Color figures from Rappaport et al., Millimeter Wave Wireless
Communications (ISBN‐13: 9780132172288)
Note: Only selected figures in the book are available in color. Black and white figures have been omitted from this file.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 19:47 Page 4 #4
4 Chapter 1 Introduction
Cellular, 50 MHz,1983
PCS, 150 MHz,1995
UNII, 300 MHz, 1997
LMDS, 1300 MHz, 1998
60 GHz Unlicensed, 5000 MHz, 1998
Figure 1.1 Areas of the squares illustrate the available licensed and unlicensed spectrum bandwidthsin popular UHF, microwave, 28 GHz LMDS, and 60 GHz mmWave bands in the USA. Other countriesaround the world have similar spectrum allocations [from [Rap02]].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 5 #5
1.1 The Frontier: Millimeter Wave Wireless 5
Figure 1.2 Wireless spectrum used by commercial systems in the USA. Each row represents a decade in frequency. For example, today’s 3G and 4G cellular and WiFi carrier frequencies are mostlyin between 300 MHz and 3000 MHz, located on the fifth row. Other countries around the worldhave similar spectrum allocations. Note how the bandwidth of all modern wireless systems (throughthe first 6 rows) easily fits into the unlicensed 60 GHz band on the bottom row [from [Rap12b] U.S.Dept. of Commerce, NTIA Office of Spectrum Management].
AM radio
FM radio
Wi-Fi
3G / 4G LTEcellular
TV broadcast
28 GHz – LMDS(5G cellular)
Active CMOS IC research
77 GHzvehicular
radar
60 GHz unlicensedWiGig (802.11 ad)
UNITEDSTATESFREQUENCY ALLOCATIONSTHE RADIO SPECTRUM
38 GHz5G cellular
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 7 #7
1.1 The Frontier: Millimeter Wave Wireless 7
Operating frequency (GHz)
Exp
ecte
d at
mos
pher
ic lo
ss (
dB/k
m)
010−2
10−1
100
101
102
50 100 150 200 250 300 350 400
Figure 1.3 Expected atmospheric path loss as a function of frequency under normal atmosphericconditions (101 kPa total air pressure, 22 Celsius air temperature, 10% relative humidity, and0 g/m3 suspended water droplet concentration) [Lie89]. Note that atmospheric oxygen interactsstrongly with electromagnetic waves at 60 GHz. Other carrier frequencies, in dark shading, exhibitstrong attenuation peaks due to atmospheric interactions, making them suitable for future short-range applications or “whisper radio” applications where transmissions die out quickly with distance.These bands may service applications similar to 60 GHz with even higher bandwidth, illustrating thefuture of short-range wireless technologies. It is worth noting, however, that other frequency bands,such as the 20-50 GHz, 70-90 GHz, and 120-160 GHz bands, have very little attenuation, well below1 dB/km, making them suitable for longer-distance mobile or backhaul communications.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 9 #9
1.1 The Frontier: Millimeter Wave Wireless 9
2.5 mm
3.5 mm
Figure 1.4 Block diagram (top) and die photo (bottom) of an integrated circuit with four transmitand receive channels, including the voltage-controlled oscillator, phase-locked loop, and local oscillatordistribution network. Beamforming is performed in analog at baseband. Each receiver channel containsa low noise amplifier, inphase/quadrature mixer, and baseband phase rotator. The transmit channel alsocontains a baseband phase rotator, up-conversion mixers, and power amplifiers. Figure from [TCM+11],courtesy of Prof. Niknejad and Prof. Alon of the Berkeley Wireless Research Center [ c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 10 #10
10 Chapter 1 Introduction
Figure 1.5 Third-generation 60 GHz WirelessHD chipset by Silicon Image, including the SiI6320 HRTX Network Processor, SiI6321 HRRX Network Processor, and SiI6310 HRTR RF Transceiver. These chipsets are used in real-time, low-latency applications such as gaming and video, and provide 3.8 Gbps data rates using a steerable 32 element phased array antenna system (courtesy of Silicon Image) [EWA+11] [©c IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 10 #10
10 Chapter 1 Introduction
1990
III-V HBT
Si CMOSSiGe HBT
III-V HEMT
0
100
200
300
400
f T (
GH
Z)
500
600
700
800
1995 2000Year
2005 2010
Figure 1.6 Achievable transit frequency (fT ) of transistors over time for several semiconductortechnologies, including silicon CMOS transistors, silicon germanium heterojunction bipolar transistor(SiGe HBT), and certain other III-V high electron mobility transistors (HEMT) and III-V HBTs.Over the last decade CMOS (the current technology of choice for cutting edge digital and analogcircuits) has become competitive with III-V technologies for RF and mmWave applications [figurereproduced from data in [RK09] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 11 #11
1.1 The Frontier: Millimeter Wave Wireless 11
Figure 1.7 Wireless personal area networking. WPANs often connect mobile devices such as mobilephones and multimedia players to each other as well as desktop computers. Increasing the data-ratebeyond current WPANs such as Bluetooth and early UWB was the first driving force for 60 GHzsolutions. The IEEE 802.15.3c international standard, the WiGig standard (IEEE 802.11ad), and theearlier WirelessHD standard, released in the 2008–2009 time frame, provide a design for short-rangedata networks (≈ 10 m). All standards, in their first release, guaranteed to provide (under favorablepropagation scenarios) multi-Gbps wireless data transfers to support cable replacement of USB, IEEE1394, and gigabit Ethernet.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 12 #12
12 Chapter 1 Introduction
Figure 1.8 Multimedia high-definition (HD) streaming. 60 GHz provides enough spectrum resourcesto remove HDMI cables without sophisticated joint channel/source coding strategies (e.g., compres-sion), such as in the wireless home digital interface (WHDI) standard that operates at 5 GHzfrequencies. Currently, 60 GHz is the only spectrum with sufficient bandwidth to provide a wirelessHDMI solution that scales with future HD television technology advancement.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 12 #12
12 Chapter 1 Introduction
wired network
WLAN access point
Figure 1.9 Wireless local area networking. WLANs, which typically carry Internet traffic, are a popu-lar application of unlicensed spectrum. WLANs that employ 60 GHz and other mmWave technologyprovide data rates that are commensurate with gigabit Ethernet. The IEEE 802.11ad and WiGigstandards also offer hybrid microwave/mmWave WLAN solutions that use microwave frequencies fornormal operation and mmWave frequencies when the 60 GHz path is favorable. Repeaters/relays willbe used to provide range and connectivity to additional devices.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 13 #13
1.1 The Frontier: Millimeter Wave Wireless 13
Figure 1.10 Wireless backhaul and relays may be used to connect multiple cell sites and subscriberstogether, replacing or augmenting copper or fiber backhaul solutions.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 15 #15
1.1 The Frontier: Millimeter Wave Wireless 15
700 MHz
BandUplink(MHz)
Downlink(MHz)
Carrier bandwidth(MHz)
AWS
GSM 900
GSM 1800
PCS 1900
Cellular 850
Digitaldividend
UMTS core
IMTextension
746–763
1710–1755
880–915
1710–1785
1850–1910
824–849
470–854
1920–1980
2500–2570
776–793
2110–2155
925–960
1805–1880
1930–1990
869–894
2110–2170
2620–2690
1.25 5 10 15 20
1.25 5 10 15 20
1.25 5 10 15 20
1.25 5 10 15 20
1.25 5 10 15 20
1.25 5 10 15 20
1.25 5 10 15 20
1.25 5 10 15 20
1.25 5 10 15 20
Figure 1.11 United States spectrum and bandwidth allocations for 2G, 3G, and 4G LTE-A (long-termevolution advanced). The global spectrum bandwidth allocation for all cellular technologies does notexceed 780 MHz. Currently, allotted spectrum for operators is dissected into disjoint frequency bands,each of which possesses different radio networks with different propagation characteristics and buildingpenetration losses. Each major wireless provider in each country has, at most, approximately 200 MHzof spectrum across all of the different cellular bands available to them [from [RSM+13] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 16 #16
16 Chapter 1 Introduction
Interferingbasestation
LOScommunication
mmWaverelay
mmWavebackhaul
Servingbasestation
NLOScommunication
Figure 1.12 Illustration of a mmWave cellular network. Base stations communicate to users (andinterfere with other cell users) via LOS, and NLOS communication, either directly or via heteroge-neous infrastructure such as mmWave UWB relays.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 18 #18
18 Chapter 1 Introduction
SIRCIM 6.0: Impulse response parameters
Nor
mal
ized
line
ar p
ower
00
0.2
0.4
0.6
0.8
1
100 200 300Excess delay (nanoseconds)
400 5000
0.1
0.2
Distan
ce (m
eters)
Path loss reference distance = 1 meterTopography = 20% obstructed LOSRMS delay spread = 65.9 nanosecocndsOperating frequency = 60.0 GHzBuilding plan = open
Figure 1.13 Long delay spreads characterize wideband 60 GHz channels and may result in severeinter-symbol interference, unless directional beamforming is employed. Plot generated with Simu-lation of Indoor Radio Channel Impulse Response Models with Impulse Noise (SIRCIM) 6.0 [from[DMRH10] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 21 #21
1.3 Emerging Applications of MmWave Communications 21
Signalling rate (Gbps)00.1
1
10
5 10 15
Cost (3.5 m)
Optics densityalready better**
2004 2007±1 2010±1 2013±2
Optics better
Copper betterCost (10 m)
Rat
io o
f opt
ical
to e
lect
rical
per
form
ance
Power
20
Figure 1.14 Comparison between optical and electrical performance in terms of cost and powerfor short cabled interconnects. The results show that optical connections are preferred to electricalcopper connections for higher data rates, assuming wires are used [adapted from [PDK+07] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 22 #22
22 Chapter 1 Introduction
Figure 1.15 MmWave wireless will enable drastic changes to the form factors of today’s computingand entertainment products. Multi-Gbps data links will allow memory devices and displays to becompletely tetherless. Future computer hard drives may morph into personal memory cards and maybecome embedded in clothing [Rap12a][Rap09][RMGJ11].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 23 #23
1.3 Emerging Applications of MmWave Communications 23
Figure 1.16 Future users of wireless devices will greatly benefit from the pervasive availability ofmassive bandwidths at mmWave frequencies. Multi-Gbps data transfers will enable a lifetime of con-tent to be downloaded on-the-fly as users walk or drive in their daily lives [Rap12a][Rap09][RMGJ11].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 24 #24
24 Chapter 1 Introduction
cloud
10-50 Gbpslinks
Hundreds ofwireless post-it
notes
Desk
OfficeBookshelf
10-50 Gbps links
Passivewireless-to-fiber node
1-10 Tbps link1-10 Tbps link
Desktopcomputer with
wirelessmemory
connectivity
Desk
Hundreds ofwireless post-it
notes withbooks/movies
OfficeBoo
kshe
lf
Passive wireless-to-fiber node
Figure 1.17 The office of the future will replace wiring and wired ports with optical-to-RF inter-connections, both within a room and between rooms of a building. UWB relays and new distributedwireless memory devices will begin to replace books and computers. Hundreds of devices will beinterconnected with wide-bandwidth connections through mmWave radio connections using adap-tive antennas that can quickly switch their beams [Rap11] [from [RMGJ11] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 25 #25
1.3 Emerging Applications of MmWave Communications 25
Vehicle-to-vehicle communication
Vehicular radar
Vehicular-to-infrastructure communication
Figure 1.18 Different applications of mmWave in vehicular applications, including radar, vehicle-to-vehicle communication, and vehicle-to-infrastructure communication.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch01 2014/8/12 10:20 Page 26 #26
26 Chapter 1 Introduction
seat back entertainment wireless Internet access
Figure 1.19 Different applications of mmWave in aircraft including providing wireless connec-tions for seat-back entertainment systems and for wireless cellular and local area networking. Smartrepeaters and access points will enable backhaul, coverage, and selective traffic control.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch02 2014/8/12 10:21 Page 92 #60
92 Chapter 2 Wireless Communication Background
Application Layer
Data LinkLayer
LogicalLink
Control
MAC
Physical Layer
Hardware Layer
Network Layer
Transport Layer
Session Layer
Presentation Layer
Figure 2.32 Reference system architecture for a communication network. The Physical Layer(PHY) is considered the lowest layer, and the Application Layer is the highest layer. We proposea new layer, called the Hardware Layer, that is below PHY, in order to account for complexitiesinvolved with the creation of new hardware and devices for mmWave communications.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 100 #4
100 Chapter 3 Radio Wave Propagation for MmWave
100
10
1
0.1
0.010 20
Sea
leve
l atte
nuat
ion
(dB
/km
)
40 60 80 100 120 140 160 180 200
Frequency (GHz)
220 240 260 280 300 320 340 360 380 400
Figure 3.1 The attenuation (dB/km) in excess of free space propagation due to absorption in air atsea level across the sub-terahertz frequency bands. The far left (unshaded) bubble shows extremelysmall excess attenuation in air for today’s UHF and microwave consumer wireless networks, and otherbubbles show interesting excess attenuation characteristics that are dependent on carrier frequency[from [RMGJ11], c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 107 #11
3.2 Large-Scale Propagation Channel Effects 107
100
50
20
10
5
2
1
0.5
0.2
0.1
0.05
0.02
0.011 2 5 10 20 50
Frequency (GHz)
Spec
ific
atte
nuat
ion
(dB
/km
)
100 200 500 1,000
Heavy rainfall 8 GHz1.4 dB attenuation @ 200 m
150 mm/h
100 mm/h
50 mm/h
25 mm/h
5 mm/h
1.25 mm/h
0.25 mm/h
Figure 3.2 Rain attenuation as a function of frequency and rain rate in the mmWave spectrum[from [AWW+13][RSM+13][ZL06] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 115 #19
3.2 Large-Scale Propagation Channel Effects 115
Figure 3.5 Indoor penetration measurements at 72 GHz in a building in Brooklyn, New York. TheTX location is marked by a triangle, the RX locations are shown as numbered dots. The primary raypaths for signal penetration are shown with arrows [reproduced from [NMSR13] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 121 #25
3.2 Large-Scale Propagation Channel Effects 121
Path 2
Zone 2Zone 2
Zone 1
d1 d2
Receiver
Receiver
Obstacle
Transmitter
Transmitter
Path 1Path 3
Path 2 is l/2 longer than path 1.Path 3 is l/2 longer than path 2.
Figure 3.6 Example of a diffraction object blocking the line-of-sight (LOS) path between trans-mitter and receiver. At millimeter wave frequencies, objects such as trees and people may inducefading and scattering as they move.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 134 #38
LOS: σ = 6.63dBNLOS: σ = 9.45dBnLOS = 3.76
nNLOS = 4.69
LOS: σ = 6.30dBNLOS: σ = 9.36dBnLOS = 3.57
nNLOS = 4.51
LOS: σ = 6.52dBNLOS: σ = 9.31dBnLOS = 3.28
nNLOS = 4.30
LOS: σ = 6.22dBNLOS: σ = 9.28dBnLOS = 3.03
nNLOS = 4.08
120
(a)
(b)
(c)
(d)
n = 5
n = 4
n = 3
nNLOS = 4.69
nLOS = 3.76
Path loss corresponding to one best power versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
120
n = 5
n = 4
n = 3
nNLOS= 4.51
nLOS= 3.57
Path loss corresponding to non-coherently combined two best powers versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
120
n = 5
n = 4
n = 3
nNLOS= 4.30
nLOS= 3.28
Path loss corresponding to coherently combined two best powers versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
120
n = 5
n = 4
n = 3
nNLOS= 4.08
nLOS= 3.03
Path loss corresponding to coherently combined three best powers versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
Figure 3.11 Scatter plots of measured 28 GHz cellular path loss in New York City[SR13][RSM+13][AWW+13][SWA+13]. The plots illustrate the reduction in path loss that can beachieved when a mobile handset using 10 steerable beams combines individual multipath signalsarriving at different angles from the same transmitter. In (a), the single best beam pointing direc-tion is used to make a link at each RX location. In (b), the two best beam pointing directionsare non-coherently combined (where the powers in each unique beam are simultaneously added). In(c) and (d), the two and three best beams, respectively, are coherently added (where the total voltagein each unique beam is simultaneously added and then squared to produce power).
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 134 #38
LOS: σ = 6.63dBNLOS: σ = 9.45dBnLOS = 3.76
nNLOS = 4.69
LOS: σ = 6.30dBNLOS: σ = 9.36dBnLOS = 3.57
nNLOS = 4.51
LOS: σ = 6.52dBNLOS: σ = 9.31dBnLOS = 3.28
nNLOS = 4.30
LOS: σ = 6.22dBNLOS: σ = 9.28dBnLOS = 3.03
nNLOS = 4.08
120
(a)
(b)
(c)
(d)
n = 5
n = 4
n = 3
nNLOS = 4.69
nLOS = 3.76
Path loss corresponding to one best power versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
120
n = 5
n = 4
n = 3
nNLOS= 4.51
nLOS= 3.57
Path loss corresponding to non-coherently combined two best powers versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
120
n = 5
n = 4
n = 3
nNLOS= 4.30
nLOS= 3.28
Path loss corresponding to coherently combined two best powers versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
120
n = 5
n = 4
n = 3
nNLOS= 4.08
nLOS= 3.03
Path loss corresponding to coherently combined three best powers versus distance
TX-RX Separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
100
100105
80
60
40
20
0
Figure 3.11 Scatter plots of measured 28 GHz cellular path loss in New York City[SR13][RSM+13][AWW+13][SWA+13]. The plots illustrate the reduction in path loss that can beachieved when a mobile handset using 10 steerable beams combines individual multipath signalsarriving at different angles from the same transmitter. In (a), the single best beam pointing direc-tion is used to make a link at each RX location. In (b), the two best beam pointing directionsare non-coherently combined (where the powers in each unique beam are simultaneously added). In(c) and (d), the two and three best beams, respectively, are coherently added (where the total voltagein each unique beam is simultaneously added and then squared to produce power).
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 136 #40
136 Chapter 3 Radio Wave Propagation for MmWave
Lobe 2
Lobe 1
120°90°
Track position 1
Track position 10
60°
30°
330°
300°270°
240°
210°
180°
150°
0°
120°90°
60°
30°
330°
300°270°
240°
210°
180°
150°
0°
Track position 5
120°90°
60°
30°
330°
300°270°
240°
210°
180°
150°
0°
Track position 21
120°90°
60°
-72-82-92-75-85-95
30°
330°
300°
Threshold270°
240°
210°
180°
150°
0°-75-85-95 -75-85-95
Figure 3.12 Four polar plots of 28 GHz propagation at track positions 1, 5, 10, and 21 along a21-step linear track with λ/2 step sizes show two lobes of received power across azimuth. Measure-ments are for a partially obstructed NLOS RX environment in downtown Brooklyn using 24.5 dBihorn antennas at both the TX and RX. The TX was placed on the rooftop of NYU’s Rogers Hall135 m away from the RX. Each dot represents the received power level at a particular RX azimuthangle. For NLOS RX locations, a threshold of 20 dB below maximum power level was defined for athreshold (shown as a solid-line circle) to determine lobe statistics, whereas 10 dB was used for theLOS threshold [reproduced from [SWA+13] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch03 2014/8/12 10:22 Page 141 #45
3.7 Outdoor Channel Models 141
−100
−110
−120
−130
−140
−150
−160
−200 −100 100Frequency (MHz)
Cha
nnel
pow
er g
ain
|H(f
)|2 (d
B)
TX location WRW-A RX location 14channel frequency response track 7
Relative threshold = 24 dBRX Atten = 0dBDistance = 70 mTX: 15°/−2.5°
RX: 0°/2.5°
Measurement 125 track 7
2000
−90
Figure 3.13 Frequency-selective fading occurs about the 38 GHz carrier frequency in outdoor urbanNLOS channels. Note the periodic 50 MHz fades in frequency about the carrier correspond to a RMSdelay spread that is approximately 20 ns. Here we see a channel that has deep fades as low as 30 dBfrom the peak channel gain.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
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3.7 Outdoor Channel Models 141
1
0.9On the campus of the university of Texas at AustinTX on roof of building WRW37.625 GHz carrier1 MHz sub-bands condiered +/- 222 MHz from carrier46.1dBm EIRP25 dBi 7° Beamwidth TX Antenna25 dBi 7° Beamwidth or 13.3 dBi dBi 38°Beamwidth RX AntennaLink distances 61 - 265 m
CDF of frequency selectivity outdoor 38 GHz cellular channel
0.8
0.7
0.6
Cum
ulat
ive
dist
ribu
tion
func
tion
0.5
0.4
0.3
0.2
−40 −30 −20 −10 01 MHz sub-band deviation from mean channel gain (dB)
10 20
0.1
0
Figure 3.14 When a channel frequency representation such as that shown in Fig. 3.13 is consideredover 1 MHz subbands (i.e., we evaluate the average channel gain at 1 MHz intervals and comparethese small intervals to the overall average channel gain across the band), we see that fading isnot severe for outdoor urban cellular mmWave channels. The time delay spread and the number ofresolvable multipath components directly contribute to the fading characteristics across the occupiedspectrum. Directional antennas change small-scale fading from today’s common Rayleigh fadingcharacteristics (for omnidirectional antennas) into much narrower fade depths over much widerfrequency bands.
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3.7 Outdoor Channel Models 145
0.9LOS 38 GHz
LOS 60 GHz
NLOS 38 GHz
NLOS 60 GHz
0.8
0.7
0.6
0.5
0.4
0.3
Prob
abili
ty R
MS
dela
y sp
read
< a
bsci
ssa
0.2
0.1
00 20 40 60
RMS delay spread (ns)
80 100
E[σLOS38]=1.2ns
E[σNLOS38]=23.6ns
E[σNLOS60]=0.8ns
E[σNLOS60]=7.4ns
Max[σLOS38]=1.3ns
Max[σNLOS38]=122ns
Max[σNLOS60]=0.9ns
Max[σNLOS60]=36.6ns
120 140
1
Figure 3.15 Differences in RMS delay spread and their distribution at 38 and 60 GHz in variousoutdoor environments [from [RBDMQ12] c© IEEE].
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146 Chapter 3 Radio Wave Propagation for MmWave
1
0.9
0.8
0.7
0.6Pr
obab
ility
RM
S de
lay
spre
ad <
abs
ciss
a0.5
0.4
0.3
0.2
0.1
00 20 40 60 80 100
RMS delay spread (ns)
LOS(25dBi RX Ant.)
LOS(13.3dBi RX Ant.)
NLOS(13.3dBi RX Ant.)
NLOS(25dBi RX Ant.)
120 140 160 180 200 220 240
E[σLOS25]=1.5ns
E[σNLOS25]=14.3ns
E[σLOS13]=1.9ns
E[σNLOS13]=13.7ns
Max[σLOS25]=15.4ns
Max[σNLOS25]=225ns
Max[σLOS13]=15.5ns
Max[σNLOS13]=166ns
Figure 3.16 Greater transmitter antenna heights resulted in decreased 90% RMS delay spreadcompared with situations in which the 38 GHz transmitter is near the ground, and the worst-caseRMS delay spread was found to be 225 ns on a Texas college campus using the tallest transmitterantenna height [from [RSM+13] [RGBD+13] c© IEEE].
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146 Chapter 3 Radio Wave Propagation for MmWave
180
150
120
90
−30
−60
−90
−120
−150
−180
−180 −150 −120 −90 −60 −30 30 90 150 180
TX
RX
Positiveangles
Positiveangles
Negativeangles
Negativeangles
TX azimuth angleRX
azi
mut
h an
gle
0
60
30
0 60 120
38GHz60GHz
Figure 3.17 MmWave applications in which the transmitter and receiver are close to the ground(such as peer-to-peer or vehicle-to-vehicle) will provide a wide distribution of angles at which linksmay be established [from [RBDMQ12][RGBD+13] c© IEEE].
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3.7 Outdoor Channel Models 147
100
150
120
90
30
0−30 30 60 90 120 150 180
Fewer negativeRX azimuth linksbecause of ENSbuilding to leftof RX antenna
0°
0°
TX azimuth angle
RX
azi
mut
h an
gle
Transmitterantennaazimuth
40
25 20 15 10 5 0
30
2010
0
−60
−60 −30
−90
−120
−150
−180
Link distribution10° TX angle bins
0
Lin
k di
stri
butio
n10° R
X a
ngle
bin
s
Lin
kco
unt
Linkcount
60
Figure 3.18 The antenna pointing angles found with a 37.625 GHz carrier and highly directionalantennas at the receiver and transmitter. The transmitter was elevated to 18 m [from [RBDMQ12][RGBD+13] c© IEEE].
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148 Chapter 3 Radio Wave Propagation for MmWave
120 38 GHz
38 GHz linear fit60 GHz linear fit
0.09 θcomb + 1.1
0.13 θcomb + 11
60 GHz
100
80
60
RM
S de
lay
spre
ad (
ns)
40
20
0
Angle combination = |TX Azimuth| + |RX Azimuth| (deg)
0 50 100 150 200
Figure 3.19 Steeper azimuth pointing angles are associated with higher RMS delay spreads for out-door peer-to-peer channels. The measurements from this plot were taken with 25 dBi 7 beamwidthhorns at the transmitter and receiver, and with link distances from 19 to 129 m [from [RBDMQ12]c© IEEE].
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148 Chapter 3 Radio Wave Propagation for MmWave
0 50 100 150 200
200
25dBi RX Ant.
RX Ant.
Linear fit for 25dBi
Linear fit for
0.22θcomb
0.14θcomb
θcomb = |TX azimuth| + |RX azimuth| + |∆Elev.| (deg)
−0.56
+ 3.7
13.3dBi
13.3dBi
RM
S de
lay
spre
ad (
ns)
150
100
50
0
Figure 3.20 Steeper antenna pointing angles are associated with higher RMS delay spreads. Thesemeasurements were taken at 38 GHz with at 25 dBi TX antennas, and a 25 dBi or 13.3 dBi RXantennas. Link distances ranged just beyond 900 m [from [RQT+12] c© IEEE].
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150 Chapter 3 Radio Wave Propagation for MmWave
30
10
20
30
40
Path
loss
abo
ve 3
m r
efer
ence
(dB
)
50
60
LOS
n = 5
n = 4
n = 3
n = 2snLOS = 2dB
snNLOS = 10.12dB
snNLOS-best = 10.16dB
NLOS
nLOS = 2.25
nNLOS-all = 4.22
nNLOS-best = 3.76
70
Transmitter to receiver separation (m)101 102
Figure 3.21 Due to the very high value for the break-point distance, LOS links at mmWavefrequencies are very close to free space in terms of path loss. This plot was generated with highlydirectional antennas at the receiver and transmitter with 25 dBi gain and 7 beamwidths at 60 GHz[from [RBDMQ12] c© IEEE]. Note that the oxygen absorption causes the path loss exponent to beslightly greater than 2.0.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
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150 Chapter 3 Radio Wave Propagation for MmWave
70 LOS
NLOS
n = 5
n = 4
n = 3
n = 2
nLOS = 2.0
σnLOS = 3.79dBσnLOS = 11.72dBσnNLOS-best = 8.57dB
nNLOS-all = 4.57
nNLOS-best = 3.71
60
50
40
30
20
10
0
Transmitter to receiver separation (m)
Path
loss
abo
ve 3
m r
efer
ence
(dB
)
3 101 102
Figure 3.22 This plot shows measured path loss values for 38 GHz peer-to-peer applications withhighly directional 25 dBi 7 beamwidth horn antennas [from [RBDMQ12] c© IEEE].
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3.7 Outdoor Channel Models 151
σnLOS = 4.55dB
σnLOS-all = 11.55dB
σnLOS = 13.39dB
σnNLOS-best = 11.69dB
LOS
LOS partiallyobstructedNLOS
Best NLOS
nLOS-all = 2.30
nLOS = 1.89
nNLOS-all = 3.86n=5
n=4
n=3
n=2
nNLOS-best = 3.2
70
80
60
50
40
30
20
10
0
Path
loss
abo
ve 3
m r
efer
ence
(dB
)
Transmitter to receiver separation (m)
5 101 102 103
Figure 3.23 When a highly directional antenna is used at the receiver, LOS links will be very closeto free space but NLOS links may be more heavily attenuated. This plot is for 38 GHz and themeasurements used the same highly directional antennas at both the transmitter and receiver [from[RQT+12] c© IEEE].
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3.7 Outdoor Channel Models 151
70
60
50
40
30
20
10
n = 5
n = 4
n = 3
n = 2
Transmitter to receiver separation (m)
Path
loss
abo
ve 5
m r
efer
ence
(dB
)
05 101 102 103
Clear LOS
σnLOS = 3.45dB
σnLOS-all = 9.4dB
σnLOS = 11.03dB
σnLOS-best = 8.37dB
NLOS
LOS partiallyobstructed
Best NLOS
nLOS =1.90
nLOS-all =2.21
nNLOS-best =2.56
nNLOS-all =3.18
Figure 3.24 This plot was generated from measurements using a 25 dBi 7 beamwidth horn TXantenna and a less directional 13.3 dBi 40 beamwidth horn at the receiver. NLOS paths are signif-icantly stronger as the receiver cannot filter out multipath as effectively as when a more directionalantenna is used [RQT+12] c© IEEE].
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3.7 Outdoor Channel Models 153
1 30 100 20040
60
80
100
120
140
160
T−R separation (m)
Path
loss
(dB
)
28 GHz omnidirectional PL model 1 m − Manhattanfor (RX at 1.5 m AGL)
NLOSLOSn
NLOS = 3.4, σ
NLOS = 9.7 dB
nLOS
= 2.1, σLOS
= 3.6 dB
(α, β, σ) = (79.2 dB, 2.6, 9.6 dB)n
FreeSpace
UHF (1900 MHz) nNLOS
= 2.6,
σNLOS
= 7.7 dB
Figure 3.25 28 GHz omnidirectional close-in free space reference distance (d0 = 1 m) and floatingintercept path loss models for a non-line of sight (NLOS) urban environment with a receiver antenna1.5 m above ground. A comparison is made to path loss in a 1.9 GHz urban NLOS environment asreported in [BFR+92] [FBRSX94].
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154 Chapter 3 Radio Wave Propagation for MmWave
160
70
80
90
100
110
120
130
140
150Omni NLOS Path Loss
28 GHz omni NLOS PL model 1 m - Manhattan
nNLOS = 3.4, σNLOS = 9.7 dB
(α, β, σ) = (79.2
10
T-R separation (m)
Path
loss
(dB
)
100
Figure 3.26 28 GHz omnidirectional path loss model from which the TX and RX antenna gains havebeen removed. The close-in free space reference distance model with respect to a 1 m free spacereference distance, and the floating intercept (α, β) model from [RRE14] is shown for distancesranging from 30 to 200 m.
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154 Chapter 3 Radio Wave Propagation for MmWave
dB, 2.6, 9.6)
10
10
20
30
40
50
60
70
80Omni LOS Path Loss
28 GHz omni LOS PL model 1 m - Manhattan
nLOS = 2.1, σLOS = 3.6 dB
10
T-R separation (m)
Path
loss
abo
ve 1
m F
S re
fere
nce
(dB
)
100
n=3
n=2
Figure 3.27 28 GHz omnidirectional path loss model from which the TX and RX antenna gainshave been removed. The close-in free space reference distance model with respect to a 1 m freespace reference distance is shown. Note that one point at 100 m had excessive path loss due to thefact that the antennas were not aligned on boresight at this location. By removing this single point,it is evident that the LOS path loss exponent is very close to 2.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
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3.7 Outdoor Channel Models 155
1 10 20 30 100 2000
20
40
60
80
100
120
T−R separation (m)
Path
loss
abo
ve 1
m F
S re
fere
nce
(dB
)
28 GHz Manhattan path loss versus distance
n = 2
n = 3
n = 4
n = 5NLOS Path Loss
Path LossLOS−Co Path LossLOS−Cross Path Lossn
NLOS = 4.4, σ
NLOS = 10.0 dB
nNLOS−Best
= 3.7, σNLOS−Best
= 9.2 dB
nLOS−Co
= 1.8, σLOS−Co
= 0.99 dB
nLOS−Cross
= 3.6, σLOS−Cross
= 5.1 dB
Figure 3.28 28 GHz Manhattan single beam path loss measurements as a function of T-R sep-aration distance using 24.5 dBi horn antennas with 10.9 half-power beam width at both the TXand RX and 15 dBi (28.8 degree HPBW) horn antennas at both the TX and RX. NLOS path lossesinclude LOS non-boresight and truly NLOS measurements. Co-polarized and cross-polarized LOSmeasured path losses are also shown. The close-in free space reference distance model with respectto a 1 m free space reference distance is shown. All data points represent path loss values calculatedfrom recorded PDP measurements.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
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3.7 Outdoor Channel Models 159
1 30 100 200
80
100
120
140
160
T−R separation (m)
Path
loss
(dB
)
73 GHz omnidirectional PL model 1 m − Manhattanfor hybrid (RX at 2 m and 4.06 m AGL)
NLOSLOSn
NLOS= 3.4, σ
NLOS= 7.9 dB
nLOS
= 2.0, σLOS
= 4.8 dB
(α, β, σ) = (80.6 dB, 2.9, 7.8 dB)n
FreeSpace
Figure 3.29 73 GHz omnidirectional path loss model from which the TX and RX antenna gainshave been removed for a combination of cellular and backhaul (hybrid) RX antenna heights. Theclose-in free-space reference distance model for d0 = 1 m and the floating intercept (α, β) modelfrom [RRE14] over 30-200 m are shown.
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160 Chapter 3 Radio Wave Propagation for MmWave
1 10 20 30 100 20040
60
80
100
120
140
160
T−R separation (m)
Path
loss
(dB
)
73 GHz omnidirectional PL model 1 m − Manhattanfor access (RX at 2 m AGL)
NLOSLOSnNLOS = 3.3, σNLOS = 7.6 dB
nLOS
= 2.0, σLOS
= 5.2 dB
(α, β, σ) = (81.9 dB, 2.7, 7.5 dB)n
FreeSpace(73 GHz)
UHF (1900 MHz) nNLOS
=
σNLOS
= 7.7 dB
Figure 3.30 73 GHz omnidirectional path loss model from which the TX and RX antenna gainshave been removed for mobile RX antenna heights of 2 m. The close-in free-space reference distancemodel for d0 = 1 m and the floating intercept model (α, β) model from [RRE14] over 30-200 mare shown. A comparison is made to path loss in a 1.9 GHz urban NLOS environment as reported in[BFR+92].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
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3.7 Outdoor Channel Models 161
1 30 100 200
80
100
120
140
160
T−R separation (m)
Path
loss
(dB
)
73 GHz omnidirectional PL model 1 m − Manhattanfor backhaul (RX at 4.06 m AGL)
NLOSLOSn
NLOS= 3.5, σ
NLOS= 7.9 dB
nLOS
= 2.0, σLOS
= 4.2 dB
(α, β, σ) = (84.0 dB, 2.8, 7.8 dB)n
FreeSpace
Figure 3.31 73 GHz omnidirectional path loss model from which the TX and RX antenna gainshave been removed for backhaul RX antenna heights of 4.06 m. The close-in free-space referencedistance model for d0 = 1 m and the floating intercept (α, β) model from [RRE14] over 30-200 mare shown.
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162 Chapter 3 Radio Wave Propagation for MmWave
1 10 20 30 100 20070
90
110
130
150
170
190
n = 2
n = 3
n = 4
n = 5
T−R separation (m)
Path
loss
(dB
)73 GHz unique pointing angle path loss versus distance with
RX height: 2 m using 27 dBi, 7° 3dB BW TX antennas and
27 dBi, 7° 3dB BW RX antennas in Manhattan
NLOSNLOS−bestLOSnNLOS = 4.7, σ
NLOS 12.6 dB
nNLOS−best = 3.6, σNLOS−best = 10.6 dB
nLOS = 2.2, σLOS 5.2 dB
NLOS Omni: α = 81.9 dB, β = 2.7,σ = 7.5 dB
Figure 3.32 New York City cellular RX height (2 m) path loss measurements at 73 GHz as a func-tion of T-R separation distance using vertically polarized 27 dBi, 7 half-power beam width TX andRX antennas. All data points represent path loss values calculated from recorded PDP measurements.Crosses indicate all NLOS pointing angle data points, diamonds indicate best NLOS pointing angledata points for each RX location and each T-R combination, and circles indicate LOS data points.The measured path loss values are relative to a 1 m free-space close-in reference distance. NLOS PLEsare calculated for the entire data set and also for the best recorded link. LOS PLEs are calculated forstrictly boresight-to-boresight scenarios. n values are PLEs and σ values are shadow factors. The solidline spanning 30 to 200 m is the omnidirectional (α, β) model from [RRE14] [ALS+14] depicted inFig. 3.30.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
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3.7 Outdoor Channel Models 163
1 10 20 30 100 20070
90
110
130
150
170
190
n = 2
n = 3
n = 4
n = 5
T−R separation (m)
Path
loss
(dB
)73 GHz unique pointing angle path loss vs. distance with
RX height: 4.06 m using 27 dBi, 7° 3dB BW TX antennas and
27 dBi, 7° 3dB BW RX antennas in Manhattan
NLOSNLOS−bestLOSn
NLOS = 4.7, σ
NLOS12.7 dB
nNLOS−best
= 3.7, σNLOS−best
= 11.2 dB
nLOS
= 2.4, σLOS
6.3 dB
NLOS Omni: α = 84.0 dB, β = 2.8,σ = 7.8 dB
Figure 3.33 New York City backhaul measurements with RX heights of 4.06 m path losses at73 GHz as a function of T-R separation distance using vertically polarized 27 dBi, 7 half-power beamwidth TX and RX antennas. All data points represent path loss values calculated from recorded PDPmeasurements. Crosses indicate all NLOS pointing angle data points, diamonds indicate best NLOSpointing angle data points for each RX location and each T-R combination, and circles indicateLOS data points. The measured path loss values are relative to a 1 m free-space close-in referencedistance. NLOS PLEs are calculated for the entire data set and also for the best recorded link. LOSPLEs are calculated for strictly boresight-to-boresight scenarios. n values are PLEs and σ valuesare shadow factors. The solid line spanning 30 to 200 m is the omnidirectional (α, β) model from[RRE14] depicted in Fig. 3.31.
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164 Chapter 3 Radio Wave Propagation for MmWave
0−75
−70
−65
−60
−55
Void 1:Duration: 2.7 ns
Void 2:Duration: 9.5 ns
Void 3:Duration: 13.3 ns
Void 4:Duration: 23.9 ns
20 40 60
Time (ns)
Rec
eive
d po
wer
(dB
m/n
s)
80 100
Cluster 5:Duration: 1.9 ns1 Sub-path
Cluster 3:Duration: 2.00 ns2 Sub-paths
Cluster 1:Duration: 9.1 ns7 Sub-paths
Cluster 4:Duration: 11.81 ns8 Sub-paths
Cluster 2:Duration: 31.0 ns7 Sub-paths
Figure 3.34 Illustration of some of the key temporal modeling parameters used for modeling thetemporal clusters in an omnidirectional SSCM wideband mmWave channel. This example shows fivetime clusters, with time durations ranging from 2 to 31 ns, and voids between clusters ranging from2.7 to 23.9 ns [SR14a].
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3.7 Outdoor Channel Models 165
Lobe segment
Lobe Azimuth Spread
RMS LAS
60°
-84 -74 -64 -54
30°
90°dBm
120°
150°
180°
210°
240°
270°
300°
330°
0°
AOA
Lobe power
Lobe
Figure 3.35 Illustration of some of the key spatial modeling parameters used to model thespatial lobes in an omnidirectional SSCM wideband mmWave channel. The polar plot (in theazimuthal/horizontal plane only) shows five distinct lobes with various lobe azimuth spreads andAOAs. Each dot is a lobe angular segment simulated for a particular discrete pointing angle andrepresents the total integrated received power over a particular beam width (and corresponds to thearea under a PDP for the particular RX pointing angle). The lobe power is the sum of powers fromeach segment within the lobe (e.g., the sum of powers from each lobe segment in a lobe).
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168 Chapter 3 Radio Wave Propagation for MmWave
60
Path
loss
abo
ve 3
m r
efer
ence
(dB
)50
40
30
20
10
0101
Transmitter to receiver separation (m)
npeer-NLOS = 4.19
nveh-LOS = 2.66
npeer-LOS = 2.23
Peer-to-peer NLOS
Peer-to-peer LOS
Vechicle LOS
n=5
n=4
n=3
n=2
Vechicle NLOS
nveh-NLOS = 7.17
102
Figure 3.36 Path loss for 60 GHz for peer-to-peer applications and communication from a ground-based transmitter to a receiver in a vehicle. These measurements used highly directional 25 dBi 7
beamwidth antennas as the transmitter and receiver [from [BDRQL11] c© IEEE].
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168 Chapter 3 Radio Wave Propagation for MmWave
All NLOS meas.
E[σNLOS] = 6.48 ns
Max[σNLOS] = 36.6 ns
E[σLOS] = 0.76 ns
E[σpeer] = 6.02 ns
E[σvehicle] = 2.73 ns
Max[σLOS] = 0.88 ns
Vehicleall measurements
RMS delay spread (ns)
Prob
abili
ty R
MS
dela
y sp
read
< A
bsci
ssa
00
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
5 10 15 20 25 30 35 40
Peer-to-peerall measurements
All LOS meas.
Figure 3.37 These measurements used highly directional 25 dBi 7 beamwidth antennas at thetransmitter and receiver. When the transmitter communicates to a receiver inside a vehicle, muchlower RMS delay spreads result than when the transmitter and receiver are in the open [from[BDRQL11] c© IEEE].
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172 Chapter 3 Radio Wave Propagation for MmWave
Excess time
Tcurser−t
AOA’s of rays areclustered
Clustered in space
Pre-curserpower-growth
Numberof post-curser
rays Nb
Cluster arrivalrate Λ
Post-curserpower-decay
Post-curser MPCarrival rate λbPre-Curser MPC
arrival rate λ f
NLOS clusterCluster decayrate Γ Γe
Channel impulseresponse
LOS componentgain of β
φ
Numberof pre-curser
rays Nf
γf
t−Tcurser
γf
Receiveing antenna
Clustered in time−t
e
e
Figure 3.38 Representation of key parameters used to specify multipath channels. Statistics ofthe key channel parameters are generated from measured data, as collected by wideband channelsounders, to determine the temporal and spatial channel models that can be used by researchers andstandard bodies for modem design and signaling protocols.
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PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 203 #17
4.3 The On-Chip Antenna Environment 203
Metal guard ringSlotted metal
Figure 4.6 There are several considerations for on-chip antennas related to CMOS productionrules: 1) All metal layers must meet a minimum fill requirement. This is reflected in the figure by thefact that there are no large portions of the chip left empty (the lighter-shaded portions of the figure).2) A metal guard ring must often surround the chip to prevent damage during dicing. 3) Largeareas of metal must be slotted to meet design rules. 4) Metal structures must meet a minimum sizerequirement, which in practice is usually satisfied by most designs.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 207 #21
4.3 The On-Chip Antenna Environment 207
100 −10
−20
−30
−40
−50
80
60
40
20
00 50 100 150
Si substrate thickness [µm]
Rad
iatio
n ef
fici
ency
(e r
ad)
(%)
Tra
nsm
issi
on c
oeff
icie
nt (
S 21)
[dB
]
200 250
L= 6 mmp= 10Ω cmd= 5 mm
300
Figure 4.10 This figure indicates that the efficiency of on-chip antennas is reduced greatly by athick substrate [from [KSK+09] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 209 #23
4.4 In-Package Antennas 209
tanδd=0.001100
10
1
0.1
0.011 10 100
Resistivity (Ω−cm)
Atte
nuat
ion
(dB
/mm
)
1k 10k 100k
tanδd=0.003
tanδd=0.01
tanδd=0.03
Figure 4.12 For low resistivity substrates, the loss due to currents carried by substrate dopants(i.e., conductive losses) is the major loss mechanism hurting performance [from [LKCY10] c© IEEE].
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PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 216 #30
216 Chapter 4 Antennas and Arrays for MmWave Applications
120
50
40
30
20
40 50 60 70 80Width, W (µm)
Thickness (µm)
Ret
urn
loss
(+d
B)
Ret
urn
loss
(+d
B)
90 10040 50 60 70 80 90 100
110
10
0
50
45
35
25
55
40
30
10
8
6
4
2
0
100
L = 4 mm
L = 3 mm
L = 4 mm
L = 5 mm
L = 6 mm
L = 3 mmW = 100 µm
W = 90µm
W = 80 µm
W = 70 µm
L = 6 mm
Cr/Au: 10 nm/60 nm
Cr/Au: 10 nm/60 nm
Cr/Au: 10 nm/60 nm
L = 5 mm
L = 3 mm
L = 2 mm
80
60
40
20
00 5 10 15
Frequency, f (GHz)
Wid
th, W
(µm
)
Len
gth,
L (
mm
)
20 25 30 5 10 15 20Frequency, f (GHz)
25 30 35 4035
Figure 4.18 [MHP+09] presented these plots for the design of a planar dipole antenna on a625 µm GaAs substrate with relative permittivity of 12.9 and 625 µm thick [reproduced from[MHP+09] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 222 #36
222 Chapter 4 Antennas and Arrays for MmWave Applications
0X
Y
30−30
Silicon substrate
Patch (Pad layer)
Ground (M1−M5)Slot (2 µm wide)
−7.00
−14.00
−21.00
−28.00
60−60
dB(Gain total)
Phi
XZ plane (Phi = 0°)
dB(Gain total)YZ plane (Phi = 90°)
90
120
150−180
−150
−120
−90
YX
Z
Figure 4.26 This figure shows that a patch antenna typically radiates above the top metal layerof the antenna. The top figure illustrates the metal slots that are usually required for on-chip patchantennas due to their large size. In the lower figure, [CGLS09] used two parallel metal strips on theedge of the patch to increase bandwidth. [This figure is a combination of figures from the literature([SCS+08] above, [CGLS09] below) c© IEEE.]
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 223 #37
4.5 Antenna Topologies for MmWave Communications 223
Feeding via
Patch ground planePackage substrate
Patch
Chip dielectric
Chip substrate
H1
H2
εr2
εr1Feeding substrate
Packaging
Patch
W
L
A
Air
Ball connector
CPWFA
Hcav
Scav
Figure 4.27 There are various methods for feeding an in-package patch from a packaged chip.The ball connector (left) may, for example, be used in a flip chip connection. [KLN+11] found thatthis type of ball connector improves with a smaller radius of the ball and a smaller metal pad for theball (represented here as a small rectangular piece of metal below the ball) [right portion from[HRL10] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 228 #42
228 Chapter 4 Antennas and Arrays for MmWave Applications
Lc
Wc
Figure 4.34 This element was cascaded periodically below an on-chip microstrip antenna to forman AMC to achieve a gain of −1.5 dBi [from [CGLS09] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 255 #69
4.8 Characterization of On-Chip Antenna Performance 255
Manyscatterers
Measurementarea
Proberadiation
Metalstage
Figure 4.58 A probe station is often used to characterize mmWave antennas. These measurementsmay be inaccurate due to radiation from probes and scattered fields from the many surrounding metalobjects [from [MBDGR11] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 256 #70
256 Chapter 4 Antennas and Arrays for MmWave Applications
180°
Direction of movement
Probe Probe
90°
0°
C1
C2
Figure 4.59 The two antennas were swept in angle across each other. The chips on which theantennas were fabricated are represented by squares, and the antennas are represented by smallerblack boxes [from [MBDGR11] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 256 #70
256 Chapter 4 Antennas and Arrays for MmWave Applications
With de-embedding
No de-embedding
Yagi gain at chip horizon (65 GHz)
Angle (degrees)
Ant
enna
gai
n (d
Bi)
Figure 4.60 The de-embedding method indicates that the on-chip Yagi pattern was distorted bythe presence of other nearby metal structures [from [MBDGR11] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch04 2014/8/12 10:22 Page 257 #71
4.9 Chapter Summary 257
–15–100 –75 –50 –25 0
Curve infoYagi onlyYagi w/ metal barYagi w/ metal in Q2Yagi w/ metal in all QMeasured
Phi [deg]
Q2
xx
zy
y
yx
60 GHz Yagi antenna pattern normalized
Ant
enna
gai
n-ph
i dir
ectio
n (d
Bi)
25 50 75 100
–13
–10
–8
–5
–3
0
Figure 4.61 Simulations confirmed measurements that indicated the distortion of the antennapattern was caused by surrounding metal structures. This indicates that isolation between integratedantennas and other nearby structures on the chip or in the package is key to successful design [from[MBDGR11] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 271 #13
5.4 Simulation, Layout, and CMOS Production of MmWave Circuits 271
Probe
PadT-line
Top view
T-line ABCD matrixPad Pad
Z0, β
60 µm 60 µm340 µ
m550 µm
Side viewCpad
Figure 5.7 The S-parameters of a transmission line can be used to determine the effective relativepermittivity and loss tangent of a CMOS process. The effects of the probe pads must be de-embeddedfor this measurement to be accurate [from [GJRM10] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 272 #14
272 Chapter 5 MmWave RF and Analog Devices and Circuits
Extracted IC dielectric permittivity from measured probepad and transmission line
T line epsilon (real part)
Probe pad epsilon (enhanced)
Probe pad epsilon (simple)
5.5R
elat
ive
perm
ittiv
ity ε
r 5
4.5
4
3.5
10 20 30 40 50Frequency (GHz)
60 70 803
Figure 5.8 The effective relative permittivity may be measured in a number of ways and is a vitalparameter for the design of passive structures [from [GJRM10] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 273 #15
5.5 Transistors and Transistor Models 273
Loss tangent of IC dielectric from extracted transmissionline complex permittivity
Los
s ta
ngen
t (ta
nδ)
0.2
0.15
0.1
0.05
00 10 20 30 40
Frequency (GHz)50 60 70
Figure 5.9 The effective loss tangent is a vital parameter to predict loss of passive structures [from[GJRM10] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 298 #40
298 Chapter 5 MmWave RF and Analog Devices and Circuits
Metal removed tomeet CMP metalfill requirements
CPW signal traces
Figure 5.26 A common ground plane is evident in this layout. Portions of the metal have beenremoved in order to meet metal fill requirements [based on a figure from [MTH+08] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 315 #57
5.9 Sensitivity and Link Budget Analysis for MmWave Radios 315
1E+11
Frequency (Hz)
00
–2
–4
–6
–8
–10
–12
5E+10
S11 dB
Figure 5.42 The amplifier accepts energy in only a certain range of frequencies.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 315 #57
5.9 Sensitivity and Link Budget Analysis for MmWave Radios 315
Frequency (Hz)
2.00E + 01
1.50E + 01
1.00E + 01
5.00E + 00
0.00E + 00
– 5.00E + 00
– 1.00E + 01
5E+10 1E+11
S21 dB
0
Figure 5.43 The value of S21 gain is only high in a certain band of frequencies.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 316 #58
316 Chapter 5 MmWave RF and Analog Devices and Circuits
Baseband
Mixer
Antennas
Mixer
Baseband
VCOVCO
Poweramp. (PA)
Low noiseamp (LNA)
Figure 5.44 A direct conversion architecture for a transmitter and receiver. This is a populararchitecture for today’s cellphones. In many designs, the VCO is part of a phase-locked loop (PLL).
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 320 #62
320 Chapter 5 MmWave RF and Analog Devices and Circuits
Ideal
Real(compression)
Input power, mW
Output power, mW20
18
16
14
12
10
8
6
4
2
00 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8
Real vs. ideal output power
Psat
OP1dB
P1dB
1dB
Figure 5.46 The non-linearity of most devices results in the compression of the output power ofthe fundamental harmonic.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 331 #73
5.11 Analog MmWave Components 331
3
2
1
0
0
0.5
1
1.5
0
0 1 2 3 4 5 6 7
1 2 3
Amplifier classes
Normalized time
Normalized time
Drain conduction current
IAIBIABIC
Threshold voltageIn
put v
olta
ge
Nor
mal
ize
drai
nco
nduc
tion
curr
ent
4 5 6 7
RL
Vdd
Vin = Vin,DC + Vin,RF
VinAVinBVinABVinC
Figure 5.50 The bias point of the amplifier determines the amplifier’s class. Class A amplifiersconduct current over the entire period. Class B amplifiers conduct over half the period, Class Cconduct over less than half the period, and Class AB conduct over more than half the period, butless than the entire period.
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PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 348 #90
348 Chapter 5 MmWave RF and Analog Devices and Circuits
LO v
olta
ge
Normalized time
Normalized time
Bet
a fa
ctor
Beta
Beta
1 2
6
5
4
3
2
1
0
–1
–2
–0.2
0
0.2
0.4
0.6
0.8
1
1.2
3 4 5 60
1 2 3 4 5 60
VddRD
VoutRoCAS
Ro
RS
gmCAS
Vin
p
T
LO voltage
Cascode threshold voltage
Figure 5.60 Based on the nature of the LO signal magnitude and bias point, we may treat thetransconductance of the cascode as a square wave that switches on and off. This figure shows howthe gain of the switching mixer is gated by the LO voltage to create the switching effect of the mixer.This approach is used in double balanced mixers, such as Gilbert cells.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 353 #95
5.11 Analog MmWave Components 353
Open-loop gain
00
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
20 40 60 80 100 120 140 160 180
GHz
Phase
–60
–40
–20
00
20
40
60
80
100
20 40 60 100 120 140 160 180
GHz
80
Figure 5.63 The open-loop gain (top) and phase shift (bottom) of the oscillator in Fig. 5.62.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 355 #97
5.11 Analog MmWave Components 355
Open-loop gain magnitude
00
0.2
0.4
0.6
0.8
1
1.2
20 40 60 80 100 120 140 160 180
GHz
Open-loop phase
−1.50E+02
−1.00E+02
−5.00E+01
0
5.00E+01
1.00E+02
1.50E+02
0 20 80 100 120 140 160 180
GHz
6040
Figure 5.65 The gain and phase of a simple LC tank oscillator.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 359 #101
5.11 Analog MmWave Components 359
Open-loop gain magnitude
00
0.2
0.4
0.6
0.8
1
1.2
20 40 60 80 100 120 140 160 180
GHzOpen-loop phase
−1.50E+02
−1.00E+02
−5.00E+01
0.00E+00
5.00E+01
1.00E+02
1.50E+02
0 20 80 100 120 140 160 180
GHz
6040
Closed-loop gain
00
5
10
15
20
25
30
20 40 60 80 100 120 140 160 180GHz
Figure 5.71 For real-world oscillator circuits, the output spectrum will be polluted by power atfrequencies other than the intended operating frequency.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 360 #102
360 Chapter 5 MmWave RF and Analog Devices and Circuits
Noisy waveform
Time
Noiseless waveform
Noise impulse
Figure 5.72 Noise events, such as the noise impulse represented here, will affect both the amplitudeand phase of a circuit. In general, the amplitude impulse response of the circuit will act to removeamplitude noise over time. But phase noise persists, as is evident when we compare the phase of thenoisy waveform to the noiseless waveform.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 361 #103
5.11 Analog MmWave Components 361
Output spectrum Q comparison
00
5
10
15
20
25
30
20 40 60 80 100 120 140 160 180
GHz
L = 1 nH
L = 0.1 nH
Figure 5.73 The output spectrum of an LC oscillator becomes more spectrally pure as the qualityfactor of the tank increases.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 363 #105
1.2
0.8
0.6
0.4
0.2
00.00E+00 5.00E+10 1.00E+11
Hertz
Gai
nD
egre
esG
ain
Hertz
Hertz
Open-loop gain
1.50E+11 2.00E+11
1
150
100
50
−50
−100
00.00E+00 5.00E+10 1.00E+11 1.50E+11 2.00E+11
Open-loop phase
250
200
150
100
50
00.00E+00 5.00E+10 1.00E+11 1.50E+11 2.00E+11
Closed-loop gain
Figure 5.76 The transfer characteristics of a subharmonic oscillator.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 363 #105
1.2
0.8
0.6
0.4
0.2
00.00E+00 5.00E+10 1.00E+11
Hertz
Gai
nD
egre
esG
ain
Hertz
Hertz
Open-loop gain
1.50E+11 2.00E+11
1
150
100
50
−50
−100
00.00E+00 5.00E+10 1.00E+11 1.50E+11 2.00E+11
Open-loop phase
250
200
150
100
50
00.00E+00 5.00E+10 1.00E+11 1.50E+11 2.00E+11
Closed-loop gain
Figure 5.76 The transfer characteristics of a subharmonic oscillator.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 379 #121
5.12 Consumption Factor Theory 379
Energy expenditure per bit vs. transmission distance
Distance (m) [Log Scale]
PNP =1 W, max distance=67 m
PNP =10 W, max distance=120 m
PNP =100 W, max distance=213 m
PNP =1000 W, max distance=379 m
5 15 50 160 500
RX Gain 30 dbCarrier frequency 20 GHzPath loss exponent: 4
Capacity 1 GbpsHTX = 1HRX = 1
1580 5000
Ene
rgy
per
bit (
dBJ)
−30
−40
−50
−60
−70
−80
−90
−100
Figure 5.86 For a system with high signal path efficiency and high non-path power consumption,we see that the energy expenditure per bit is dominated by non-path power, indicating little advantageto shortening transmission distances [from [MR14b] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 380 #122
380 Chapter 5 MmWave RF and Analog Devices and Circuits
−1005 15 50 160 500 50001580
Ene
rgy
expe
nditu
re p
er b
it (d
BJ)
Distance (m) [log scale]
Energy per bit vs. transmission distance
−90
−80
−70
−60
−50
−40
−30
PNP =100 W, max distance=67 m
PNP =1000 W, max distance=120 m
PNP =1 W, max distance=21 m
PNP =10 W, max distance=38 m
RX gain 30 dBCarrier frequency 20 GHzPath loss exponent: 4
Capacity 1 GbpsHTX = 0.01HRX = 0.01
Figure 5.87 When signal path components are less efficient, as illustrated here, then shorter trans-mission distances start to become advantageous, as signal path power starts to represent a largerportion of the power expenditure per bit [from [MR14b] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 380 #122
380 Chapter 5 MmWave RF and Analog Devices and Circuits
−1005 15 50 160 500 50001580
Ene
rgy
expe
nditu
re p
er b
it (d
BJ)
Distance (m) [log scale]
Energy per bit vs. transmission distance
−90
−80
−70
−60
−50
−40
−30
PNP =100 W, max distance=40 mPNP =1000 W, max distance=71 m
PNP =1 W, max distance=13 mPNP =10 W, max distance=22 m
RX Gain 30 dBCarrier frequency 180Path loss exponent: 4
HTX =1Capacity 10 GbpsHRX =1
Figure 5.88 A higher frequency system that can provide a much higher bit rate capacity withoutsubstantially increasing non-path power consumption may result in a net reduction in the energyprice per bit [from [MR14b] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch05 2014/8/12 10:24 Page 381 #123
5.12 Consumption Factor Theory 381
−1005 15 50 160 500 50001580
Ene
rgy
expe
nditu
re p
er b
it (d
BJ)
Distance (m) [log scale]
Energy per bit vs. transmission distance
−90
−80
−70
−60
−50
−40
−30
PNP =100 W, max distance=13 mPNP =1000 W, max distance=22 m
PNP =1 W, max distance=4 mPNP =10 W, max distance=7 m
HTX = 0.01HRX = 0.01
Capacity 10 GbpsRX gain 30 dBCarrier frequency 180 GHzPath loss exponent: 4
Figure 5.89 Lower efficiencies of signal path components motivate the use of shorter transmissiondistances [from [MR14b] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch06 2014/8/12 10:24 Page 385 #3
6.2 Review of Sampling and Conversion for ADCs and DACs 385
Arbitrary signal1
0.8
0.6
0.4
0.2
-0.2
-0.4
-0.6
-0.8
-10 0.1 0.2 0.3 0.4 0.5
Time (seconds)0.6 0.7 0.8 0.9 1
´10-5
0
Vol
tage
(V
)
Figure 6.1 An arbitrary baseband analog signal having an approximate bandwidth of 100 MHz.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch06 2014/8/12 10:24 Page 386 #4
386 Chapter 6 Multi-Gbps Digital Baseband Circuits
Spectrum of arbitrary signal
-0.2-0.3-0.4
-90
-100
-80
-70
-60
-50
-40
-30
-20
-10
0
-0.1 0Frequency (GHz)
Am
plitu
de (
dB)
0.1 0.2 0.3 0.4
Figure 6.2 The spectrum of the arbitrary analog waveform shown in Fig. 6.1. The spectrum hasbeen normalized such that the strongest spectral component has an amplitude of 0 dB.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch06 2014/8/12 10:24 Page 387 #5
6.2 Review of Sampling and Conversion for ADCs and DACs 387
100 MHz BW signal sampled at 400 MHz
-0.2
-0.3
-0.1
0
4 4.02 4.04 4.06 4.08 4.1 4.12 4.14 4.16 4.18 4.2
Sampled signalOriginal signal
0.1
0.2
0.3
Time (seconds)
Vol
tage
(V
)
´ 10-6
Figure 6.3 A zoomed-in version of the arbitrary signal in Fig.6.1 showing how it has been sampled.The bandwidth (BW) of the original signal is 100 MHz, and the sampling rate is 400 MHz.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch06 2014/8/12 10:24 Page 388 #6
388 Chapter 6 Multi-Gbps Digital Baseband Circuits
Spectrum of 100 MHz bandwidth signal sampled at 400 MHz
-0.2-0.3-0.4-0.5
-90
-80
-70
-60
-50
-40
-30
-20
-10
-0.1 0Frequency (GHz)
Am
plitu
de (
dB)
0.1 0.2 0.3 0.4 0.5
Figure 6.4 The result of sampling the signal in Fig. 6.1 in the time domain is to make the spectrumperiodic in the frequency domain.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch06 2014/8/12 10:24 Page 389 #7
6.2 Review of Sampling and Conversion for ADCs and DACs 389
Aliased signal0
−10
−20
−30
−40
−50
−60−3 −2 −1 0
Frequency (GHz)
Am
plitu
de (
dB)
1 2 3
Figure 6.5 If the signal of Fig. 6.1 with a baseband bandwidth of 100 MHz is sampled at 100 MHz, the result is an aliased signal. In the frequency domain, overlapping copies of the original signal’s spectrum completely distort the resulting sampled signal.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch06 2014/8/12 10:24 Page 390 #8
390 Chapter 6 Multi-Gbps Digital Baseband Circuits
100 MHz BW signal, sampled at 400 MHz and discretized to 4 bits
-0.6
-20
-15
-10
-5
0
-0.4 -0.2 0Frequency (GHz)
Am
plitu
de (
dB)
0.2 0.4 0.6
Figure 6.6 Quantizing the signal of Fig. 6.1 with 4 bits reduces the dynamic range to 24 dB.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch06 2014/8/12 10:24 Page 392 #10
392 Chapter 6 Multi-Gbps Digital Baseband Circuits
Sampling with jitter
4.164.144.124.1
-0.2
-0.15
-0.1
-0.05
0
0.05
0.1
0.15
0.2
4.18 4.2 4.22 4.24 4.26
Time (seconds) ´ 10-6
Vol
tage
(V
)
Figure 6.7 An example of sampling with jitter, where uncertainty in the time interval betweensamples results in decreased dynamic range.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 438 #6
438 Chapter 7 MmWave Physical Layer Design and Algorithms
Positive signal limit
Voltage signalQuantization levelSample points
TIME
Negative signal limit
Vol
tage
Figure 7.2 Amplitude thresholding and quantization for a 3-bit ADC with uniform quantizationlevels in the receiver. In a wireless communications receiver, ADCs quantize both the in-phase andquadrature channels independently. Automatic gain control (AGC) is used to normalize the energyof the received complex baseband signal so that the ADC thresholds and quantization levels can befixed (i.e., do not depend on fading). As shown in Chapter 3, fading will be less pronounced withdirectional antennas.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 440 #8
440 Chapter 7 MmWave Physical Layer Design and Algorithms
0.5
0.4
0.3
0.2
0.1
0
14
12
10
8
6
4
2
0
Out
put (
volts
)
Out
put p
hase
shi
ft (
degr
ees)
Input (volts)0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4
MeasurementFitted model
Figure 7.3 AM-PM and AM-PM measurements and modified Rapp model for a CMOS 65 nmPA. For more detail on mmWave PA statistics, please consult the tables provided in Chapter 5 [plotcreated with data from [EMT+09] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 442 #10
442 Chapter 7 MmWave Physical Layer Design and Algorithms
–40 dBc/Hz
–50 dBc/Hz
–60 dBc/Hz
–70 dBc/Hz
–80 dBc/Hz
–90 dBc/Hz
–100 dBc/Hz
–110 dBc/Hz
Ideal component
IBM VCO — smoothed
Pole/zero model
2 MHz per division
67.278 GHz 67.296 GHz 67.314 GHz
Figure 7.4 Measured power spectral density of single sideband (SSB) VCO output for desired signalat 67.3 GHz. PSD measurements normalized to desired carrier output power (represented by dBc/Hz).A similar figure is shown in Chapter 5. Here, however, we have included a comparison to a pole/zeromodel that facilitates physical layer performance simulations [data taken from [FRP+05] and smoothedto create the plot c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 443 #11
7.2 Practical Transceivers 443
Pow
er s
pect
ral d
ensi
ty (
dB/H
z)
−30 dB/decade = flicker FM
−40 dB/decade = random walk
−20 dB/decade = white FM
−10 dB/decade = flicker phase
White noise
Frequency (Hz)
Figure 7.5 Different phase noise effects and their contribution to the power spectral density in theLeeson model [Lee66].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 449 #17
7.3 High-Throughput PHYs 449
SC-FDE, RS(255,239) — CM1.3
SC-FDE/OFDM — AWGNSC-FDE/OFDM — AWGN + mask
SNR (dB)
QPS
K B
ER
SC-FDE, LDPC(672,336) — CM1.3
SC-FDE, LDPC(672,432) — CM1.3OFDM, RS(255,239) —
OFDM, LDPC(672,432) — CM1.3
OFDM, LDPC(672,336)
100
10-1
10-2
10-3
10-4
10-5
-2 0 2 4 6 8 10 12
Figure 7.6 Bit error rate as a function of SNR for OFDM and SC-FDE in AWGN and LOSchannels with QPSK constellations. The coding options are RS(255,239) and LDPC(672,336). The802.15.3c spectral mask is added to the AWGN channel to demonstrate bandwidth conservation ofeach modulation strategy. Hardware impairments are not considered.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 450 #18
450 Chapter 7 MmWave Physical Layer Design and Algorithms
SNR (dB)
SC-FDE, RS(255,239) — CM1.3
SC-FDE/OFDM — AWGNSC-FDE/OFDM — AWGN + mask
SC-FDE, LDPC(672,336) — CM1.3
SC-FDE, LDPC(672,432) — CM1.3OFDM, RS(255,239) — CM1.3
OFDM, LDPC(672,432) — CM1.3
OFDM, LDPC(672,336) — CM1.3
4 6 8 10 12 14 16 18 20
16–Q
AM
BE
R
100
10-1
10-2
10-3
10-4
10-5
Figure 7.7 Bit error rate as a function of SNR for OFDM and SC-FDE in LOS CM1.3 channelswith 16-QAM constellations. Hardware impairments are not considered.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 451 #19
7.3 High-Throughput PHYs 451
SC-FDE, RS(255,239) — CM2.3
SC-FDE, LDPC(672,336) — CM2.3
SC-FDE, LDPC(672,432) — CM2.3OFDM,
OFDM, LDPC(672,432) — CM2.3
OFDM,
SNR (dB)
QPS
K B
ER
100
10-1
10-2
10-3
10-4
10-5
-2 0 2 4 6 8 10 12 14 16 18 20
Figure 7.8 Bit error rate as a function of SNR for OFDM and SC-FDE in NLOS CM2.3 channelswith QPSK constellations. No hardware impairments are considered.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 452 #20
452 Chapter 7 MmWave Physical Layer Design and Algorithms
SNR (dB)
16–Q
AM
BE
RSC-FDE, RS(255,239) — CM2.3
SC-FDE, LDPC(672,336) — CM2.3
SC-FDE, LDPC(672,432) — CM2.3OFDM, RS(255,239) — CM2.3
OFDM, LDPC(672,432) — CM2.3
OFDM, LDPC(672,336) — CM2.3
5 10 15 20 25
100
10-1
10-2
10-3
10-4
10-5
Figure 7.9 Bit error rate as a function of SNR for OFDM and SC-FDE in NLOS CM2.3 channelswith 16-QAM constellations. No hardware impairments are considered.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 453 #21
7.3 High-Throughput PHYs 453
SC-FDE, RS(255,239) —
w/ CMOS PA, 1 dB Backoffw/ CMOS PA, 2 dB Backoffw/ CMOS PA, 4 dB Backoff
— CM1.3
SC-FDE, LDPC(672,432)OFDM, RS(255,239) — CM1.3
OFDM, LDPC(672,432) — CM1.3
OFDM, LDPC(672,336) — CM1.3
SNR (dB)
Increasing
backoff
-2 0 2 4 6 8 10 12
QPS
K B
ER
100
10-1
10-2
10-3
10-4
10-5
Figure 7.10 Bit error rate as a function of SNR for OFDM and SC-FDE in LOS CM1.3 channelsand CMOS PA non-linearity with QPSK constellations.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 454 #22
454 Chapter 7 MmWave Physical Layer Design and Algorithms
SNR (dB)4 6 8 10 12 14 16 18 20 22
SC-FDE, RS(255,239) — CM1.3
w/ CMOS PA, 1 dB Backoffw/ CMOS PA, 2 dB Backoffw/ CMOS PA, 4 dB Backoff
SC-FDE, LDPC(672,336) — CM1.3
SC-FDE, LDPC(672,432) — CM1.3OFDM, RS(255,239) — CM1.3
OFDM, LDPC(672,432) — CM1.3
OFDM, LDPC(672,336) —
16–Q
AM
BE
R
100
10-1
10-2
10-3
10-4
10-5
Figure 7.11 Bit error rate as a function of SNR for OFDM and SC-FDE in LOS CM1.3 channelsand CMOS PA nonlinearity with 16-QAM constellations.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 455 #23
7.3 High-Throughput PHYs 455
SC-FDE,
w/ CMOS PA, 1 dB Backoffw/ CMOS PA, 2 dB Backoffw/ CMOS PA, 4 dB Backoff
LDPC(672,336) — CM2.3
SC-FDE,OFDM, RS(255,239) — CM2.3
OFDM, LDPC(672,432) — CM2.3
OFDM, LDPC(672,336) — CM2.3
QPS
K B
ER
0 5 10 15 20SNR (dB)
100
10-1
10-2
10-3
10-4
10-5
Figure 7.12 Bit error rate as a function of SNR for OFDM and SC-FDE in NLOS CM2.3 channelsand CMOS PA non-linearity with QPSK constellations.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 456 #24
456 Chapter 7 MmWave Physical Layer Design and Algorithms
SC-FDE, RS(255,239) —
w/ CMOSw/ CMOSw/ CMOS
— CM2.3
SC-FDE, LDPC(672,432)OFDM, RS(255,239) — CM2.3
OFDM, LDPC(672,432) — CM2.3
OFDM, LDPC(672,336) — CM2.3
16–Q
AM
BE
R
SNR (dB)
5 10 15 20 25 30 35
100
10-1
10-2
10-3
10-4
10-5
Figure 7.13 Bit error rate as a function of SNR for OFDM and SC-FDE in NLOS CM2.3 channelsand CMOS PA non-linearity with 16-QAM constellations.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 457 #25
7.3 High-Throughput PHYs 457
SC-FDE, RS(255,239)
w/ 5-bit ADC, 1.0σ peak mag.w/ 5-bit ADC, 1.5σ peak mag.w/ 5-bit ADC, 2.0σ peak mag.
SC-FDE, LDPC(672,336) — CM1.3
SC-FDE, LDPC(672,432) — CM1.3OFDM, RS(255,239) — CM1.3
OFDM, LDPC(672,432) — CM1.3
OFDM, LDPC(672,336) — CM1.3
SNR (dB)
Increasing
peak mag.
QPS
K B
ER
0 2 4 6 8 10 12
100
10-1
10-2
10-3
10-4
10-5
Figure 7.14 Bit error rate as a function of SNR for OFDM and SC-FDE in LOS CM1.3 channelsand 5-bit ADC samples with QPSK constellations.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 458 #26
458 Chapter 7 MmWave Physical Layer Design and Algorithms
SC-FDE, RS(255,239)
w/ 5-bit ADC, 1.0σ peak mag.w/ 5-bit ADC, 1.5σ peak mag.w/ 5-bit ADC, 2.0σ peak mag.
SC-FDE, LDPC(672,336) — CM1.3
SC-FDE, LDPC(672,432) — CM1.3OFDM, RS(255,239) — CM1.3
OFDM, LDPC(672,432) — CM1.3
OFDM, LDPC(672,336) — CM1.3
SNR (dB)
16–Q
AM
BE
R
6 8 10 12 14 16
100
10-1
10-2
10-3
10-4
10-5
Figure 7.15 Bit error rate as a function of SNR for OFDM and SC-FDE in LOS CM1.3 channelsand 5-bit ADC samples with 16-QAM constellations.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 460 #28
460 Chapter 7 MmWave Physical Layer Design and Algorithms
bN bN bN... ...
1st sequencecorrelationstarts here.
2nd sequencecorrelationstarts here.
Ns repetitions Ns repetitions
Pre-/post-fixes preventmultipath destruction ofzero sidelobe properties.
... ...
ann = N−NCP
N−1bn
n = N-NCP
N−1an
n = 0
NCP−1bn
n = 0
NCP−1aN aNaN
Figure 7.16 Illustration of how length-N complementary Golay sequence pair, aN and bN , maybe used to construct a training sequence that enables channel impulse response estimation throughcomplementary correlation. Pre- and post-fixes are added before and after Ns repetitions of eachcomplementary sequence to prevent excess multipath from disrupting zero-sidelobe properties. At thereceiver we correlate with the channel distorted versions of each N -length complementary sequenceand add them together to yield an estimate of a single tap. Delayed correlations are computed foreach tap, and each of the Ns repetitions is used to improve estimate robustness in the presence ofnoise.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 465 #33
7.5 Future PHY Considerations 465
Spec
tral
eff
icie
ncy
(bit/
sec/
Hz)
SNR (dB)0
0
0.5
1
1.5
2
2.5
3
3.5
5 10 15 20
Unquantized Capacity1-bit ADC Capacity2-bit ADC Capacity3-bit ADC Capacity
Figure 7.18 Capacity comparison with 1-, 2-, 3-, and, ∞-bit ADC precision for a discrete memory-less channel with perfect synchronization [SDM09].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 466 #34
466 Chapter 7 MmWave Physical Layer Design and Algorithms
(a) Analog equalization
analog signal
analog signal
AnalogEQ
ADC
ADC
DAC
Digitalreceiver
Digitalreceiver
Feedbackfilter
Symbol
decisions
Symbol
decisions
Energy-normalized
Energy-normalized
Tap delay line coefficients
(b) Mixed-signal equalization
Figure 7.19 Analog and mixed signal equalization architectures can reduce the ADC bit resolu-tion of the overall receiver (assuming that synchronization and other receiver functionality can bemaintained). Mixed signal equalization can be considered a DFE with an analog feedback filter anda digital feedforward filter.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch07 2014/8/12 10:25 Page 467 #35
7.5 Future PHY Considerations 467
Range reference (r)
Transmitter ULA Receiver ULA
8×8 Channel
Antenna element
θ φ
Figure 7.20 LOS MIMO channel with arbitrary uniform linear array (ULA) alignment and Nr =Nt = 8 elements on each ULA. The range reference of the link is denoted by r, and the total antennaarray lengths at the receiver and transmitter are Lr and Lt, respectively.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 485 #15
8.3 Beam Adaptation Protocols 485
−100 −80 −60 −40 −20 0 20 40 60 80 1000
2
4
6
8
10
12
14
16
18
Angle (degree)
Ant
enna
Pow
er G
ain
(dB
)
beams on codebook W3beams on codebook W2beams on codebook W1
Figure 8.7 A multilevel codebook proposed in [HKL+11] for wireless backhaul. Higher levels ofthe codebook have narrower beams, thus enhanced resolution [from [HKL+11, Figure 2] c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 488 #18
488 Chapter 8 Higher Layer Design Considerations for MmWave
1st time slot2nd time slot
Full duplex Half duplex Multi-hop
Figure 8.9 Different relay configurations with source, relay, and destination. In theory, a commu-nication link with a relay may exploit both the direct link from the source to the destination andthe indirect link through the relay. With a full duplex relay, the relay listens and retransmits at thesame time. A practical example of a full duplex relay is a repeater. With a half duplex relay, the relayeither transmits or receives and communication may be broken into two phases: transmission fromthe source, and transmission from the relay. In a multi-hop channel, the relay is half duplex and thesource to destination link is not exploited (from [Hea10]).
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 492 #22
492 Chapter 8 Higher Layer Design Considerations for MmWave
0 50 100 150 200 2500
0.360.6
11.2
2
2.5
3
4
5
Coverage range (m)
PHY
dat
a ra
te (
Gbp
s)
LOS-60GHz
LOS-5GHz
NLOS-60GHz
NLOS-5GHz
Figure 8.12 Coverage range and data rate at the physical layer for a 5 GHz link and 60 GHzlink under two different channel conditions: LOS and NLOS. It can be seen that the microwave linkprovides higher coverage at the expense of smaller data rates. A multi-band protocol could obtainthe rate benefits of 60 GHz and the coverage benefits of lower frequencies [from [YP08, Figure 1]c© IEEE].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 496 #26
496 Chapter 8 Higher Layer Design Considerations for MmWave
Data (pixels)
UEPMSB
MSB only No retransmission
scheduled
MSB in error LSB in error
Transmission
Acknow-
ledgment
Transmission
Acknow-ment
Retransmission
MSB only
LSB
MSB LSB
MSB LSB
MSB LSB
MSB LSBUEP
216 bytes 8 bytes RS (224, 216, t = 4)
ParityBad codeword
RS code 1 RS code 100RS code 2 RS code j
RS code 1 RS code 2
RS code 1 RS code 2
RS code 1
RS code 100
RS code 100
RS code 100RS code 2
RS code j
RS code j
RS code j
…
…
…
…
Video packet 1 (partition 1)
Video packet 2 (partition 1)
Video packet 3 (partition 1)
Video packet 4 (partition 1)
(e)
(b)(a)
(d)(c)
Stream 1
Partitionindex
MSB(video pixel)
LSB(video pixel)
Payload-N(video packet)
Payload-1(video packet)
Videoheader
MACheader
PHYheader
Partition 1packet
Pixeltype 1
Horizontal
1234...
1 2 3 4 5 . . .
Pixeltype 2
Pixeltype 3
Pixeltype 4
Partition 2packet
Partition 3packet
Partition 4packet
HCS
MSBCRC
LSBCRC
Length MCS
Payload
MCS0 (EEP) MCS1 or MCS 2
Length MCS
Payload
H and Vpositions
Videoframe #
Interlace field
Video control field of payload NIn
crea
sing
impo
rtan
ce
I channel
Q channel
Firsthalf
g1
g2
Secondhalf
Rate r1
Rate r2
MUX
Stream 2
StreamN–1
RaterN–1
Rate rN Stream N
…
Tim
eV
ertic
al
Type 1Type 2Type 3Type 4
Figure 8.13 Different strategies for supporting video in mmWave systems. (a) Pixel partition-ing. (b) Frame format. (c) Uncompressed video with automatic repeat request. (d) Unequal errorprotection. (e) Error concealment using Reed-Solomon codes. [From [SOK+08, Figure 13] c© IEEE]
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 496 #26
496 Chapter 8 Higher Layer Design Considerations for MmWave
Data (pixels)
UEPMSB
MSB only No retransmission
scheduled
MSB in error LSB in error
Transmission
Acknow-
ledgment
Transmission
Acknow-ment
Retransmission
MSB only
LSB
MSB LSB
MSB LSB
MSB LSB
MSB LSBUEP
216 bytes 8 bytes RS (224, 216, t = 4)
ParityBad codeword
RS code 1 RS code 100RS code 2 RS code j
RS code 1 RS code 2
RS code 1 RS code 2
RS code 1
RS code 100
RS code 100
RS code 100RS code 2
RS code j
RS code j
RS code j
…
…
…
…
Video packet 1 (partition 1)
Video packet 2 (partition 1)
Video packet 3 (partition 1)
Video packet 4 (partition 1)
(e)
(b)(a)
(d)(c)
Stream 1
Partitionindex
MSB(video pixel)
LSB(video pixel)
Payload-N(video packet)
Payload-1(video packet)
Videoheader
MACheader
PHYheader
Partition 1packet
Pixeltype 1
Horizontal
1234...
1 2 3 4 5 . . .
Pixeltype 2
Pixeltype 3
Pixeltype 4
Partition 2packet
Partition 3packet
Partition 4packet
HCS
MSBCRC
LSBCRC
Length MCS
Payload
MCS0 (EEP) MCS1 or MCS 2
Length MCS
Payload
H and Vpositions
Videoframe #
Interlace field
Video control field of payload N
Incr
easi
ng im
port
ance
I channel
Q channel
Firsthalf
g1
g2
Secondhalf
Rate r1
Rate r2
MUX
Stream 2
StreamN–1
RaterN–1
Rate rN Stream N
…
Tim
eV
ertic
al
Type 1Type 2Type 3Type 4
Figure 8.13 Different strategies for supporting video in mmWave systems. (a) Pixel partition-ing. (b) Frame format. (c) Uncompressed video with automatic repeat request. (d) Unequal errorprotection. (e) Error concealment using Reed-Solomon codes. [From [SOK+08, Figure 13] c© IEEE]
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 499 #29
8.6 Multiband Considerations 499
Backhaul connections between all base stations
Microwavelink
mmWavelink
Interference
Figure 8.14 A model for coexistence between mmWave and microwave cellular where themicrowave cellular network forms an umbrella network to facilitate the management of manymmWave communication links and to simplify functions like handoff. On the left side, a mobiledevice may connect either to a microwave or mmWave base station or to both simultaneously usingthe phantom cell concept. Interference on the microwave frequencies comes from other microwavebase stations and on the millimeter wave frequencies from other millimeter wave small cells. Asshown in Chapter 3 and elsewhere, the directionality of the beam patterns reduces the impact ofmmWave interferers [BAH14][SBM92][RRE14][RRC14][ALS+14].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 502 #32
502 Chapter 8 Higher Layer Design Considerations for MmWave
-5 0 5 10 15 200.4
0.5
0.6
0.7
0.8
0.9
1
SINR threshold in dB
SIN
R c
over
age
prob
abili
ty
Rc = 50 m
Rc = 100 m
Rc = 200 m
Rc = 300 m
(a) (b)
50 100 150 200 250 3000.4
0.5
0.6
0.7
0.8
0.9
1
Avg. cell radius Rc in meters
Cov
erag
e pr
obab
ility
wit
h SI
NR
> 10
dB
Figure 8.15 SINR coverage probability with different base station densities, where Rc =√
1/πλand λ is the density of base stations.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch08 2014/8/12 10:25 Page 504 #34
504 Chapter 8 Higher Layer Design Considerations for MmWave
Cel
l thr
ough
put
in M
Hz
3177.2
432
929
1.2 1.4124
848
0100200300400500600700800900
1000
Average5% tail
mmWave MUMassive MIMOSU-MIMSIS
Figure 8.16 Comparison of cell throughput of mmWave networks and microwave networks.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 512 #6
512 Chapter 9 MmWave Standardization
Freq
uenc
y al
loca
tions
(G
Hz)
65
60
55North America Europe Australia Korea Japan
Figure 9.1 International frequency allocation for 60 GHz wireless communication systems.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 517 #11
9.3 IEEE 802.15.3c 517
ParentPNC
ChildPNC
Parentpiconet
Neighborpiconet
Figure 9.7 Neighbor piconet. The parent piconet (controlled by the set-top box) manages coexis-tence with the neighbor piconet (controlled by the personal computer).
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 522 #16
522 Chapter 9 MmWave Standardization
Channel 1(center at 58.32 GHz)
Channel 2(center at 60.48 GHz)
Channel 3(center at 62.64 GHz)
Channel 4(center at 64.80 GHz)
edge at57.2406 GHz
edge at59.4006 GHz
edge at61.5600 GHz
edge at63.7200 GHz
edge at65.8800 GHz
Figure 9.8 Channelization in IEEE 802.15.3c provides four different channels for mmWave PHY.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 530 #24
530 Chapter 9 MmWave Standardization
... ...
......
~1,830 MHz
2,640 MHz
352 Data and pilot sub-carriers 141 Zeroed guard sub-carriers
3 Zeroed DC sub-carriers
... ...
16 Zeroed reserve sub-carriersSpectral mask
Figure 9.15 OFDM symbol formatting in the HSI PHY in IEEE 802.15.3c. The sub-carrier fre-quency spacing is 5.15625 MHz for all 512 sub-carriers. Three null DC tones prevent carrier feedthrough as well as ADC/DAC offset problems. The guard tones are usually nulled to meet spectralmask requirements, although customized guard tone values may optimize front end effects.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 532 #26
532 Chapter 9 MmWave Standardization
... ...
~1,760 MHz
2,538 MHz
... ...
3 Zeroed DC sub-carriersSpectral mask
141 Zeroed guard sub-carriers352 Data and pilot sub-carriers
Figure 9.16 OFDM symbol formatting for the HRP. The sub-carrier frequency spacing is ≈ 4.96MHz for all 512 sub-carriers. Three DC tones and all guard tones are nulled.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 533 #27
9.3 IEEE 802.15.3c 533
... ...
~ 92 MHz
317.25 MHz
34 Data and Pilot Subcarriers 91 Zeroed Guard Subcarriers
3 Zeroed DC Subcarriers
... ...
Spectral mask
98 MHz
Figure 9.18 OFDM symbol formatting for the LRP. The sub-carrier frequency spacing is 2.48 MHzfor all 128 sub-carriers. There are 37 data and null subcarriers, each with a subcarrier width of 2.48MHz, resulting in an occupied bandwidth of 91.76 MHz (∼ 92 MHz). The specified spectral maskpassband bandwidth (at 10 dB down) is 98 MHz, allowing for roll-off in the LRP mode. Three DCtones and all guard tones are nulled.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 540 #34
540 Chapter 9 MmWave Standardization
skewing
QI
1.25 dd
d
d
d
Figure 9.24 UEP Type 3 through skewing of 16-QAM. Here, the minimum distance betweenin-phase constellation points (where MSB bits are mapped) is increased by a factor 1.25.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 545 #39
9.3 IEEE 802.15.3c 545
Quasi-omnipatterns
Sectorpatterns
Hi-respatterns
Beampatterns
Figure 9.29 Four levels of patterns in antenna beamforming codebook for eight-element uniformlinear array (patterns visualized on the azimuthal plane (top view) for a vertical array orientation)[802.15.3-09].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 547 #41
9.3 IEEE 802.15.3c 547
Hi-res beam
Elevation
Azimuthal
Figure 9.30 High-resolution beam cluster in 3-dimensional space.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 549 #43
9.3 IEEE 802.15.3c 549
Q
I
StandardOOK
DAMIOOK
Q
I
Figure 9.32 OOK and DAMI constellations for optional use within the SC-PHY [802.15.3-09].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 552 #46
552 Chapter 9 MmWave Standardization
Host
A/Vcontroller
Adaptationsub-layer
MACsub-layer
PHYsub-layer
Privacy
Stat
ion
man
agem
ent e
ntity
Beamsteering
andtracking
A/Vformatting
Contentprotection
MAC
LRP HRP
Figure 9.33 Layering of WirelessHD device.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 553 #47
9.4 WirelessHD 553
Video/Audio
Source ProcessorSubsystem
Content ProtectionSubsystem
Audio/Video/Data Multiplexer
Data
HRP Data
Switch
Figure 9.34 WirelessHD packetizer diagram [Wir10].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 556 #50
556 Chapter 9 MmWave Standardization
Type CPLCP
Type BPLCP
Type APLCP
Type CPHY
Type BPHY
Type APHY
Arrayconfig.
Type AMAC
Type BMAC
Type CMAC
... Control
Ant
enna
arra
yPH
YM
AC MAC layer
managemententity (MLME)
PHY layermanagement
entity (PLME)
Devicemanagemententity (DME)
Protocol adaptation layer (PAL)
Figure 9.35 Protocol structure of ECMA-387 [ECMA08].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 558 #52
558 Chapter 9 MmWave Standardization
... Control
60 GHz PHYOOB PHY
60 GHz MACOOB MAC
Convergence MAC
Higher layers
Figure 9.36 Out-out-band (OOB) control channel layered architecture in ECMA-387 [ECMA08].
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 570 #64
570 Chapter 9 MmWave Standardization
Link change interval
DATA(to STA)
DATA(to STA)
DATA(to STA)
ACK(to STA)
ACK(to Relay)
ACK(to STA)
Never received bydestination STA
No ACK Relay RX focused on source,TX focused on destination
STA(dest)
STA(source)
Relay
DATA(to Relay)
Link change interval Link change interval
Figure 9.46 Transmissions by source STA, amplify-and-forward relay, and destination STA in alink switching example in normal mode.
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
PTG-Rappaport Rappaport Ch09 2014/8/12 10:26 Page 580 #74
580 Chapter 9 MmWave Standardization
...STP
... ...
0 1
167
168
169
335
Pair 167Pair 1Pair 0
Figure 9.53 Static tone pairing (STP) in MCS 13-17. Even and odd sub-carriers are paired (outof 336 sub-carriers total) and mapped to maximize the minimum sub-carrier distance between evenand odd sub-carriers (168 sub-carrier distance).
From Rappaport et al., Millimeter Wave Wireless Communications, ISBN-13: 978-0-13-217228-8. Copyright © 2015 Pearson Education, Inc.
Group 0DTP
0 1
164
1672 3
165
166
G41
+ 1
68G
41 +
169
G41
+ 1
70G
41 +
171
G0 +
168
G0 +
169
G0 +
170
G0 +
171
...
............
Group 41
Figure 9.54 Dynamic tone pairing (DTP) in MCS 13-17. For DTP, the group pair index (GPI) isgiven to the PHY and is defined by the transmitter. The mapping GPI is hence a permutation whereGPI : 0, 1, . . . , 41 → 0, 1, . . . , 41 and GPI : k → Gk. Note that although the even elements ofeach group have a fixed mapping, the odd elements may be mapped more generally. In other words,Gk may vary for a fixed k, depending on the link configuration. The ends of the DTP transformedsub-carriers are not shown to maintain generality, although one of the 42 groups must be mappedto the last DTP group in practice.