Design World / EE Network Power Electronics Handbook 2016

2016 Designing with permanent magnets Page 14 Teardown: What’s inside an electric toothbrush Page 08

H A N D B O O K

POWER ELECTRONICS

160101_DT16_EEW_US_Snipe.indd 1 12/18/15 2:28 PM

EE_FEB 2016_Cover_Vs1.indd 1 2/24/16 5:11 PM

160101_INNOV_EEW_Handbook_US.indd 1 12/18/15 3:04 PMDigi Key 2-16 EE.indd 1 2/23/16 12:35 PM

Open VPX is a trademark of VITA.

VPXtra is a trademark of Behlman.

Not only do the Behlman VPXtra™ 1000CD-IQ and 1000CM-IQ Power Supplies deliver the highest VPX power available today, they also have high-level intelligence functions not available in any other VITA 62 compliant power supplies.

This lets designers create higher power systems having more intelligent communication, measurement and control functions that can monitor and report the status of multiple parameters, and support ANSI/VITA 46 signals.

Whether you want to design and build better VPX systems, or simply operate them, the extra performance and reliability achieved by using Behlman VPXtra™ 1000CD-IQ and 1000CM-IQ Power Supplies can increase your success.

With these 6U Behlman VPXtra™ Power Supplies, users can monitor and report output voltage, output current, input voltage, input current, and temperature. ANSI/VITA 46 signal support is provided for geographical addressing, NVMRO (Non-Volatile Memory Read Only) and SYSRESET input.

Both provide user-adjustable warning/fault levels for voltage, current or temperature; inventory management information (part number, serial number, and revision status); SMBALERT # signal power supply fault warning; over 200K storage memory; extensive PMBUS command set and status registers support.

VPXtra™ 1000CD-IQ: 1000 WDC via 2 outputs VPXtra™ 1000CM-IQ: 700 WDC via 5 outputs

The highest power of all, plus intelligent communication, measurement and control.

PROUDLY MADE

IN THE USA

www.behlman.com • 1 631 435-0410 • [email protected]

A POWER SUPPLY

CAN DO A LOT MORETHAN SUPPLY POWER.

CONTRARY TO WHAT YOU THOUGHT,

VPXtra-IQ_9x10.875_REV01_FINAL.indd 1 11/2/15 4:17 PMBehlman_EE_ad.indd 1 2/23/16 12:34 PM

2 2016 powerelectronictips.com | designworldonline.comDESIGN WORLD — EE Network

Pow

er E

lect

roni

cs

Anotherdifficultywithdependingtooheavilyonamathematical approach is that it tends to foster a fear factor when it comes to designing real magnetics. “It all seems to start inthefirstcircuitstheorycourse,”Ridleysaid.“Whenyoudrawaschematic,youputinlinesshowingwherecurrentsgoandwherevoltagesappear.Butwhenyouputamagneticfieldinthere,the lines go all over the place. Youcan’tcapturethatinyourschematic. It is inconvenient from theteachingpointofview.” Making the problem worse isthatmostuniversitycoursesoncircuitsspendlittletimediscussingmagneticcomponents.“Irecentlylookedatthenotesforamajoruniversity’scircuittheorycourse.Theyspent20minutesontheworkingsofcapacitors,fiveoninductors,andzerotimeontransformers,”saidRidley.“Sothereisamassiveholeintermsofwhatuniversitiescouldbeteaching.Perhaps5%of engineers will learn these thingsbyosmosisinindustry.Butitappearsasthoughfewuniversitiesareprovidingagoodpracticaleducationonabouthowto approach magnetic design problems.” ThislackofeducationalresourcesisonereasonRidleyisontheagendafortheupcomingAPECAppliedPowerElectronicsConferenceinLongBeach.He’steachingashortcourseonthedifficultiesofmodelingtransformers.He’salsochairingaplenarysessionandanindustrysession on how to grapple with magnetic core losses at the higher switchingfrequenciesatwhichstate-of-the-artsuppliesnowoperate. Judgingbythecomplexitiesofmagneticdesign,hissessionswillprobablybewellattended.

Youmightthinksojudgingbythe“blackart”labelthat’softenappliedtothisaspectofpowerelectronics.Thereputationforwitchcraftherecomesinlargepartfromtherolemagneticmaterialsplayingettingsuchcircuitstoworkproperly.Theproblem:Seemingly,fewpeopleunderstandhowmagneticcoresreallywork,letalonehowtodesigncircuitsthatusethemeffectively. “Themisconceptionsandmisunderstandingsaboutmagneticsareimmensebothinuniversitiesandinindustry,”saidDr.RayRidley,anexpertinmagneticdesignpractices.Hesaidoneproblemwithcircuitsinvolvingmagneticsisthattheydon’tlendthemselvestoneatanalyticalsolutions.“Theydon’tfitinamodernspreadsheetsimulationorCADsolutionbox.Peoplewantsomethingthatiscannedthattheycanpickupwithoutdoinganythingextra.Magneticcircuitsaren’tlikethat.” Onereasonmagneticsdon’tlendthemselvestostraightforwardanalysisisthattherearenostandardsforexpressingthepropertiesofmagneticmaterials.“Twodifferentvendorsofsimilarmaterialswillpresentdataintwodifferentways.Youcan’tjustputthematerialpropertiesonaspreadsheetandthenfigurehowthecorelossesvarywithfrequency,temperatureanddutycycle.Thereisnosinglewayofdoingthatrightnow,”saidRidley. Thislackofstandardsthwartsteachingmethodsinuniversitiesthattendtogravitatetowardneat,codifiedsolutions.“Academicsliketogosolelywithequations.Theymakethemistakeoftryingtofindonethatworksforeverybody.Theyarechasingsomethingthatcan’tbedone,”Ridleysaid.“Inindustry,peoplewhoactuallydomagneticsdesignwelldon’tapproachitwithsuchcomplicatedmethods,butwithaheavydoseofpracticality.”

Fear of

Leland TeschlerExecutive Editor@DW_LeeTeschler

MAGNETICSDOES IT REALLY TAKE A VOODOO PRACTITIONER TO DESIGN A SWITCHING POWER SUPPLY?

Editorial_EE_Feb 2016_Vs2.MD.indd 2 2/23/16 1:23 PM

MPDm e m o ry p r o t e c t i o n d e v i c e s

M P D i s a g l o b a l m a n u f a c t u r e r o f b a t t e r y h o l d e r s a n d o t h e r e l e c t r o n i c c o m p o n e n t s . W e b e l i e v e t h a t o u r c o m p o n e n t s s h o u l d f i t e a s i l y i n t o y o u r d e s i g n s , w h i c h i s w h y w e a r e a l w a y s c r e a t i n g i n n o v a t i v e n e w p r o d u c t s .

b a t t e r y h o l d e r s . c o m

ConstantlyKeeping you

connectedm p d d ev ic e s ca n b e f o u n d i n

S E R V E R E Q U I P M E N T A R O U N D T H E WO R L D

Memory Protection Devices 2-16 EE.indd 3 2/23/16 12:36 PM

02 Fear of Magnetics Does it really take a voodoo practitioner to design a switching power supply?

08 Teardown: What’s inside a Phillips Sonicare electric toothbrush

A novel ac rectification and coil-driving scheme may characterize the electronics found in this widely used oral hygiene instrument.

14 Basics of designing with permanent magnets

It can be tricky to correctly specify magnetic materials for power circuits and actuators. An understanding of definitions and common properties serves as a good starting point.

20 GaN components boost power density in supplies

To understand why gallium-nitride components shrink the size of power supplies, examine how they dissipate energy at high frequencies.

CONTENTS

8 58

4 DESIGN WORLD 2016 www.powerelectronictips.com

4826 Test instruments help

tame power factor and signal integrity Harmonic analysis helps design electronics that will behave well on the power grid.

30 A close look at the Level VI power supply spec DOE power supply efficiency specifications target a broader array of power supplies to save energy and reduce greenhouse-gas emissions.

36 Multiple charging modes from a single amplifier

Though there are two standards for charging appliances wirelessly, a single circuit can be devised to serve as a charging node for both of them.

42 Compensating for temperature in

high-frequency rectifier diodes Over-temperature can be a problem in high-performance switching supplies. New power rectifier diodes build in safeguards to head off heat build-up.

48 How and when MOSFETs blow up

High temperatures and operating conditions outside the safe operating area can sabotage MOSFETs used in switching circuits.

52 Measuring wireless charging efficiency in the real world When it comes to energy efficiency, there are significant differences in competing wireless charging methods.

58 Digital inrush current controller reference design A reference design shows how to control current inrush in

ac-to-dc rectifiers.

Contents_EE_Feb 2016_Vs3.MD.indd 4 2/24/16 8:18 AM

Keysight 2000 X-Series

Keysight 3000T X-Series



Bandwidth 70 MHz – 200 MHz 100 MHz – 1 GHz 200 MHz – 1.5 GHz 1 GHz – 6 GHz

Sample Rate 2 GSa /s 5 GSa /s 5 GSa /s 20 GSa /s

Waveform Update Rate >100,000 wfms/s >1,000,000 wfms/s >1,000,000 wfms/s >450,000 wfms/s

Display 8.5″ 8.5″ touch 12.1″ touch 12.1″ multi-touch

Zone Touch Triggering N/A YES YES YES

Instrument Integration Arbitrary waveform generator, digital voltmeter, protocol analyzer, counter, MSO

Value and performance at your ngertips.Solve your toughest measurement challenges with the Keysight In niiVision X-Series family of oscilloscopes. Scopes comestandard with high-end features such as large touch displaysand industry-leading waveform update rates. Plus, Keysightoffers a wealth of application-speci c software forthese fully upgradable scopes. With Keysight,not only can you do more, you get more.

© Keysight Technologies, Inc. 2015

US A: 800 829 4444 CAN: 877 894 4414

Knowledge at your ngertips: get app notes, videos, webcasts and more:www.keysight.com/find/infiniivision-knowledge

Keysight Technologies 2-16 EE.indd 5 2/23/16 12:37 PM

EDITORIAL

Editorial DirectorPaul J. [email protected]@dw_editor

Managing Editor Leslie Langnau [email protected]@dw_3Dprinting

Executive EditorLeland [email protected]@dw_LeeTeschler

Senior EditorMiles [email protected]@dw_Motion

Senior EditorMary [email protected]@dw_marygannon

Senior EditorLisa [email protected]@dw_LisaEitel

Associate EditorMike [email protected]@dw_mikesantora

Assistant Editor Michelle [email protected]@wtwh_Michelle

Contributing Editor Aimee [email protected]

Contributing Editor Danielle [email protected]

NEW MEDIA/WEB/

BUSINESS DEVELOPMENT

Web Development ManagerB. David [email protected]@wtwh_webdave

Web Development SpecialistPatrick [email protected]@amigo_patrick

Digital Marketing SpecialistAndrew [email protected]

VideographerJohn [email protected]@wtwh_jhansel

VideographerKyle [email protected]@wtwh_kyle

Videographer Alex [email protected]

Digital Media ManagerPatrick [email protected]@wtwhseopatrick

Online CoordinatorJennifer [email protected]@wtwh_jennifer

Director, Audience DevelopmentBruce [email protected]

ControllerBrian [email protected]

Traffic ManagerMary [email protected]

Production AssociateTracy [email protected]

GRAPHICS

VP, Creative ServicesMark [email protected]@wtwh_graphics

Art Director Matthew [email protected]@wtwh_designer

Graphic DesignerAllison [email protected]

2011 - 20152014 Winner

SALES

Mike [email protected]

Todd [email protected]@wtwh_todd

Jessica [email protected]@wtwh_MsMedia

Michael [email protected]@mrference

Mike [email protected]

David [email protected]@wtwh_david

Neel [email protected]@wtwh_ngleason

Tom [email protected]@wtwh_Tom

Jim [email protected]@jpowers_media

Courtney [email protected] @wtwh_CSeel

LEADERSHIP TEAM

Publisher Mike [email protected]@wtwh_memich

Managing DirectorScott [email protected]@SMMcCafferty

EVPMarshall [email protected]@mmatheson

Follow the whole team on twitter @DesignWorld

CONNECT WITH US!

WTWH Media, LLC

6555 Carnegie Avenue, Suite 300

Cleveland, OH 44103

Ph: 888.543.2447

Fax: 888.543.2447

DESIGN WORLD does not pass judgment on subjects of controversy nor enter into dispute with or between any individuals or organizations. DESIGN WORLD is also an independent forum for the expression of opinions relevant to industry issues. Letters to the editor and by-lined articles express the views of the author and not necessarily of the publisher or the publication. Every effort is made to provide accurate informa-tion; however, publisher assumes no responsibility for accuracy of submitted advertising and editorial information. Non-commissioned articles and news releases cannot be acknowledged. Unsolicited materials cannot be returned nor will this organization assume responsibility for their care.

DESIGN WORLD does not endorse any products, programs or services of advertisers or editorial contributors. Copyright© 2016 by WTWH Media, LLC. No part of this publication may be reproduced in any form or by any means, electronic or mechanical, or by recording, or by any information storage or retrieval system, without written permission from the publisher. Subscription Rates: Free and controlled circulation to qualified subscribers. Non-qualified persons may subscribe at the following rates: U.S. and possessions: 1 year: $125; 2 years: $200; 3 years: $275; Canadian and foreign, 1 year: $195; only US funds are accepted. Single copies $15 each. Subscriptions are prepaid, and check or money orders only.

Subscriber Services: To order a subscription or change your address, please email: [email protected], or visit our web site at www.designworldonline.com

POSTMASTER: Send address changes to: Design World, 6555 Carnegie Ave., Suite 300, Cleveland, OH 44103


Pow

er E

lect

roni

cs

MARKETING

Marketing ManagerStacy [email protected]@wtwh_Stacy

Marketing and Events CoordinatorJen [email protected]@wtwh_jen

Marketing CoordinatorLexi [email protected]@medtech_lexi

Digital Marketing SpecialistJosh [email protected]@wtwh_joshb

Medical Design& OUTSOURCING

Staff_EE_FEB 2016_Vs1.indd 6 2/24/16 1:32 PM

Mean Well 2-16.indd 7 2/23/16 12:39 PM


Pow

er E

lect

roni

cs

Teardown: What’s inside a Phillips Sonicare electric toothbrush

T he Philips Sonicare Elite series toothbrush is billed as generating over 31,000 brush strokes per minute thanks to “sonic motion” driving its

brush. A teardown reveals what Philips is talking about in its marketing material: The brush head drive involves torsion bars positioned electromagnetically rather than by an electric motor. Cut through the plastic handle of the Philips device and you’ll see a plastic frame holding a battery—a nickel-cadmium 1.2-V unit from Sanyo—the pickup coil in the base, which is used for inductive charging, and a massive electromagnet on an E-shaped frame. This electromagnet is used to move the brush head back and forth to get the brushing movement. The circuit board that contains the electronics solders onto this frame with seven pins making connections for the pickup coil, the battery and the brush-head drive coil. Like most electric toothbrushes these days, the Philips Sonicare Elite employs wireless charging while it sits in a charging unit that doubles as a holder for the toothbrush. When we park the toothbrush in the holder and put a scope on the inductive pickup coil, we get a waveform that turns out to be at a frequency of about 100 kHz. So the base unit is generating a 100-kHz signal and the brush unit picks it up. The base unit evidently contains a frequency converter that ups the mains frequency from 60 Hz to 100 kHz. Unfortunately, we couldn’t see the details of this circuitry because the base unit electronics are potted in some kind of epoxy.

On the toothbrush itself, the charging circuit is interesting in that there is extremely little to it. One can find teardowns of other brands of electric toothbrushes on the web that reveal what might be called conventional rectifier components: rectifier diodes and a capacitor or two for filtering, used to rectify the ac signal picked up by the charge coil into a dc signal that then goes to the rest of the components on the board. But that’s not at all what goes on in the Philips Sonicare brush. Following the tiny traces on the circuit board in the handle shows that the pickup coil connects directly to a 24-pin chip. The chip doesn’t carry markings for any kind of commercially available device, so its functions can only be discerned by examining its connections. Chip markings indicate the IC comes from ams, or Austriamicrosystems. There are no intervening components for rectification (or for anything else) between the wireless charging coil and this 24-pin IC, so the only conclusion can be that the chip does the rectifying. But that’s not all it does. An examination of its connections shows it probably also contains an 8-bit processor. ams doesn’t make a processor chip, let alone one that contains some kind of ac rectifier. So we must conclude this is an ASIC of some kind that ams did for Philips. Teardowns of other electric toothbrush brands have found 4-bit, rather than 8-bit, processors handling the brush controls. But we counted what seem to be

Pow

er E

lect

roni

cs

LELAND TESCHLERExecutive Editor

A novel ac rectification and coil-driving scheme may characterize the

electronics found in this widely used oral hygiene instrument.

Teardown_EE_2016_Power_Vs3.MD.indd 8 2/23/16 1:29 PM

Keep a Hand on Precise Frequency Control

www.ctscorp.com

PMS 2925

PMS 541-theirs

PMS 313

CTS puts the frequency-control solutions you need in the palm of your hand!

CTS places the frequency-control products you need in the palm of your hand. Serving some of the most challenging military/defense and industrial applications with compelling SWaP-C benefits, CTS offers a wide range of high-performance frequency control solutions for demanding ultra-low-power, low-phase-noise, and low-g-sensitivity requirements in extremely compact configurations. In addition to CTS miniature low-power OCXOs, Frequency Control products include flexible and configurable RF modules to 2.5 GHz with industry-best jitter and phase-noise performance as well as Hi-Rel COTS clock oscillators with stable frequencies to 800 MHz, full military temperature range, and low-jitter options. CTS offers custom solutions and quick-turnaround prototypes to meet the most demanding frequency-control requirements:

• Low-power, miniature OCXOs, 10 to 250 MHz with 10-ppb stability, -40 to +85°C

• Low-power, miniature OCXOs with low-g sensitivity to 2E-10/g Gamma

• Ruggedized TCXO modules to beyond 1 GHz

• Ruggedized RF modules to beyond 2.5 GHz

• Hi-Rel COTS crystal oscillators (XOs) from 16 kHz to 800 MHz, -55 to +125°C

CTS_EE_ad.indd 9 2/23/16 12:40 PM

10 2 • 2016 powerelectronictips.com | designworldonline.comDESIGN WORLD — EE Network

Pow

er E

lect

roni

cs

looks as though a couple of these capacitors might be part of the clock oscillator. But the other three don’t seem to have any function other than perhaps to work in the rectifier circuit somehow. Also, there are no inductors on the circuit board. One ac-to-dc rectification scheme that uses capacitors only, no inductors, is a charge pump. It basically employs a network of diodes and charging or discharging capacitors to change an ac signal to a dc voltage. While the ac signal is positive, current flows through diodes and charges capacitors. When the ac signal goes negative, the diodes don’t conduct and the capacitors discharge. Multiple stages can be stacked to get better rectification. The components found on the board are consistent with this


Pow

er E

lect

roni

cs

10 I/O lines on the ams chip that would seem to indicate the use of at least an 8-bit controller: The board contains two momentary contact switches. One is a main turn-on switch, the other changes the speed of the brush if the operator desires. The ams processor chip has an input from both of them. Six pin connections go to six LEDs on the board that show the state of battery charging. Two more of the IC’s pins go to another chip that basically drives the brush back and forth, for a total of 10 outputs. Our guess: It is easier to manage 10 I/O connections with an 8-bit processor rather than a 4-bit device. One thing is for sure: Regardless of its processing abilities, the ams chip is a clocked device. The presence of a crystal oscillator (again, with non-standard markings) on the backside of the board is a dead giveaway that we have a clocked circuit here.

POWER CONVERSIONThe lack of markings on the ams chip makes the toothbrush charging scheme somewhat of a mystery. But there is a clue on the board in the form of five capacitors. It

The Sonicare PCB attached to the plastic frame with seven solder connections: two for the wireless charging coil, three for the electromagnet coils on the E frame, and two for the nickel-cadmium battery.

The crystal clock oscillator was the sole component residing on the back of the PCB.

approach. So it is plausible that this toothbrush uses some kind of a simple charge pump to get dc that charges up the battery. Also, a look at the specs for the Sanyo battery reveals it slow-charges at 150 mA in about 16 hours. The Sonicare manual says the brush takes 24 hours to fully charge, so the charge current may be below 150 mA. That seems to be well within the level that an ASIC this size should be able to handle. Perhaps the most interesting part of the circuit comes into play when the brush moves back and forth. With a push of the on/off switch, the processor turns on a circuit that generates an oscillating electric field. This happens using another chip, a Toshiba dual n-channel MOSFET. An output from the ams chip goes to the gate of

Teardown_EE_2016_Power_Vs3.MD.indd 10 2/23/16 1:30 PM

The UltimatePower Couple

With their high K and small size, these 1:1 coupled inductorsare the perfect match for your SEPIC and fl yback applications

WWW.COILCRAFT.COM

®

High Current MSD Series

Offered in eleven body sizes and hundreds of inductance/current rating combinations, our MSD/LPD families are perfectly coupled to all your SEPIC and fl yback designs.

The MSD Series offers current ratings up to 16.36 Amps, low DCR, coupling coeffi cients as high as K ≥ 0.98, and up to 500 Vrms winding-to-winding isolation.

With profi les as low as 0.9 mm and footprints as small as 3.0 mm square, the LPD Series offers current ratings up to 5.6 Amps, DCR as low as 0.042 Ohms and coupling coeffi cients as high as K ≥ 0.99.

You can see all our coupled inductors, including models with turns ratios up to 1:100, at www.coilcraft.com/coupled.

Low Profi le LPD Series

Coilcraft_EE_ad.indd 11 2/23/16 12:41 PM


Pow

er E

lect

roni

cs

each MOSFET. Each MOSFET output ties to one end of a coil driving the brush. There is a center tap on the coil that connects directly to the positive terminal of the battery. So in operation, the processor drives one side of the coil, then the other, to generate an alternating magnetic field that vibrates the brush back and forth. We were unable to measure the frequency of that oscillating field, but one of the original patents for electric toothbrushes available online said it is in the range of 250 Hz. The brush oscillation circuit contains two diodes. (We initially thought these had something to do with rectifying the 100-kHz charging waveform until we mapped out

all the PCB traces.) These diodes are there to bleed off the inductive kick that arises when suddenly switching on big coils. The same kind of diodes can be found to reduce inductive kicks on circuits that switch electromechanical relays. Without the diodes, there’s a risk of destroying the MOSFETs. Completing the connection to the MOSFETs are two resistors serving as pull-down resistors for the MOSFET gates. The oscillating field that appears at the end of the E-shaped frame interacts with the bottom of the brush head. And on the bottom of the brush head are two magnets, and they appear to be pretty strong magnets at that. We have no way of quantifying their magnetic fields, but they act as though they might be of the rare earth variety. This

Based on the components found on the PCB, one possibility is that the ac rectification scheme on the Sonicare toothbrush might be something along the lines of a simple charge pump, as depicted here.

The base unit for the toothbrush had its electronics potted in the epoxy visible here; so there wasn’t much to see or analyze.

The schematic for the connections we found on the PCB for the ams chip look like this; quite simple indeed.

The drive circuit for the brush head centers on a Toshiba n-channel MOSFET chip. It includes two kickback diodes and two pull-up resistors.

Teardown_EE_2016_Power_Vs4.MD.indd 12 2/24/16 11:20 AM

132 • 2016 powerelectronictips.com | designworldonline.com DESIGN WORLD — EE Network

TOOTHBRUSH TEARDOWN

132016 powerelectronictips.com | designworldonline.com DESIGN WORLD — EE Network

might also help explain why replacement brush heads for the Sonicare Elite series go for about $8 a piece or more as of this writing. Asthefieldchanges,itbasicallypushes the magnets back and forth. The magnets in turn connect to the brush through three torsion bars which transmit the magnet motion to the brush itself. And after a couple minutes of this 250-Hzoscillatingaction,thebrushuserhas clean teeth.

REFERENCES

Austriamicrosystems (ams) ams.com/eng

Acoustic toothbrush patent: A lot of the circuit details on the Philips toothbrush resemble those described in this patent:google.com/patents/US20020092104.

Philips USA usa.philips.com/c-m/consumer-products

ProminentonthePCBfortheSonicarebrusharethetwoICsfromamsandToshiba,fivecapacitorsthatseemtobepartoftheclockoscillatorandrectifier,twomomentaryswitches,twokick-backdiodesandpull-upresistors,andthesixLEDsservingaschargeindicators.Theclockoscillatormountsontheback.

Twostrongmagnetssitontheendofthreetorsionbarsinthereplaceablebrushhead.ThemagnetsarepositionedontheendoftheEframeandareoscillatedbackandforthbytheelectricfield.

Torsion bars

Brushhead magnets

Teardown_EE_2016_Power_Vs4.MD.indd 13 2/24/16 11:31 AM


Pow

er E

lect

roni

cs

Basics of designing with permanent magnets

T hough magnetic materials are widely used in power components, engineers frequently have misunderstandings about their magnetic

properties. For example, the energy product (BH)max is the most frequently used permanent magnet property, but probably the least understood. And most engineers do not learn much about permanent magnets in school. Likely they will learn a few bits of jargon, like units and parameters, but will never really master the subtle points. So it can be overwhelming the first time an engineer becomes involved in a project involving a permanent magnet.

People designing with magnets frequently get surprised by several subtle points. These include magnet temperature characteristics, testing and the interactive nature of magnetic design; any single change has consequences elsewhere. It usually takes most engineers several tries before they feel comfortable designing with magnets. And it is quite common for designers to start the process by looking back at the last design for guidance, rather than starting from scratch.

Some magnetic materials are more difficult to design with than others. Alnico magnets, for example, are an

DESIGN WORLD — EE Network

Pow

er E

lect

roni

cs

STAN TROUTOwner/Metallurgist

Spontaneous Materials

It can be tricky to correctly specify magnetic materials for power circuits

and actuators. An understanding of definitions and common properties

serves as a good starting point.

Commonly used units for permanent magnetsQuantity Symbol CGS Unit SI Unit ConversionsMagnetic Induction B Gauss (G) Tesla (T) 1 T = 104 G = 10 kG

1 G = 10-4 T

Magnetic Field H Oersted (Oe) Ampere/meter (A/m)

1 A/m = 4π × 10-3 Oe1 Oe = 79.58 A/m

Magnetization 4πM (CGS)M (SI)µ0M (SI)

Gauss (G) Ampere/meter (A/m) Tesla (T)

1 G = 79.58 A/m1 A/m = 4π × 10-3 G1 T = 104 G = 10 kG1 G = 10-4 T

Energy Product (BH)max Gauss-Oersted (G-Oe)

Joule/meter3 (J/m3)

1 J/m3 = 40π G ∙ Oe = 125.7 G ∙ Oe1 G ∙ Oe = 7.958 × 10-3 J/m3

The three vectors describing magnet performance are related to each other and the relationships take different forms in different unit systems.

Magnetics_EE_2016_Power_Vs5.MD.indd 14 2/23/16 1:33 PM

It’s what’s on the InsIde that counts®

E L E C T R O N I C S C O R P.

www.keyelco.com • (718) 956-8900 • (800) 221-5510

aNTI -v IbRaTION gROmmETS

• Designed for noise suppression in office or lab equipment • Isolates sensitive instrumentation from unpredictable vibration and shock • Provides the flex needed to dampen unwanted noise and vibration • Shoulder screws

specially designed to provide correct spacing • Request Catalog M65

™

EE-EH THiNK SiloSun+AntiVibGrom_2-16.indd 1 1/8/16 12:13 PMKeystone 2-16 EE.indd 15 2/23/16 12:42 PM


Pow

er E

lect

roni

cs

H, Magnetic field. A vector describing the intensity of the magnetic field created by a field source, such as current moving through a wire or a permanent magnet. Correct units for reporting this quantity are Oersted in CGS and Ampere-turn/meter (A/m) in SI.

M, Magnetization. A quantity describing the magnetic state of the material, representing the vector sum of individual atomic magnetic moments per unit volume. The magnetic moments arise from unpaired spinning electrons, typically located in the 3d or 4f electron shell of each atom. The CGS unit for M is Gauss and the SI unit for M is A/m.

These three vectors are not independent, they are related. Fundamentally, induction is a combination of magnetization and magnetic field, but the exact relationship is slightly different between the two systems of units. The defining equations are

B = H + 4πM (CGS) (1a) or B = µ0 (H + M) (SI) (1b)

In CGS units the B and 4πM quantities are given units of Gauss, and H is in Oersteds. But equation (1a) makes it clear that Gauss and Oersteds are dimensionally equivalent. It is not unusual to hear someone refer to the intensity of a magnetic field in Gauss, although strictly speaking, it is not correct. The constant µ0 = 4π x 10–7 Tesla-m/A is called the permeability of free space. It is used only in SI units to relate H and M to B. Occasionally, µ0M = J is reported as a polarization in SI units of Tesla.

Magnetic materials display hysteresis. Hard magnetic materials (for example,


Pow

er E

lect

roni

cs Major hysteresis loop, M vs. H

odd material compared to ferrite, samarium-cobalt or neodymium-iron-boron. Because alnico has a relatively low intrinsic coercive field, alnico magnets must be relatively longer to prevent demagnetization. This leads to a geometry that looks long and skinny rather than the traditional short and fat. It also explains why there are few new designs in alnico.

Additionally, it is tough to develop a feel for relative magnetic strength. Mathematics only goes so far in describing magnetic behavior. As a demonstration, I’ve put ferrite, SmCo and NdFeB of the same size on a steel plate. You can feel the difference in the force as you pull the magnets off the plate. Because the magnets are the same size, it is clear to the observer that the difference they are feeling must be due to the materials itself. This demonstration works well in-person. It is much tougher to visualize these differences rather than experience them.

The bottom line is that magnetic design can be a complicated subject. But a firm grasp of the basics can help engineers get started on the right track.

THREE VECTORSMost magnetic behavior is characterized in terms of three interrelated vector quantities that either describe what is happening inside the permanent magnet or in the region around it. These quantities are:

B, Magnetic induction or flux density. A vector quantity describing the concentration of magnetic flux at a point in space. It is expressed in terms of flux lines per unit of cross-sectional area. This quantity is reported as Gauss in CGS units (centimeter-gram-second) and as Tesla in SI units (Système International d’Unités).

The major hysteresis loop for magnetic material typically takes on a rectangular shape when plotted with the magnetic field on the X axis and magnetization on Y. Application of a sufficiently large magnetic field saturates the sample in both directions, making the enclosed area as large as possible for the sample in question.


PERMANENT MAGNETS

172016 DESIGN WORLD — EE Network

permanent magnets) show a large amount of it, and soft magnetic materials show little hysteresis. A word of Greek derivation, hysteresis describes the observation that magnetic materials are highly nonlinear, meaning their response to a stimulus lags behind the applied stimulus in a consistent and repeatable manner, a hysteresis loop. The stimulus in this case is an applied magnetic field and the material’s response is the magnetization or induction.

In typical graphs of hysteresis loops, the X-axis shows the applied magnetic field, H, and the Y-axis shows the magnet’s internal response, the magnetization, M. Starting from the origin,

magnetization rises with increasing field until it hits a maximum, defined as the saturation magnetization, MS. The minimum applied magnetic field necessary for saturation is called HS; it is an important practical parameter, although frequently ignored and difficult to define precisely.

As the field drops to zero, we find much of the magnetization remains, defined as Br, the remanent magnetization. As the applied magnetic field becomes negative, a significant field is necessary to reduce the magnetization to zero, defined as Hci, the intrinsic coercive field. Applying a larger negative field will

Major hysteresis loop B vs. H Demagnetization curves for a permanent magnet

The variation of saturation magnetization with temperature in nickel, demonstrating a Curie temperature of 358° C

Typical irreversible loss behavior of a permanent magnet

HysteresisloopscanbeplottedwithmagneticfieldontheXaxisandfluxdensityon the Yaxis.Insteadofflatteningoutathighfields,thiscurveassumesafixedandconstantpositiveslope.

Demagnetizationcurvesareconstructedbyshowingthesecondquadrantofmagneticfieldversusmagnetizationandversusfluxdensityplots.Thiseliminatesinformationthatisredundantinthecompletehysteresisloops.

Thetemperaturequalitiesofnickelmimicthoseofmostpermanentmagnetswhichallhavealargesaturationmagnetizationatlowtemperature.Themagnetizationfallsaccordingtothepatternvisiblehere,finallyreachingzeroattheCurietemperatureTc.

Atypicalcurveforirreversiblelossshowshowfluxlossrisesthelongeramagnetseeshightemperatures.Herethesoaktemperaturewas150°C.Thetimescaleinthese plots is typically logarithmic.

Magnetics_EE_2016_Power_Vs5.MD.indd 17 2/24/16 11:17 AM


Pow

er E

lect

roni

cs

saturate the sample in the opposite direction, and a symmetric pattern can be seen in the rest of the curve that finally closes back on itself again in the first quadrant.

The loop in plots of hysteresis is called a major hysteresis loop because sufficient magnetic field is applied to saturate the sample in both directions, making the enclosed area as large as possible for the sample in question. Applying a larger maximum field will not affect the enclosed area of the loop, nor does it yield any additional information about the sample. Any data supplied about magnetic materials should be based on a major hysteresis loop, unless clearly stated otherwise.

Applying a smaller maximum field, one that does not fully saturate the sample, yields a minor hysteresis loop. Typically, minor loops should be avoided for permanent magnets, as the size and features of a minor loop depend heavily on the maximum field supplied and not solely on the magnetic properties of the sample.

Hysteresis loops can alternatively be drawn with flux density, B, which is the externally available magnetic flux, plotted against the applied magnetic field. Instead of flattening out at high fields, this curve assumes a fixed and constant positive slope. The remanent magnetization, Br happens at the same place, but the field to reduce the flux density to zero, Hc, is less than Hci. The B versus H curve also gives the energy product, (BH)max, which is the largest product of B × H in the second quadrant. (This entity represents the area of the rectangles traced out in B-H major hysteresis loops and demagnetization curves.)

There is substantial redundancy in showing the complete hysteresis loop. The main interest is when the

applied magnetic field is in the opposite direction of the magnetization or induction. So the standard convention in the permanent magnet industry is to show just the second quadrant of both curves together. The resulting graphs are called demagnetization curves.

Assuming ideal behavior—meaning no drop in magnetization once a sample is saturated—all the following equations are true as equalities. When real cases are considered, the greater than (>) applies.

4πMs π Br (CGS) (2a)

µ0Ms ≥ Br (SI) (2b)

Br ≥ Hc (CGS) (3a)

Br ≥ µ0Hc (SI) (3b)

Hci ≥ Hc (CGS and SI) (4)

(Br/2)2 ≥ (BH)max (CGS) (5a)

(Br/2)2 ≥ µ0(BH)max (SI) (5b)

TEMPERATURE AND MAGNETSPermanent magnets are sensitive to temperature in several ways. We will consider three.

The first thermal quality to consider is the Curie temperature. Like nickel, most permanent magnets have a large saturation magnetization at low temperature. The magnetization begins to fall rapidly at higher temperatures, finally reaching zero at the Curie temperature Tc. Therefore, Tc is always worthy of note, but there are other thermal considerations that should be considered as well.

Because of the behavior of magnetization over temperature, there are small and reversible changes

Typical properties—The four families of permanent magnets Ferrite Alnico SmCo NdFeB

PROPERTY CERAMIC 8 ALNICO 5 REC-20 REC-26 MQ1-B NMX-42BH

Br (kG) 4.0 12.5 9.0 10.5 6.9 13.1

≥ (%/αC) -0.18 -0.02 -0.05 -0.03 -0.105 -0.11

(BH)max MGOe 3.8 5.5 20 26 10 42

Hci (kOe) 3.3 0.64 20+ 10+ 9 14

β (%/°C) +0.4 -0.015 -0.3 -0.3 -0.4 -0.6

Hs (kOe) 10 3 20 30 35 25

Tc (°C) 450 890 727 825 360 310

Electrical conductivity Poor Good Good Good Fair Good



PERMANENT MAGNETS

in the magnetic parameters around room temperature. These variations are quantified in two temperature coefficients, α and β, which report the behavior of Br and Hci, respectively, the second thermal qualities to consider. The defining equations for these two parameters are:

α = Br ΔT Hci ΔT 1 ΔBr 1 ΔHci β =

To design a magnet application correctly, a temperature range should be given along with the reported values of α and β, for instance, 20 to 150° C. These are also called reversible temperature coefficients because these effects are completely reversed when the magnets are returned to room temperature.

The third and most conceptually difficult thermal quality is called irreversible loss, which arises when a magnet sees elevated temperatures for an extended period of time. In other words, one might see a flux loss of a few percent when a disc-shaped magnet with L/D = 0.5 is held at 150° C for a few hours. Unlike reversible loss, this loss is not reversed by returning the sample to room temperature. However, all the magnetic flux can be recovered by remagnetizing the part. In


practice, remagnetization may not always be practical, so the losses are irreversible in this context.

Note that the time scale in a typical irreversible loss plot is logarithmic. The exact loss observed depends on the time, temperature and shape of the sample. Usually, the losses flatten out over time, so this is a stable situation. Irreversible loss is commonly observed in rare earth magnets and should be considered in the design process.

FAMILIES OF PERMANENT MAGNETSThere are four major permanent magnet materials. Each material has at least one compelling reason to maintain its commercial interest.

Ferrite magnets cost the least in terms of dollars-per-kilogram of material, but they have the weakest magnetic properties, and Hci drops as the temperature decreases.

Alnico magnets have the lowest temperature coefficient of Br and the highest maximum operating temperature, but they have the lowest Hci. This latter property makes them easy to magnetize, but also prone to demagnetization.

Samarium-cobalt magnets, both the SmCo5 and the Sm2Tm17 (where Tm is a combination of transition metals: typically Co, Fe, Cu and Zr or Hf) compositions, have exceptionally high Hci values and a relatively low temperature coefficient of Br, but the combined high cost of both samarium and cobalt makes these the most expensive magnets in use today.

Neodymium-iron-boron magnets come in bonded and sintered forms. Sintered neodymium magnets offer the highest (BH)max, while bonded magnets offer more design flexibility. Corrosion resistance and maximum operating temperature have been long- standing objections, but generally overcome with appropriate coatings and an understanding of the thermal properties of NdFeB, when designing a new device.

All in all, an understanding of the basics of magnetic materials begins with understanding their hysteresis loops and the basic permanent magnet parameters, for example, Br, Hci and (BH)max. Each type of permanent magnet material offers one or more reasons to consider it for use. The obligation of design engineers is to understand these reasons and the application under consideration to make good material selections.

REFERENCESSpontaneous Materialsspontaneousmaterials.com

The International System of Units (SI) NIST Special Publication 330 (2008) Barry N. Taylor and Ambler Thompson, Editors.

R. B. Goldfarb and F. R. Fickett, U.S. Department of Commerce, National Bureau of Standards, Boulder, CO, March 1985, NBS Special Publication 696.

Rainer Hilzinger and Werner Rodewald, Magnetics Materials, Vacuumschmelze 2013 pp. 134-135, Publicis Publishing, ISBN 978-3-89578-352-4.

A plot of magnetic strength over time, from Spontaneous Materials, reveals why rare-earth magnets have become widely used. But each magnetic material has its own set of design issues.

Magnetics_EE_2016_Power_Vs5.MD.indd 19 2/24/16 11:16 AM


Pow

er E

lect

roni

cs

GaN components boost power density in supplies

S ilicon power FETs have come a long way over the past 35 years. Modern high-voltage superjunction FETs have developed over the

past 15 years to even exceed what was thought to be the “theoretical limit of performance” for silicon. As a result, power supplies have become more efficient, with many suppliers offering 96% peak efficiency today. Power density has benefited as well. Today’s high- performance server power supplies have power densities of 40 W/in.3 or better. Yet the industry goal is to significantly boost this power density for the future. You might wonder how this can be accomplished. First, it is important to look at what is limiting density right now. Many power supply functions occupy a relatively fixed volume for given power

supply requirements. For example, the size of the dc bus capacitor is typically dictated by the holdup-time requirement. But the EMI (electromagnetic interference) filters, PFC (power factor correction) and dc-to-dc stages, along with their thermal management, represent more than half the power supply volume. These functions can potentially be much smaller if the operating frequency can significantly rise without the corresponding penalty of increased switching loss. So, why not simply increase the operating frequency of existing power supply topologies to improve the density? Often, the limiting factor is the power semiconductors in both the PFC and dc-to-dc circuits. These power transistors and rectifiers operate in switching modes that have

Pow

er E

lect

roni

cs

ERIC PERSSONInfineon Technologies

To understand why gallium-nitride components shrink the size

of power supplies, examine how they dissipate energy at high

frequencies.

Typical power supply volume contribution by section

% of Total Volume Power Supply Section

25% Input—connector, EMI filter, Inrush limiter, protection network

22% DC bus capacitors—energy storage for holdup time

10% Thermal management—heatsinks, fans

10% Bias power supply

15% PFC stage

18% DC-DC stage and O-ring FETs

A review of typical volumes occupied by the main functions in a modern 40 W/in.3 power supply as a percentage of total volume shows that the volume of some functions don’t change much for given power supply requirements. For example, the size of the dc bus capacitor is typically dictated by the holdup-time requirement. But the EMI filters, PFC and dc-dc stages along with thermal management—which represents more than half of the entire power supply volume—can become much smaller if the operating frequency can rise without incurring more switching loss.

Infineon_EE_2016_Power_Vs3.MD.indd 20 2/23/16 1:44 PM

GaN COMPONENTS


Comparing typical parameters of 600-V Superjunction FET versus 600-V GaN HEMT

For 55 mΩ nominal Rds(on)

Best superjunction E mode GaN HEMT

Qg (typ) 68 nC 6 nC

Qoss

(typ) 420 nC 44 nC

Eoss

(typ) 8 μJ 7 μJ

Qrr (typ) 6,000 nC 0 nC

A comparison of a silicon superjunction FET to a GaN HEMT reveals the HEMT is still better in Eoss, but by a much smaller margin than Qoss, where it is 10x better.

frequency-dependent switching losses. Thus, boosting the switching frequency also increases the switching loss in the power semiconductors. This is exactly the opposite of what is needed: If the power supply density is to rise, the losses in the power supply will have to drop.

DENSITY AND EFFICIENCYMany of the common methods of improving efficiency do not necessarily improve power density. In fact, oftentimes the opposite is true. For example, lowering the operating frequency of a power supply reduces the frequency-dependent switching losses. But the lower frequency also necessitates use of bigger magnetics. Thus the tradeoff: The highest-efficiency power supply will have the lowest density for a given design approach. But improved efficiency (reduced power loss) is necessary to improve density for two reasons: First, lower losses also correspondingly reduce the size of heat sinks, fans and other thermal management devices. Moreover, for a given maximum internal temperature limit, the internal power dissipated must drop as the physical volume of the power supply shrinks. Form factors with more optimal surface area/volume ratio can help, but the overall trend is toward less ability to dissipate power as physical volume shrinks. So efficiency and density are linked: Smaller power supply volume requires a proportional reduction in losses. It is possible to boost frequency without additional switching loss through use of a different control strategy.

Regardless of the transistor technology, zero-voltage switching (ZVS) is one key to minimizing switching loss and enabling use of higher frequencies. The majority of power supply topologies are based on the concept of using transistors to switch a voltage source into an inductive load. The goal of ZVS is to use energy stored in the parasitic capacitance of the switching device along with inductor current to losslessly commutate the switch capacitance. This is instead of hard-switching where the transistor forces

Graphed here is a typical boost converter stage switching waveform at the moment the transistor turns on. At t0- the inductor current flows through the diode into the bus capacitor. The switch node voltage Vsw equals the bus voltage. At t0, Q1 turns on and begins conducting current, ramping up to the inductor current IL at t1. Note that the voltage at the switch node Vsw has not moved yet. If the diode was perfect and there was no reverse-recovery charge, the switch node voltage would begin to move toward zero at t1. But if D1 is a PN junction diode (or the body diode of a synchronous rectifier), the diode cannot immediately stop conducting, so the current in Q1 continues to ramp up, as does the corresponding reverse current in the diode. This continues until t2, when the diode stops conducting. Here, there is a significant reverse-recovery current in addition to the steady-state inductor current, and the transistor Q1 carries the total current while seeing the full bus voltage across its drain-source.

commutation and dissipates the energy stored in the device capacitance.

HARD SWITCHINGHard switching is common in the PFC stage of a power supply. Consider a typical boost converter stage consisting of an inductor, diode, bus capacitor and transistor acting as a switch. Also consider the switching waveform at the moment the transistor turns on. Initially, current flowing through the inductor charges the bus capacitor, and the switch node voltage (Vsw) therefore

Reverse Recovery

Loss

EOSSLoss

CrossoverLoss

t2

t3t1t0IL

VSW = VDS (Q1)

IL VSW



Pow

er E

lect

roni

cs

equals the bus voltage. When the switching transistor turns on, it begins conducting current. Transistor current ramps up to the level of the inductor current (IL). But the voltage at the switch node Vsw initially doesn’t drop. If the diode was perfect and there was no reverse-recovery charge, Vsw would begin to drop toward zero immediately. But if the diode is a PN junction diode (or the body diode of a synchronous rectifier), then it cannot immediately stop conducting, so the current in the transistor continues to ramp up, as does the corresponding reverse current in the diode. Current continues to ramp up until the diode recovers its ability to block voltage and stops conducting. At this point, there is a significant reverse-recovery current on top of the steady-state inductor current. The transistor is supporting the total current while still supporting the full bus voltage across its drain-source. This leads to the high peak power dissipated in the transistor during the turn-on interval. The power dissipation, P(t), curve is the product of device current times voltage. It peaks at the same instant as the inductor current. Finally, the current through the transistor discharges the capacitance of the


Pow

er E

lect

roni

cs A PFC circuit operated in critical conduction mode can effect soft switching. Here, when the inductor current reaches zero, the equivalent circuit is an LC. The capacitance is the combined output capacitance of the switch plus the parasitic capacitance of the diode and inductor. The initial condition of the circuit is that this capacitance is charged to the bus voltage and at t0 it will resonate and ring down to negative bus voltage. But the switch will clamp it as it crosses zero volts. Q1 can then be turned on to begin the next cycle. Since it turns on when Vds is already zero, there is no turn-on loss.

switch node and drives the voltage to zero, thus dissipating the energy stored in the diode capacitance and the transistor’s own self-capacitance. To summarize, in a hard-switched turn-on, there are three main energy loss mechanisms each cycle:1. Commutation or crossover loss—proportional to current rise time; faster turn-on means lower loss2. Reverse recovery loss (does not apply for Schottky diode)—depends mostly on the diode characteristic; diodes with large Qrr, like the body diode of a superjunction FET, can have an extremely large Qrr and completely dominate the turn-on loss3. Eoss loss—this is the energy stored in the capacitance of the switch node (including the switch itself, the diode, and parasitic capacitance in the inductor) that gets dissipatively discharged each time the switch turns-on.

ZVSAs a simple example of ZVS, consider the same boost PFC circuit as in the previous example, but with a different control strategy. The previous example operated in continuous conduction mode (CCM) with the current through the inductor never falling to zero. Now suppose the

current is allowed to reach zero each cycle. Of course this means the ripple current of the PFC stage has a much higher magnitude (so there is both more rms current and corresponding conduction loss). But allowing the inductor to fully discharge sets up the condition for lossless commutation of the diode—essentially free ZVS. When the inductor current reaches zero, the equivalent circuit is an LC, but the capacitance is not the big bulk capacitor on the dc bus—it is blocked by the diode. Instead, the total capacitance is the combined output capacitance of the switch plus the parasitic capacitance of the diode and inductor. The initial condition of the circuit is that C is charged to the bus voltage. The capacitance will resonate, and its voltage will ring down to negative bus voltage. But the switch will clamp the voltage as it crosses zero volts. When the switch does turn on again, the voltage across it is already zero, thus eliminating the turn-on switching loss. This mode of operating the PFC circuit is known as critical conduction mode, or CrCM. The concept of using small amounts of energy stored in the inductor or in device capacitance is common practice for enabling ZVS in a variety of topologies and control strategies. The LLC converter is a

VSW, ILVSW

IL

VSWt0

Vbus

IL

PFC CIRCUIT


GaN COMPONENTS


good example of a dc-to-dc stage that uses resonance to realize ZVS in the back half of a power supply. As previously mentioned, ZVS can work with any type of switch, but here is where the big difference between conventional silicon FETs and GaN HEMTs becomes important: If the effective capacitance of the switch is made much smaller, the time required to make the ZVS transition also drops correspondingly. Or alternatively, the time can be made the same, but the amount of stored charge needed can be reduced correspondingly. It is desirable to operate at a higher frequency to realize higher energy density. This is where the shorter transition time becomes important. Normally in the LLC circuit, the ZVS transition only takes a small percentage of the total resonant period—for example, it may be 330 nsec, about 5% of the period, for one cycle of a typical 150-kHz operating frequency. But if the frequency rises 4x to 600 kHz, the 330 nsec (per edge) becomes 20% of the period. The transition time (also known as dead time) is “non-productive” power transferring time—it is simply time spent waiting for the lossless ZVS transition. This

means that as the dead time becomes a larger percentage of the total period, the productive portion of the resonant period is proportionally smaller, and this drives the rms current much higher due to the higher peak-to-average ratio. In other words, to boost frequency significantly in a ZVS circuit, the circulating energy needed to realize ZVS must drop proportionally. Otherwise, the penalty of higher rms current on both the primary and secondary sides will kill the efficiency of the power supply, making it impossible to improve density. But the relationship between capacitance, charge and energy in modern high-voltage (superjunction) MOSFETs is complex because the capacitance is so nonlinear. This nonlinearity makes it difficult to compare devices based on datasheet capacitance values as they can change three orders of magnitude depending on voltage. It also makes a big difference between devices that are optimal for hard switching (low Eoss), versus those best for ZVS soft switching (low Qoss). A graph of Qoss versus Eoss clearly illustrates this difference. Consider the case of a 650-V, 70-mΩ-rated high-performance superjunction FET compared to a GaN

HEMT with the same rated on-resistance. The superjunction charge Qoss rises steeply, reaching 90% of its final value within the first 20 V applied. The slope then abruptly diminishes so applying another 380 V only adds 10% more charge. This behavior arises because of how charge distributes in the columnar structures of a superjunction FET. This behavior has an interesting effect: The energy needed to pump charge into the Coss at low voltage is smaller due to the V2 relationship in E = ½ CV2. Though 90% of the charge gets stored in the first 20 V, a far smaller portion of the total energy is stored by this point. This characteristic nonlinearity explains why superjunction FETs can have a relatively low Eoss for a given Qoss. It is this nonlinearity that makes them excellent (low Eoss) for hard-switching applications compared to the other silicon alternatives. In stark contrast, the GaN HEMT is a lateral device and has a low, nearly linear capacitance versus voltage. Its graph of Qoss versus voltage has a shallow slope rising to a value an order of magnitude smaller than that of the superjunction Qoss. This even distribution of charge along the voltage axis results in an Eoss having a final

The dotted lines represent charge (left axis) and the solid lines represent energy (right axis) on this graph of Qoss and Eoss versus Vds for a 650-V, 70-mΩ-rated high-performance superjunction FET compared to a GaN HEMT with the same rated on-resistance. The superjunction charge (dotted blue line) rises steeply to 90% of its final value within the first 20 V, then rises the final 10% over the remaining 380 V because of how charge distributes in the columnar structures of a superjunction FET. This characteristic nonlinearity explains why superjunction FETs can have a relatively low Eoss for a given Qoss, and can be excellent (low Eoss) for hard-switching circuits. Note that the GaN HEMT has ~10x lower Qoss, which is the important parameter for ZVS applications.

Qos

s (n

C)

E oss

(μJ)

Vds (V)


DISRUPTIVE TECHNOLOGIES TO GLOBAL MARKETS

Our manufacturers’ disruptive technologies drive GaN, SiC and ultra-capacitor customer solutions over a wide range of power applications, including battery charger, induction heating, laser, medical, motor drive, renewable energy, RF generator, UPS and welding. Get started at rellpower.com.

More than just another distributor.

40W267 Keslinger RoadP.O. Box 393LaFox, IL 60147-0393

Copyright © 2016 Richardson Electronics, Ltd. All rights reserved.

[email protected] RELLPOWER.COM

®

800.348.5580 / 630.208.2200

C

M

Y

CM

MY

CY

CMY

K

RELL-Power-WTWH-Handbook-2016-02.pdf 1 2/11/2016 1:46:02 PM

Richardson 2-16 EE.indd 24 2/23/16 12:43 PM

GaN COMPONENTS


value (at 400 V) almost equaling that of the superjunction device, since Eoss is the integration of charge times voltage. All in all, the HEMT is better in Eoss, but by a much smaller margin than Qoss where it is 10x improved. To further illustrate the effect of Qg and Qoss, consider the well-known LLC circuit. Suppose the same LLC circuit is used for comparison of both Si and GaN on the primary side running at about 325 kHz, delivering 750 W from a 385-V bus. The waveforms for the superjunction FETs on the primary of the LLC show once the upper gate turns off, the drain voltage takes more than 350 nsec to slew from bus to zero volts. The superjunction nonlinear charge creates long, shallow tails on the voltage that mandate the long dead time. Even with a 350-nsec dead time, the voltage has not yet reached zero (so it is near ZVS) when

the lower gate turns on. It may seem that turning on slightly early, before the voltage across the switch is really zero, is a small compromise, but it is not. Don’t forget that nearly half the Eoss still remains at only 20 V on the drain because of the nonlinearity. In other words, this dead time is pushed to be as short as possible without significant compromises in power loss (and efficiency). With the GaN HEMT in the same circuit, under the same conditions, the gate voltage has a much faster rise and fall time than in the superjunction device. This is a result of the gate driver having a much easier time driving the low-charge gate of the GaN device. Moreover, the low Qoss of the HEMT makes the drain voltage linear and much faster as well. This lets the dead time be 3x shorter, and there is have no additional loss from non-ZVS.

Thus, for power supply applications that require higher density, GaN HEMTs provide far superior properties that enable higher operating frequencies while simultaneously reducing the overall losses. In the example and conditions described here with a 750-W output, the overall efficiency of the LLC converter is 96.5% for superjunction, 97.8% for GaN. This is a 38% reduction in power loss, from 27.2 to 16.9 W because of the GaN HEMT.

REFERENCESInfineon Technologies AG infineon.com/gan


Using an example of an LLC converter, the ZVS waveforms for superjunction FET Vgs (top graph) are displayed at

5 V/div, Vds at 100 V/div, and a time scale of 100 nsec/div. Below it are ZVS waveforms for a GaN HEMT with the same settings. In the superjunction FET, the upper gate turns off and the drain voltage takes more than 350 nsec to slew from bus to zero volts. The superjunction nonlinear charge creates long, shallow tails on the voltage that mandate the long dead time. Even after a 350-nsec dead time, when the lower gate turns on the voltage has not yet reached zero (so it is near ZVS). In viewing the same waveforms for the GaN HEMT, note the gate voltage has a much faster rise and fall time than the superjunction device. This is a result of the gate driver having a much easier time driving the low-charge gate of the GaN device. Moreover, due to the low Qoss of the HEMT, the drain voltage is linear and much faster as well. Because of this, the dead time can be 3x shorter and have no additional loss from non-ZVS.

GaN COMPONENTS

LLC CONVERTER



Pow

er E

lect

roni

cs

Test instruments help tame power factor and signal integrity

E lectronics that include switch-mode power supplies (SMPSs) and VFDs depend on non-linear components to minimize size and cost. This approach, however, forces designers to note

the impact non-linear current flows have on the signal integrity of the input mains power

SMPSs, for example, typically draw non-linear current and are therefore significant sources of harmonics on the power network. The same applies to VFDs, used with ac motors as often found in HVAC equipment and industrial fans. Even LED lamps, despite their low power consumption, can produce significant harmonics when a multitude of LED lamps are installed in a building or house.

The negative effects of harmonics are well-known. They can include overheating of cabling and transformers, current flowing through neutral ac conductors, nuisance tripping of circuit breakers, high electromagnetic emissions and reduced life of motors and transformers. In addition, the presence of these harmonics forces transformers and cables to have a higher rated capacity than would otherwise be necessary. To alleviate such problems, designers

begin by assessing the actual harmonic levels. Unfortunately, getting a handle on the amplitude and character of harmonics present can be a challenging task.

Measuring the Total Harmonic Distortion (THD), power factor, and harmonic levels of the input mains provides a good indicator of how a device-under-test impacts power quality.

IEEE and IEC guidelines describe how measurements must take place and also specify maximum harmonic levels for power electronics connected to the power network (IEEE 519, IEC 61000-3-2). Many standards related to power quality are written in reference to harmonics, because this provides a rational way to specify testable limits on distortion.

If a power supply meets the limits specified in standards, it will be a minimal burden to power quality. Harmonic analysis provides an excellent tool to help identify the source of signal integrity issues and meet the specifications that standards require.

Pow

er E

lect

roni

cs

SIMRAN NANDAKeysight Technologies

Harmonic analysis helps design

electronics that will behave well

on the power grid.

A screen capture of the mains input to a switch-mode power supply reveals a slightly distorted read out of the voltage (top trace). The current drawn by the device (middle trace) is clearly a non-linear load. The bottom trace is the power transferred.

Keysight_EE_2016_Power_Vs4.MD.indd 26 2/23/16 2:11 PM

PROVEN Critical Component Integrity

DC-DC Converters

2V to 10,000 VDC Output

Electronics, Inc.143 Sparks Ave., Pelham, NY 10803

2 to 5000 VDC OutputsUltra MiniatureSurface Mount and Thru HoleSingle and Dual Isolated OutputsMilitary Upgrades AvailableAV/AV/SM/AVR Series

100 to 10,000 VDC OutputProportional Control Up to 10 WattsVV Series

Programmable to6000 VDC OutputHVP Series

HIGH POWER-Wide Input Range2 to 350 VDC OutputIsolated to 300 Wattsin 1/2 and Full BrickLP/HP/XP Series

New Products Released!

Military Components-550 to +850C Operating TempMilitary EnvironmentalScreening Available3.3 to 500 VDC OutputsM/MV SeriesIsolated - Regulated

Military Applications-400 and -550 to +850C Operating TemperaturesWide Input Range3.3 to 350 VDC outputsin 1/2 Brick and Full BrickMilitary Upgrades AvailableLF/LM/FM Series

Wide Input Range8 to 60 VDC Input2 to 100 VDC OutputIsolated-RegulatedOR/IR/JR/KR Series2-20 Watts

36 to 170 VDC InputsTerminal Strips - Thru Hole3.3 to 48 VDC OutputsSingle and Dual OutputLV/HV Series

5 to 500 VDC Outputto 50 WattsWide Input RangeIsolated / Regulated / QP Series

www.picoelectronics.comComplete Listing of Entire Product Line

DC-DC Converters • AC-DC Power Supplies • Transformers and Inductors.

E Mail: [email protected]

Also AC-DC Single and 3 Phase.Power Factor Corrected. 3 Watts to 2000 Watts Models.

Call Toll Free: 800-431-1064

• Miniature DC-DC 6,000-10,000 VDC Outputs(0.5” x 1.0” x 0.5”, 9.5 grams)

• Adjustable V out, from 100 to 1,500 VDC, 3 Watts• New High Input Voltages available to 900VDC in

Outputs from 5 VDC to 500 VDC, to 300 Watts• AC-DC with Three Phase Input in Single Brick Isolated

Outputs to 300 VDC Power Factor Corrected to 300 Watts

Fax: 914-738-8225

A40_DesigWrld_9x10_875_Layout 1 1/19/16 10:27 AM Page 1

PICO Electronics 2-16 EE.indd 27 2/23/16 12:44 PM


Pow

er E

lect

roni

cs

IEC 61000-3-2 Class A (equipment that draws <16 A per phase) dictates harmonic limits as delineated in this table.

Pow

er E

lect

roni

cs

Maximum current draw for harmonics as specified by IEC 61000-3-2 Class A

Odd harmonics (n)

Max current (A)

Even harmonics (n)

Max current (A)

3 2.3 2 1.08

5 1.14 4 0.43

7 0.77 6 0.30

9 0.40 8-40 0.23×8/n

11 0.33

13 0.21

15-39 0.15×15/n

BASICS OF HARMONIC ANALYSISHarmonic analysis uses a Fourier series to decompose a complicated periodic waveform into a set of simple sinusoids. The original waveform may be arbitrarily complex and most likely doesn’t have an analytical equation. The waveform can be digitized, however, and run through a harmonic analysis to generate a set of sinusoids (harmonics) that, when added together, approximate the original waveform.

Readers may recall that harmonics are organized by frequency. The lowest non-zero-frequency harmonic is the fundamental (or first harmonic). All other harmonics have frequencies that are integer multiples of the fundamental. The second harmonic is twice the fundamental frequency, the third is three times, and so on.

Now consider a simple system of an ac source and a load. The average power delivered to the load from the source over one power line cycle is given by:

v(t)i(t) dt (1)1T

T

0∫Pavg=

where T is the period of the power line cycle, v(t) is the voltage across the load, and i(t) is the current through the load. Substituting the harmonics formulation for the v(t) and i(t) waveforms gives:

2∑Pavg= V0 I0 + cos (Øvn - Øin) (2)

∞

n=1

VnIn

This is a more useful form. Decomposing Pavg into sums of harmonics now lets you compute the power being delivered

by a particular harmonic frequency. For example, if you want to know how much power the ninth harmonic delivers, you can compute it directly by multiplying the amplitudes of the ninth harmonic voltage and current times the cosine of the angle difference between them:

2Pavg9 = cos (Øv9 - Øi9)

V9I9

An interesting prediction from the above decomposition is that both voltage and current must have corresponding harmonics present. Otherwise there will be no average power transfer from that particular harmonic. For example, if the voltage contains just the fundamental, and the current contains just the third harmonic, the average power will be zero.

If the voltage is a clean sine wave and the current waveform is non-sinusoidal, only the fundamental will transfer power. All the higher harmonics in the current waveform will be unproductive.

One goal of power system design is to maximize the power factor, PF, defined as:

VA VrmsIrms

PowerFactor = =W Pavg

But the harmonics that don’t transfer power work against this goal. They don’t contribute to Pavg, but they boost the VrmsIrms. The extra harmonic voltage and/or current is not used, but the power system still must carry the extra harmonic voltage and/or current and incur the associated losses. To maximize power transfer efficiency, it’s therefore beneficial to minimize higher harmonics.

A sensible first step before harmonic analysis is to measure the THD, especially if the purpose is to troubleshoot power quality problems. This measurement can be done with a true-RMS digital multimeter with a bandwidth and sampling rate high enough to capture the higher harmonic frequencies. If the THD level is low (less than 3 to 5% depending on amplitude), there are no harmonics issues. However, if either THD is too high or you want to characterize the performance of your device, the THD measurement is not enough. You need the full breakdown of harmonic amplitudes.

You may also want to conduct multiple runs, evaluating the device in different operating modes and varying load conditions. With general-purpose instruments (digital multimeters, spectrum

Keysight_EE_2016_Power_Vs4.MD.indd 28 2/23/16 2:12 PM


TAME POWER FACTOR

A power analyzer is a useful tool for measuring the power in a test setup. Here a function generator produces a voltage signal with just the fundamental (60 Hz) and a current with just the third harmonic (180 Hz). The bottom trace shows the power waveform. Note the average power transferred per cycle is effectively zero, though the voltage, current, and power traces have significant amplitude.

This example of a harmonics measurement shows the power line spectrum for a device generating mostly odd harmonics but with a fifteenth harmonic that is too high to meet the IEC 610000-3-2 standard for grid power quality.

analyzers or scopes), you may find the data collection and post-processing for harmonics analysis extremely time-consuming. In addition, the process of comparing the harmonic levels to published standards can be tedious.

The modern power analyzer simplifies and speeds up such procedures with features specifically intended for harmonic analysis. Data collection, frequency domain processing, and harmonics analysis are built-in. Power analyzers typically have high sample rates and a graphical display that permits viewing the waveform in a manner analogous to that of a scope. This visualization is invaluable because you can quickly verify what you are measuring and that cabling or misconnections aren’t causing issues.

The power analyzer can simultaneously display other relevant information such as THD, frequency, amplitude, phase of the voltage, current, and power waveforms. Once the measurement setup is verified, you can simply turn on the harmonic analysis and the power analyzer will display the harmonics in a table and/or visual bar graph format.

Many power analyzers also have built-in pass/fail limits based on widely used standards. This makes for quick, convenient and immediate feedback on how the harmonics of the device under test stack up against the compliance limits.

All in all, use of a power analyzer to minimize higher harmonics and maximize power transfer permits a quick investigation of harmonic content. Harmonic analysis can provide actionable guidance to address signal integrity issues in power systems with periodic switching waveforms. Using a modern power analyzer simplifies measurements and lets you model complicated interacting voltage and current waveforms in a way that is easy to understand.

REFERENCESKeysight Technologies keysight.com

Keysight_EE_2016_Power_Vs5.MD.indd 29 2/24/16 10:03 AM


Pow

er E

lect

roni

cs

A close look at the Level VI power supply spec

Market enthusiasm for new high-tech devices is undiminished. The Consumer Electronics Association, in its July 2015 U.S. Consumer Electronics Sales and

Forecasts report, predicted a total industry revenue rise of 2.4% to $285B. Major contributors include continued growth in smartphone and notebook/netbook sales and new categories giving buyers more reasons to spend. These include 4K ultra-high-definition televisions, connected-home technologies and wearable devices.

On the other hand, governments are trying to limit demand for electrical energy, both to ensure grid stability and to combat climate change. As sales of high-volume consumer electronics continue rising, the resulting squeeze on resources force greater energy efficiency measures. A key concern has been the energy wasted when electrical devices are in standby modes. In 1998, this was estimated at about 5% of all electricity generated.

Eco-design initiatives, such as Energy Star in the U.S. and ErP in the E.U., have evolved to limit both the active and standby power of equipment such as televisions, set-top boxes, computers and domestic appliances. The external power adapters of equipment such as notebook computers and office machinery have also come under

Pow

er E

lect

roni

cs

JEFF SCHNABELCUI Inc.

DOE power supply efficiency

specifications target a broader

array of power supplies to save

energy and reduce greenhouse-

gas emissions.

A plot of efficiency levels versus power supply output shows how the Level VI average efficiency specification for power supplies stacks up against earlier Level IV and Level V specifications.

Efficiency versus Output

CUI_EE_2016_Power_Vs3.MD.indd 30 2/24/16 11:45 AM

IF TRAVIS AND ANDY CAN’T DESIGN YOUR TRANSFORMER - NO ONE CAN!

@rencoelecRenco is always looking for good people to relocate to Florida headquarters.Please, email resume to [email protected]

No-charge samples can be shipped with-in 48 hours*

[email protected] [email protected]

Andy & Travisare up for

the challenge

Give them try,you will not bedisappointed

Cool & efficientoperation

Reliability &life tested

Smallest packageavailability

Low cost automatedwinding design

WIRE: round, square, rectangular, and flat foilINSULATION: thermal varnish, epoxy, silicone, oilTEMPERATURE: up to Class H 180 CelsiusSAFETY: UL approval service, including class 2FREQUENCY: 50/60 Hz, 400 Hz, up to 500 kHzPOWER: 10 watts to 10,000 wattsTOOLING: bobbins, cores, tubes, laminations, basesTOPOGRAPHIES: too many to mention!*No-charge samples will only be provided with an approved price quotation

Renco 2-16 EE.indd 31 2/24/16 2:03 PM


Pow

er E

lect

roni

cs


Pow

er E

lect

roni

csscrutiny. In 2004, California became the first authority to introduce mandatory specifications to reduce the energy dissipated by external power supplies.

Today, the International Energy Efficiency Marking Protocol for External Power Supplies is the globally accepted framework setting out limits on standby consumption and average efficiency for such adapters. Within this framework, the latest specification is Level VI, which was introduced by the U.S. Dept. of Energy (DOE) on February 10, 2016. This requires any external power supply manufactured after this date and shipped into the U.S. to meet new efficiency targets. DOE estimates this more rigorous standard will save 47 million tons of CO2 emissions annually.

NEW SPECIFICATION, TOUGHER TARGETSThe new Level VI specifications demand higher average efficiency and lower no-load power consumption compared to current Level V power supplies. The new Level VI specification is also significantly more complex than previous versions. Several categories of power supplies are now defined, and new classifications will be regulated for the first time. Some of the newly regulated products include multi-voltage power supplies as well as single-voltage power supplies over 250 W. By including these high-power adapters, Level VI will have a significant impact in markets for industrial equipment not previously covered by the marking protocol.

Examples of basic single-voltage supplies include those used by routers, set-top boxes, portable industrial devices and other consumer electronic devices in the 1 to 49-W power range. Telecommunication network infrastructure equipment, appliances and office equipment are typical applications in the 49 to 250-W range while industrial devices featuring motors or lasers will be covered by the above-250-W category.

Required efficiency levels for several categories spelled out in the Level VI standard are specified as a calculation involving supply output power. Cranking through a few of these calculations for basic and low-voltage supplies reveals that the standard’s distinction between basic- and low-voltage appears to apply a less stringent requirement for low-voltage supplies. This certainly helps ensure the manufacture of such supplies is economically viable in the more price sensitive market for consumer products and home automation devices that fall into this category. Similarly, running the standard’s efficiency calculation reveals the ≤1-W category appears to have a relatively undemanding efficiency specification of 66% or less. But currently, external adapters at these low power levels are rare.

The new specification also introduces a distinction between direct operation and indirect operation. It

Power supply type ≤1 W 1 to 49 W 49 to 250 W ≥250 W

Single-Voltage ac/dc (basic

voltage)

No load (max.)Average

efficiency (min.)

0.1 W

0.5 × Pout + 0.16

0.1 W

0.071 × ln(Pout) – 0.0014 × Pout + 0.67 (see note)

0.21 W

0.880

0.5 W

0.875

Single-Voltage ac/ac (basic

voltage)

No load (max.)

Average efficiency (min.)

0.21 W

0.5 × Pout + 0.16

0.21 W

0.071 × ln(Pout) – 0.0014 × Pout

+ 0.67

0.21 W

0.880

0.5 W

0.875

Single-Voltage ac/dc (low voltage)

No load (max.)


0.1 W

0.517 × Pout + 0.087

0.1 W

0.0834 × ln(Pout) –

0.0014 × Pout + 0.609

0.21 W

0.870

0.5 W

0.875

Single-Voltage ac/ac (low voltage)

No load (max.)


0.21 W

0.517 × Pout + 0.087

0.21 W

0.0834 × ln(Pout) –

0.0014 × Pout + 0.609

0.21 W

0.870

0.5 W

0.875

Multiple-Voltage

No load (max.)


0.3 W

0.497 × Pout + 0.067

0.3 W

0.075 × ln(Pout) + 0.561

0.3 W

0.860

-

Level VI power-supply categories and specifications

Note: ln = natural logarithm e.g. for Pout = 40W then ln(Pout) = 3.6888 so the efficiency would calculate as 87.5%

Here are the specifications for all categories defined in the Level VI specification. Note that low-voltage power supplies are defined as having output voltage less than 6 V and output current greater than 550 mA. Basic voltage refers to a power supply that is not a low-voltage power supply.

Learn more about the MDA800 and sign up to receive a free Power Basics Poster: teledynelecroy.com/static-dynamic-complete

MDA800 Series Motor Drive Analyzers

One Instrument, One Solution Come see us at APEC 2016 Booth 2037

teledynelecroy.com/motor-drive-analyzer | teledynelecroy.com/contactus© 2015 Teledyne LeCroy, Inc. All rights reserved.

Line Voltage, Current, and Power – The BasicsTHREE-PHASE

LINE VOLTAGE

Single-phase line voltage consists of one voltage vector with:

• Magnitude (voltage) • Angle (phase)

Typically, the single-phase is referred to as “Line” voltage, and is referenced to neutral.

Neutral Line

The single-phase voltage vector rotates at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

At any given moment in time, the voltage magnitude is V * sin(α) • V = magnitude of voltage vector • α = angle of rotation, in radians

The resulting time-varying “rotating” voltage vector appears as a sinusoidal waveform with a fixed frequency

• 50 Hz in Europe • 60 Hz in US • Either 50 or 60 Hz in Asia • Other frequencies are sometimes used in non-utility supplied power, e.g. • 400 Hz • 25 Hz

SINGLE-PHASE

Three-phase line voltage consists of three voltage vectors. • By definition, the system is “balanced” • Vectors are separated by 120° • Vectors are of equal magnitude • Sum of all three voltages = 0 V at Neutral

Typically, the three phases are referred to as A, B, and C, but other conventions are also used: • 1, 2, and 3 • L1, L2, and L3 • R, S, and T

The three voltage vectors rotate at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

The resulting time-varying “rotating” voltage vectors appear as three sinusoidal waveforms • Separated by 120° • Of equal peak amplitude

Voltage value = VX*sin(α) • VX = magnitude of phase voltage vector • α = angle of rotation, in radians

VA-B

VA-N

120° VA

VB

VC

Neutral

A

B

C

ω (rad/s) or freq (Hz)

120°

120°

120°Neutral

Neutral Line


200

Time

AC Single-Phase “Utility” Voltage120VAC

Volts

(Pea

k), L

ine-

Neut

ral 150

100

50

0

-50

-100

-150

-200

120 VAC Example

800

Time

AC Three-Phase “Utility” Voltage480VAC , Measured Line-Line

Volts

(Pea

k), L

ine-

Line

600

400

200

A-B VoltageB-C VoltageC-A Voltage

0

-200

-400

-600

-800

480 VAC Example

800

Time

AC Three-Phase “Utility” Voltage480VAC , Measured Line-Neutral

Volts

(Pea

k), L

ine-

Neut

ral

600

400

200

A-N VoltageB-N VoltageC-N VoltageThree-phase Rectified DC

0

-200

-400

-600

-800

480 VAC Example

Line-Line Voltage Measurements

Line-Neutral Voltage Measurements

Important to Know • Voltage is stated as “VAC”, but this is really VRMS • Rated Voltage is Line-Neutral • VPEAK = 2 * VAC (or 2 * VRMS ) • 169.7 V in the example below • VPK-PK = 2 * VPEAK • If rectified and filtered • VDC = 2 * VAC = VPEAK

Voltages can be measured two ways: • Line-Line (L-L) • Also referred to as Phase-Phase • e.g. from VA to VB, or VA-B • Line-Neutral (L-N) • Neutral must be present and accessible • e.g. from VA to Neutral, or VA-N • VL-L conversion to VL-N • Magnitude: VL-N * 3 = VL-L • Phase: VL-N - 30° = VL-L

Important to Know • Voltage is stated as “VAC”, but this is really VRMS • Rated Three-phase voltage is always Line-Line (VL-L) • Line-Line is A-B (VA-B), B-C (VB-C), and C-A (VC-A) • Line-Line is sometimes referred to as Phase-Phase • VPEAK(L-L) = 2 * VL-L • 679 V in the example to the right • VPK-PK(L-L) = 2 * VPEAK(L-L)

If a neutral wire is present, three-phase voltages may also be measured Line-Neutral • VL-N = VL-L/ 3 • 277 VAC (VRMS) in this example • VPEAK = 2 * VL-N • 392 V in the example to the right • VPK-PK = 2 * VPEAK

If all three phases are rectified and filtered • VDC = 2 * VL-N * 3 = VPEAK * 3 = 679 V in the example to the right

LINE CURRENT

Like voltage, the single-phase current vector rotates at a given frequency • Typically, 50 or 60 Hz

At any given moment in time, the current magnitude is I*sin(α) • I = magnitude of current vector • α = angle of rotation, in radians

The resulting time-varying “rotating” current vector appears as a sinusoidal waveform

Like voltage, the resulting time-varying “rotating” current vectors appear as three sinusoidal waveforms • Separated by 120° • Of equal peak amplitude for a balanced load

Current value = IX*sin(α) • IX = magnitude of line current vector • α = angle of rotation, in radians

Neutral Line


By definition, the system is “balanced” • Vectors are separated by 120˚ • Vectors are of equal magnitude • Sum of all three currents = O A at neutral (provided there is no leakage of current to ground)

Like voltage, three-phase current has three different line current vectors that rotate at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

SINGLE-PHASE

THREE-PHASE

A

B

C


120°

120°

120°Neutral

Line Current Measurements

10 ARMS Example

Important to Know • Current is stated as “lAC”, but this is really IRMS • Line currents can represent either current through a coil, or current into a terminal (see image below) depending on the three-phase winding connection • IPEAK = 2 * IRMS

• 14.14A for a 10 ARMS current in the example to the right • IPK-PK = 2 * IPEAK

-15

-12

-9

-6

-3

0

3

6

9

12

15

Line

Cur

rent

(Pea

k)

Time

AC Three-Phase "Line" Currents

A Current B Current C Current

I C I A

I B

A

B

C

NA

B

C I C I A

I B

LINE POWERSINGLE-PHASE

Electric Power • “The rate at which energy is transferred to a circuit” • Units = Watts (one Joule/second)

For purely resistive loads • P = I2R = V2/R = V * I • The current vector and voltage vector are in perfect phase

IInductive load

V

P ≠ V * I

φN

I

Capacitive load

P ≠ V * I

Vφ

N

Real Power

Imag

inar

y Pow

er

S

P

Qφ

φ

φ

φ

QB

QC

QA

SB

SA

SC

PB

PA

PC

Phase Angle (φ) • Indicates the angular difference between the current and voltage vectors • Degrees: - 90° to +90° • Or radians: -π/2 to + π/2

For capacitive and inductive loads • P ≠ V * I since voltage and current are not in phase

• For inductive loads • The current vector “lags” the voltage vector angle φ • Purely inductive load has angle φ = 90°

• Capacitive Loads • The current vector “leads” the voltage vector by angle φ • Purely capacitive load has angle φ = 90°

THREE-PHASE

For purely resistive loads • PA = VA-N * IA • PB = VB-N * IB • PC = VC-N * IC • PTOTAL = PA + PB + PC

As with the single-phase case, Power is not the simple multiplication of voltage and current magnitudes, and subsequent summation for all three phases.

Apparent Power for each Phase • |S|, in Volt-Amperes, or VA • = VRMS * IRMS for a given power cycle

Real Power for each Phase • P, in Watts • = instantaneous V * I for a given power cycle

Voltage is measured L-L • Neutral point may not be accessible, or • L-L voltage sensing may be preferred

Current is measured L-N

L-L voltages must be transformed to L-N reference:

Calculations are straightforward, as described above: • PTOTAL= PA + PB + PC • STOTAL = SA + SB + SC • QTOTAL = QA + QB + QC

Voltage is measured L-L on two phases • Note that the both voltages are measured with reference to C phase

Current is measured on two phases • The two that flow into the C phase

Mathematical assumptions: • Σ(IA + IB + IC) = 0 • Σ(VA-B + VB-C + VC-A) = 0

This is a widely used and valid method for a balanced three-phase system

VB-C

VA-C

I A

I B

A

B

C

N

A

B

C

VB-C

VA-CI A

I B

Single-phase, Non-resistive Loads

Phase Angle

Single-phase Real, Apparent and Reactive Power

Three-phase, Resistive Loads Three-phase, Non-resistive Loads

Two Wattmeter Method – 2 Voltages, 2 Currents with Wye (Y or Star) or Delta (∆) Winding

VB-N

VA-N

VC-N

I C I A

I B

A

B

C

N

VA-B

VB-C

VC-A

I CI A

I B

A

B

C

N

Power Factor (PF, or λ) • cos(φ) for purely sinusoidal waveforms • Unitless, 0 to 1, • 1 = V and I in phase, purely resistive load • 0 = 90° out of phase, purely capacitive or purely inductive load • Not typically “signed” – current either leads (capacitive load) or lags (inductive load) the voltage

VoltageCurrent

Note: Any distortion present on the Line voltage and current waveforms will result in power measurement errors if real power (P) is calculated as |S|*cos(φ). To avoid measurement errors, a digital sampling technique for power calculations should be used, and this technique is also valid for pure sinusoidal waveforms.

Resistive loadV

P=V * I

NI

Power Factor

Line-Line Voltage Sensing Case

V B-N

V A-N

PTOTAL = VA-N * IA + VB-N * IB + VC-N * ICV C-N

I B

I A

I C

N φ

V B-N

V A-N

PTOTAL ≠ VA-N * IA + VB-N * IB + VC-N * ICV C-N

I B

I AI C

N

Three-phase, Any Load

Apparent Power • |S|, in Volt-Amperes, or VA • = VRMS * IRMS for a given power cycle

Real Power • P, in Watts • = instantaneous V * I for a given power cycle

Reactive Power • Q, in Volt-Amperes reactive, or VAr • Q = S2 - P2

• Does not “transfer” to load during a power cycle, just “moves around” in the circuit

Reactive Power for each Phase • Q, in Volt-Amperes reactive, or VAr • Q = S2 - P2

• PTOTAL = PA + PB + PC • STOTAL = SA + SB + SC • QTOTAL = QA + QB + QC

PTOTAL = VA-C * IA + VB-C * IBSTOTAL= VRMSA-C * IRMSA + VRMSB-C * IRMSBQTOTAL = STOTAL2 - PTOTAL2

“Not True” RMS

Wye (Y) 3-phase Connection • Neutral is present in the winding • But often is not accessible • Most common configuration

Delta (∆) 3-phase Connection • Neutral is not present in the winding (in most cases)A

B

C

N

A

B

CFor one power cycle

• The digital samples are grouped into measurement cycles (periods) • For a given cycle index i…. • The digitally sampled voltage waveform is represented as having a set of sample points j in cycle index i • For a given cycle index i, there are Mi sample points beginning at mi and continuing through mi + Mi -1. • Voltage, current, power, etc. values are calculated on each cycle index i from 1 to N cycles.

Digital Sampling Technique for Power Calculations

Period 1Mi = 18 points


mi = point 7 mi = point 25

VPK-PK

“True” RMS

For one power cycle

Three-Phase Winding Connections

VRMS

IRMS

Real Power(P, in Watts)

Apparent Power(S, in VA)

Reactive Power(Q, in VAR)

Power Factor (λ)

Phase Angle (φ)

VRMSi = Vj2Mi

1mi + Mi - 1

Σj=mi

IRMSi = Ij2Mi

1mi + Mi - 1

Σj=mi

Pi = Vj * IjMi

1mi + Mi - 1

Σj=mi

Si = VRMSi * IRMSi

magnitude Qi = Si2 - Pi2

Sign of Qi is positive if the fundamental voltage vector leads the fundamental current vector

λi =Pi

Si

magnitude Φi = cos-1λi

Sign of Φi is positive if the fundamental voltage vector leads the fundamental current vector

Formulas Used for Per-cycle Digitally Sampled Calculations

VRMS = VPK-PK 2 21 VRMS = VAC

2

MDA800 SeriesMotor Drive Analyzers8 channels, 12-bits, 1 GHz

tlec-2015_power-poster-final-cmyk-HIRES.pdf 1 11/9/15 2:54 PM

tlec-mda-poster-designworld-handbook.indd 1 2/5/16 2:48 PMCUI_EE_2016_Power_Vs3.MD.indd 32 2/24/16 2:14 PM

LEVEL VI POWER SPEC

applies only to units intended for direct operation. Direct power supplies are defined as being able to function in the end product without a battery. An indirect power supply is not a battery charger, but cannot operate the end product without the assistance of a battery.

Some classes of direct power supplies are exempt from the need to comply with Level VI. These include devices that require FDA approval as a medical device in accordance with section 360c of title 21, and ac/dc power supplies with nameplate output voltage less than 3 V and nameplate output current greater than or equal to 1,000 mA that charges the battery of a product that is fully or primarily motor operated.

CHANGES UNDER THE SKINPower supply makers have generally found that Level VI efficiency levels couldn’t be met with the same supply designs used for Level V or earlier designs. So new topologies have generally been necessary to

AcomparisonoftheefficiencyofequivalentLevelVandLevelVIpowersupplies,measuredat25,50,75and100%offullload(7.5V,4A)showsthatLevelVIefficiencyimprovesatalllevels,andparticularlyatlowerloads.

handle the more stringent efficiency demands. For example, CUI anticipated the new Level VI regulations by introducing compliant adapters as early as the latter half of 2014. Important design changes were necessary to satisfy the new, stricter targets, based on CUI’s established topologies. For units under 120 W, this is a flyback topology, while adapters over 120 W use an LLC resonant topology.

To meet the tougher standards, low-voltage/high-current models now feature

synchronous rectification in the secondary side. Replacing conventional rectifier diodes with low-RDS(ON) MOSFETs has eliminated diode losses resulting in a net saving when the power to run the associated MOSFET controller IC is taken into account.

The PWM control strategy is significantly different compared to the previous generation. In CUI’s Level V adapters, the main control IC typically operates at a fixed frequency of 65 kHz, but the latest

Learn more about the MDA800 and sign up to receive a free Power Basics Poster: teledynelecroy.com/static-dynamic-complete

MDA800 Series Motor Drive Analyzers

One Instrument, One Solution Come see us at APEC 2016 Booth 2037

teledynelecroy.com/motor-drive-analyzer | teledynelecroy.com/contactus© 2015 Teledyne LeCroy, Inc. All rights reserved.

Line Voltage, Current, and Power – The BasicsTHREE-PHASE

LINE VOLTAGE

Single-phase line voltage consists of one voltage vector with:

• Magnitude (voltage) • Angle (phase)

Typically, the single-phase is referred to as “Line” voltage, and is referenced to neutral.

Neutral Line

The single-phase voltage vector rotates at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

At any given moment in time, the voltage magnitude is V * sin(α) • V = magnitude of voltage vector • α = angle of rotation, in radians

The resulting time-varying “rotating” voltage vector appears as a sinusoidal waveform with a fixed frequency

• 50 Hz in Europe • 60 Hz in US • Either 50 or 60 Hz in Asia • Other frequencies are sometimes used in non-utility supplied power, e.g. • 400 Hz • 25 Hz

SINGLE-PHASE

Three-phase line voltage consists of three voltage vectors. • By definition, the system is “balanced” • Vectors are separated by 120° • Vectors are of equal magnitude • Sum of all three voltages = 0 V at Neutral

Typically, the three phases are referred to as A, B, and C, but other conventions are also used: • 1, 2, and 3 • L1, L2, and L3 • R, S, and T

The three voltage vectors rotate at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

The resulting time-varying “rotating” voltage vectors appear as three sinusoidal waveforms • Separated by 120° • Of equal peak amplitude

Voltage value = VX*sin(α) • VX = magnitude of phase voltage vector • α = angle of rotation, in radians

VA-B

VA-N

120° VA

VB

VC

Neutral

A

B

C


120°

120°

120°Neutral

Neutral Line


200

Time

AC Single-Phase “Utility” Voltage120VAC

Volts

(Pea

k), L

ine-

Neut

ral 150

100

50

0

-50

-100

-150

-200

120 VAC Example

800

Time

AC Three-Phase “Utility” Voltage480VAC , Measured Line-Line

Volts

(Pea

k), L

ine-

Line

600

400

200

A-B VoltageB-C VoltageC-A Voltage

0

-200

-400

-600

-800

480 VAC Example

800

Time

AC Three-Phase “Utility” Voltage480VAC , Measured Line-Neutral

Volts

(Pea

k), L

ine-

Neut

ral

600

400

200

A-N VoltageB-N VoltageC-N VoltageThree-phase Rectified DC

0

-200

-400

-600

-800

480 VAC Example

Line-Line Voltage Measurements

Line-Neutral Voltage Measurements

Important to Know • Voltage is stated as “VAC”, but this is really VRMS • Rated Voltage is Line-Neutral • VPEAK = 2 * VAC (or 2 * VRMS ) • 169.7 V in the example below • VPK-PK = 2 * VPEAK • If rectified and filtered • VDC = 2 * VAC = VPEAK

Voltages can be measured two ways: • Line-Line (L-L) • Also referred to as Phase-Phase • e.g. from VA to VB, or VA-B • Line-Neutral (L-N) • Neutral must be present and accessible • e.g. from VA to Neutral, or VA-N • VL-L conversion to VL-N • Magnitude: VL-N * 3 = VL-L • Phase: VL-N - 30° = VL-L

Important to Know • Voltage is stated as “VAC”, but this is really VRMS • Rated Three-phase voltage is always Line-Line (VL-L) • Line-Line is A-B (VA-B), B-C (VB-C), and C-A (VC-A) • Line-Line is sometimes referred to as Phase-Phase • VPEAK(L-L) = 2 * VL-L • 679 V in the example to the right • VPK-PK(L-L) = 2 * VPEAK(L-L)

If a neutral wire is present, three-phase voltages may also be measured Line-Neutral • VL-N = VL-L/ 3 • 277 VAC (VRMS) in this example • VPEAK = 2 * VL-N • 392 V in the example to the right • VPK-PK = 2 * VPEAK

If all three phases are rectified and filtered • VDC = 2 * VL-N * 3 = VPEAK * 3 = 679 V in the example to the right

LINE CURRENT

Like voltage, the single-phase current vector rotates at a given frequency • Typically, 50 or 60 Hz

At any given moment in time, the current magnitude is I*sin(α) • I = magnitude of current vector • α = angle of rotation, in radians

The resulting time-varying “rotating” current vector appears as a sinusoidal waveform

Like voltage, the resulting time-varying “rotating” current vectors appear as three sinusoidal waveforms • Separated by 120° • Of equal peak amplitude for a balanced load

Current value = IX*sin(α) • IX = magnitude of line current vector • α = angle of rotation, in radians

Neutral Line


By definition, the system is “balanced” • Vectors are separated by 120˚ • Vectors are of equal magnitude • Sum of all three currents = O A at neutral (provided there is no leakage of current to ground)

Like voltage, three-phase current has three different line current vectors that rotate at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

SINGLE-PHASE

THREE-PHASE

A

B

C


120°

120°

120°Neutral

Line Current Measurements

10 ARMS Example

Important to Know • Current is stated as “lAC”, but this is really IRMS • Line currents can represent either current through a coil, or current into a terminal (see image below) depending on the three-phase winding connection • IPEAK = 2 * IRMS

• 14.14A for a 10 ARMS current in the example to the right • IPK-PK = 2 * IPEAK

-15

-12

-9

-6

-3

0

3

6

9

12

15

Line

Cur

rent

(Pea

k)

Time

AC Three-Phase "Line" Currents

A Current B Current C Current

I C I A

I B

A

B

C

NA

B

C I C I A

I B

LINE POWERSINGLE-PHASE

Electric Power • “The rate at which energy is transferred to a circuit” • Units = Watts (one Joule/second)

For purely resistive loads • P = I2R = V2/R = V * I • The current vector and voltage vector are in perfect phase

IInductive load

V

P ≠ V * I

φN

I

Capacitive load

P ≠ V * I

Vφ

N

Real Power

Imag

inar

y Pow

er

S

P

Qφ

φ

φ

φ

QB

QC

QA

SB

SA

SC

PB

PA

PC

Phase Angle (φ) • Indicates the angular difference between the current and voltage vectors • Degrees: - 90° to +90° • Or radians: -π/2 to + π/2

For capacitive and inductive loads • P ≠ V * I since voltage and current are not in phase

• For inductive loads • The current vector “lags” the voltage vector angle φ • Purely inductive load has angle φ = 90°

• Capacitive Loads • The current vector “leads” the voltage vector by angle φ • Purely capacitive load has angle φ = 90°

THREE-PHASE

For purely resistive loads • PA = VA-N * IA • PB = VB-N * IB • PC = VC-N * IC • PTOTAL = PA + PB + PC

As with the single-phase case, Power is not the simple multiplication of voltage and current magnitudes, and subsequent summation for all three phases.

Apparent Power for each Phase • |S|, in Volt-Amperes, or VA • = VRMS * IRMS for a given power cycle

Real Power for each Phase • P, in Watts • = instantaneous V * I for a given power cycle

Voltage is measured L-L • Neutral point may not be accessible, or • L-L voltage sensing may be preferred

Current is measured L-N

L-L voltages must be transformed to L-N reference:

Calculations are straightforward, as described above: • PTOTAL= PA + PB + PC • STOTAL = SA + SB + SC • QTOTAL = QA + QB + QC

Voltage is measured L-L on two phases • Note that the both voltages are measured with reference to C phase

Current is measured on two phases • The two that flow into the C phase

Mathematical assumptions: • Σ(IA + IB + IC) = 0 • Σ(VA-B + VB-C + VC-A) = 0

This is a widely used and valid method for a balanced three-phase system

VB-C

VA-C

I A

I B

A

B

C

N

A

B

C

VB-C

VA-CI A

I B

Single-phase, Non-resistive Loads

Phase Angle

Single-phase Real, Apparent and Reactive Power

Three-phase, Resistive Loads Three-phase, Non-resistive Loads

Two Wattmeter Method – 2 Voltages, 2 Currents with Wye (Y or Star) or Delta (∆) Winding

VB-N

VA-N

VC-N

I C I A

I B

A

B

C

N

VA-B

VB-C

VC-A

I CI A

I B

A

B

C

N

Power Factor (PF, or λ) • cos(φ) for purely sinusoidal waveforms • Unitless, 0 to 1, • 1 = V and I in phase, purely resistive load • 0 = 90° out of phase, purely capacitive or purely inductive load • Not typically “signed” – current either leads (capacitive load) or lags (inductive load) the voltage

VoltageCurrent

Note: Any distortion present on the Line voltage and current waveforms will result in power measurement errors if real power (P) is calculated as |S|*cos(φ). To avoid measurement errors, a digital sampling technique for power calculations should be used, and this technique is also valid for pure sinusoidal waveforms.

Resistive loadV

P=V * I

NI

Power Factor

Line-Line Voltage Sensing Case

V B-N

V A-N

PTOTAL = VA-N * IA + VB-N * IB + VC-N * ICV C-N

I B

I A

I C

N φ

V B-N

V A-N

PTOTAL ≠ VA-N * IA + VB-N * IB + VC-N * ICV C-N

I B

I AI C

N

Three-phase, Any Load

Apparent Power • |S|, in Volt-Amperes, or VA • = VRMS * IRMS for a given power cycle

Real Power • P, in Watts • = instantaneous V * I for a given power cycle

Reactive Power • Q, in Volt-Amperes reactive, or VAr • Q = S2 - P2

• Does not “transfer” to load during a power cycle, just “moves around” in the circuit

Reactive Power for each Phase • Q, in Volt-Amperes reactive, or VAr • Q = S2 - P2

• PTOTAL = PA + PB + PC • STOTAL = SA + SB + SC • QTOTAL = QA + QB + QC

PTOTAL = VA-C * IA + VB-C * IBSTOTAL= VRMSA-C * IRMSA + VRMSB-C * IRMSBQTOTAL = STOTAL2 - PTOTAL2

“Not True” RMS

Wye (Y) 3-phase Connection • Neutral is present in the winding • But often is not accessible • Most common configuration

Delta (∆) 3-phase Connection • Neutral is not present in the winding (in most cases)A

B

C

N

A

B

CFor one power cycle

• The digital samples are grouped into measurement cycles (periods) • For a given cycle index i…. • The digitally sampled voltage waveform is represented as having a set of sample points j in cycle index i • For a given cycle index i, there are Mi sample points beginning at mi and continuing through mi + Mi -1. • Voltage, current, power, etc. values are calculated on each cycle index i from 1 to N cycles.

Digital Sampling Technique for Power Calculations



mi = point 7 mi = point 25

VPK-PK

“True” RMS

For one power cycle

Three-Phase Winding Connections

VRMS

IRMS

Real Power(P, in Watts)

Apparent Power(S, in VA)

Reactive Power(Q, in VAR)

Power Factor (λ)

Phase Angle (φ)

VRMSi = Vj2Mi

1mi + Mi - 1

Σj=mi

IRMSi = Ij2Mi

1mi + Mi - 1

Σj=mi

Pi = Vj * IjMi

1mi + Mi - 1

Σj=mi

Si = VRMSi * IRMSi

magnitude Qi = Si2 - Pi2

Sign of Qi is positive if the fundamental voltage vector leads the fundamental current vector

λi =Pi

Si

magnitude Φi = cos-1λi

Sign of Φi is positive if the fundamental voltage vector leads the fundamental current vector

Formulas Used for Per-cycle Digitally Sampled Calculations

VRMS = VPK-PK 2 21 VRMS = VAC

2

MDA800 SeriesMotor Drive Analyzers8 channels, 12-bits, 1 GHz

tlec-2015_power-poster-final-cmyk-HIRES.pdf 1 11/9/15 2:54 PM

tlec-mda-poster-designworld-handbook.indd 1 2/5/16 2:48 PM

Level V versus VI

CUI_EE_2016_Power_Vs3.MD.indd 33 2/24/16 11:44 AM


Pow

er E

lect

roni

cscontrollers for Level VI units improve efficiency as load drops by reducing the switching frequency to 22 kHz. Improving light-load efficiency is important because the IEC-approved test method for assessing average efficiency (AS/NZS 4665) calls for power to be measured at 25, 50, 75 and 100% of rated load. The arithmetic average of the data from all four points is then calculated to determine overall average efficiency. Because the efficiency at 25% load is the lowest of the four points, improving performance at light load results in better average efficiency.

In addition, the latest controllers draw lower quiescent current than their predecessors. This helps meet the aggressive Level VI no-load targets.

Careful management of power-factor correction (PFC) circuitry is another feature of the new Level VI units. Although PFC is mandatory for power supplies rated at 90 W or over, the circuitry can be disabled at lower

loads. This eliminates significant energy losses, thereby helping to improve both the average efficiency and no-load consumption. Ripple noise can be slightly higher when the PFC circuitry is de-activated, but this should not be a problem for most applications.

Despite changes throughout the design, most of the new Level VI units retain the same case sizes and external appearance as the earlier models. On the other hand, greater average efficiency has helped to reduce typical working temperatures, resulting in an improvement in reliability.

MANDATORY COMPLIANCEThe new Level VI efficiency standards for external power supplies are currently only mandatory in the U.S., but in practice it is forcing OEMs worldwide to adjust their purchasing and supply-chain arrangements. In any case, historical patterns suggest that other territories, such as the E.U., are likely to adopt the

Level VI specification themselves.It is usually more economical

and straightforward to ship products with the same power supply type to all markets globally, so it makes sense to ensure that all units comply with the highest standard currently in force. This helps to minimize product-management challenges and avoid any potential for costly shipping errors.

Level VI power supplies have been available in the market for over a year now, and OEMs have had every opportunity to prepare for the transition to Level VI. Anyone who is unsure whether the power supplies they are now purchasing are compliant would be well advised to look for the Level VI marking on the label or check with their supplier.

REFERENCESMore information on the Level VI standard: cui.com/efficiencystandards

Unique Power Devices

www.ixys.com

Reverse Conducting IGBTs(BiMOSFETs™)

Reverse Blocking IGBTs MOS Gated Thyristors

C

G

E

1

2

3

A

KG

• "Free" intrinsic body diode

• High power density

• High frequency operation

• MOS gate turn on for drive simplicity

• Reverse blocking capability

• Monolithic series diode

• Soft reverse recovery

• Positive temperature coefficient of on-state voltage

• Very high current capability

• Anti-parallel diodes

• Isolated mounting surface

• Low gate drive requirement

APPLICATIONS • Resonant-mode power supplies • Uninterruptible Power Supplies • Laser and X-ray generators • High voltage pulser circuits

APPLICATIONS • Current source inverters • Matrix converters • Bi-directional switches • 3-level power converters

APPLICATIONS • Capacitive discharge circuits • Ignition circuits • Overcurrent protection • Solid state surge protection

2.62.52.9

VCE(sat)

typ.TJ=25°C

(V)

20011670

IC25

TC=25°C(A)

Part Number

IXBK75N170IXBL64N250IXBH20N360HV

170025003600

VCES

(V)

2.52.52.3

VCE(sat)

typ.TJ=25°C

(V)

252535

IC25

TC=25°C(A)

Part Number

IXRP15N120IXRA15N120IXRH25N120

±1200±1200±1200

VCES

(V)

3.53.56.4

VCE(sat)

typ.TJ=25°C

(V)

7.67.6

15.5

IC25

TC=25°C(A)

Part Number

IXHH40N150HVIXHX40N150V1HVMMJX1H40N150

150015001500

VCES

(V)

EUROPEIXYS [email protected]+49 (0) 6206-503-249

USAIXYS [email protected]+1 408-457-9042

ASIAIXYS Taiwan/IXYS [email protected]@ixyskorea.com

POWER

CUI_EE_2016_Power_Vs3.MD.indd 34 2/23/16 4:10 PM

TDK 2-16 EE.indd 35 2/23/16 12:48 PM


Pow

er E

lect

roni

cs

Multiple charging modes from a single amplifier

T he good news about wireless charging techniques is that they do away with messy wiring that tethers phones and other

appliances to wall outlets. The bad news is that there are multiple wireless charging schemes and they are generally incompatible with each other. Clearly, this incompatibility is not part of the formula for widespread adoption of wireless power technology.

This is the reason why manufacturers are trying to come up with a multi-mode capable wireless power design. Such a design could handle the two main competing wireless technologies: inductive coupling (also known as Qi), adopted by the Wireless Power Consortium (WPC) along

with the Power Matters Alliance (PMA), and the highly resonant method adopted by the Alliance for Wireless Power (A4WP). Recently PMA and A4WP merged to become known as AirFuel.

An amplifier that can operate to all the wireless power standards must be capable of operating at both high (6.78 MHz) and low (100 to 367 kHz) frequencies efficiently. A single amplifier design like this has been elusive, so multi-mode source solutions have resorted to using multiple amplifiers, each driving their own corresponding source coil. This approach has generally been expensive with limited market appeal.

However, there is a simple single-amplifier topology based on eGaN FETs that is capable of operating to all of the mobile device wireless power Block diagram for a typical wireless power system

Pow

er E

lect

roni

cs

MICHAEL A. DE ROOIJVP, Applications Engineering

Efficient Power Conversion

Though there are two standards for charging appliances wirelessly, a

single circuit can be devised to serve as a charging node for both of them.

EPC_EE_2016_Power_Vs3.MD.indd 36 2/23/16 2:42 PM

The Microchip name and logo, the Microchip logo and dsPIC are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. All other trademarks are the property of their registered owners. © 2015 Microchip Technology Inc. All rights reserved. 1/15DS40001764B

Broad Range of Solutions for Power Conversion Designers

www.microchip.com/intelligentpower

The power conversion market is constantly evolving, with a focus on increased effi ciencies and higher integration.

The devices you used in your last power conversion design may already be old news. No matter which topology or approach you choose, Microchip can support your design with our comprehensive portfolio of intelligent power products, which covers the spectrum from discrete analog solutions to sophisticated and fl exible power conversion using our digital signal controllers (DSCs).

The dsPIC® “GS” family of DSCs is optimized to provide full digital control of power conversion stages. Compensation loops implemented in software offer the ultimate in fl exibility, enabling designs leveraging numerous topologies to be tailored for energy effi ciency over widely varying load or environmental conditions. Complete reference designs for AC/DC and DC/DC power conversion are available to enable faster time-to-market and simplify designs.

Microchip 2-16 EE.indd 37 2/23/16 12:51 PM


Pow

er E

lect

roni

cs

Effectofdeviceloadresistancevariationonthereflectedimpedance(left)foratunedcoil-setfortheschematicshown(right).

ModifiedZVSclassDamplifierformulti-modeoperation

standards. The design is based on a modified ZVS (zero-voltage switching) class D amplifier topology coupled to a specially designed source coil. The system components have been individually experimentally verified to the A4WP class 2 and Qi/PMA standards and are designed to efficiently deliver up to 10 W.

WIRELESS POWER SYSTEM OVERVIEWThe general architecture of a wireless power transfer system generally consists of four basic building blocks: an amplifier, also known as a power converter; a source coil that includes a matching network; a device coil that includes a matching network; and a rectifier with high-frequency filtering. This architecture is similar regardless of inductive coupling or highly resonant wireless power technology.

It takes a thorough understanding of how the highly resonant coil and inductive coil systems operate to properly design the amplifier that can drive both modes efficiently. The complete schematic for a highly resonant wireless power coil system uses both shunt and series tuning in the device coil and only series tuning in the source coil. An equivalent circuit of a transformer represents the coupling between the two coils. This equivalent circuit is used to analyze the reflected impedance (labeled ZCoil_Tuned in the above diagram) presented at the source terminals as a result of varying load power. This reflected impedance would arise from a charging battery and manifests as a varying dc load

resistance (labeled Rload in the diagram).

Starting with a high dc load resistance value (shown by the red dot a in the accompanying Smith chart and decreasing until it reaches red dot b), the reflected impedance of the tuned source coil is shown by the green trace and moves in the opposite direction starting with a low impedance at green dot a and increasing until the end at green dot b. This is the negative impedance effect of the highly resonant wireless power coil-set.

There is a similar analysis for the inductive coil set. The device and source coils are still tuned, but at different operating frequencies and for different reasons. Here, the tuning is used to set a specific inductance for the reflected coil impedance that is used by the amplifier for optimal operation.

The coupling between the source coil and device coil will also affect the reflected impedance and must be considered in the design, particularly for the highly resonant operating mode. The reason will become apparent once the inductive system coil is integrated with the resonant coil and their effects must be analyzed in unison.

The coupling between the two coils is affected by changes in the distance and the turns ratio between them. If one assumes a fixed dc load resistance variation, then changes in distance between the coils, represented by d1, d2 and d3 in the figure at the top of page 39, show that as the coils get farther apart, the


Pow

er E

lect

roni

cs

SMITH CHART

TUNED COIL SET BLOCK DIAGRAM

SIMPLIFIED CLASS D STAGE

EPC_EE_2016_Power_Vs4.MD.indd 38 2/26/16 10:24 AM

MULTIPLE CHARGING MODES


Effectofdevicecouplingonthereflectedimpedance(bottomcenter)forvarioussourcecoildesignsanddistancesbetweenthesourceanddevice.ThetracesontheSmithcharthavebeenoffsetforclarity;theyactuallylayoneontopoftheother.

Integratedmulti-modecoilstructureandfrequencyselectivematching

reflected impedance variation drops. For a fixed number of turns and a fixed turns ratio between the coils, this can be a bad outcome. It can be corrected by changing the number of turns on the source coil, where reducing the number of turns will reduce the reflected impedance range. The source coil is adjusted for compatibility with existing devices and the charging coils they contain.

In a multi-mode system, the ability to custom design the coil is critical to finding the optimal operating points for both the coil set and amplifier, as well as for realizing high system efficiency. Charging systems based on inductive coupling typically demand good coupling between the device and the source. Consequently, the distance between the coils is kept fairly short.

In the case of charging systems based on highly resonant methods, the coils typically sit further apart with more spatial freedom. It is important to artificially reduce the coupling between the coils so as to re-align the reflected impedance of the source coil to the optimal operating point of the amplifier. One way to accomplish this is to reduce the number of turns on the source coil.

The coil set design is only the first step toward realizing a single amplifier operating multi-mode. The source coil must be assembled so it can be driven by a single amplifier. The figure to the left shows the top view and cross-section of such an integrated coil structure. Coil selectivity is implemented by changing the frequency of operation using applicable matching circuits. There is inherent coupling between the coils in the integrated structure that usually

COIL COUPLING AT VARIED DISTANCES

RESULTING SMITH CHART

MULTIMODE MATCHING CIRCUIT

MULTIMODE COIL STRUCTURE



Pow

er E

lect

roni

cs

forces designs to incorporate decoupling circuitry between the inductive mode and highly resonant mode coils. This decoupling can be realized using a parallel resonant circuit (LLF and CLF) as depicted in the figure at the bottom of page 39. The circuit effectively blocks high frequency current. At low frequencies, the highly resonant tuning capacitor (CHF) blocks current.

A MULTI-MODE AMPLIFIER TOPOLOGYA single amplifier can now be used to drive the integrated coil structure. The ZVS class D amplifier has proven to be a simple high-efficiency solution for 6.78-MHz highly resonant wireless power applications. The circuit that establishes ZVS for high-frequency operation must be disconnected to let the amplifier operate at low frequency in hard-switching mode without leading to over-current.

In the case of the low-frequency system, hard-switching operation was still able to yield amplifier efficiencies well above 90%. Using the ZVS technique for the amplifier, amplifier efficiencies exceeding 85% could be realized for most operating conditions when operating at 6.78 MHz.

eGaN FETs have proven to be efficient in the range of 100 to 367 kHz, despite operating in hard-switching mode. This is because of their lower input capacitance (CISS), output capacitance (COSS), excellent Miller ratio (QGD/QGS1, and lack of reverse recovery (QRR). The figure found on page 38 shows a multi-mode amplifier employing eGaN FETs. A small low-cost eGaN FET (Q3) disconnects the ZVS tank circuit from the amplifier to let it operate in hard-switching mode. This device can be driven directly from logic as it does not switch at high frequency. A suitable device for this application is the EPC2036.

The eGaN FET ZVS class D amplifier has been tested to both the A4WP class 2 (at 6.78 MHz) and Qi-A29 (at 130 kHz) standards independently. The A4WP system used the NuCurrent class 2 coil that was paired with a category 3 device and powered by an EPC9510 amplifier. The Qi system used a Würth coil set powered by a modified EPC9509 amplifier.

Tests show the A4WP test platform fully complies with the A4WP class 2 standard. The experimental evaluation used two different amplifier boards. To keep costs down, a single-ended amplifier was used for the highly resonant case. For the inductive system, the WPC (Qi) standard required a full bridge amplifier. Ultimately, a single-ended half-bridge configuration would be adopted for the final multi-mode amplifier as the coil design and tuning would be optimized for both high- and low-frequency operation.

• Test parameters: Vout = 7.0 V fsw = 130 kHz

• Test limits: 12 Vin_max

100˚ C

WPC - A29: 10 µH transmitter, Test #20 receiver EPC9114 A4WP class 2 wireless power system (left) and EPC9509 amplifier used to drive an A29 Qi wireless power coil with Qi receiver coil (below).

EPC_EE_2016_Power_Vs4.MD.indd 40 2/26/16 10:40 AM



REFERENCESFor more on inductive coupling: “System Description Wireless Power Transfer,” Vol. I: Low Power. Part 1: Interface Definition, Version 1.2, June 2015.

Power Matters Alliance. [Online] Available: powermatters.org

For more on highly resonant charging: A4WP Wireless Power Transfer System Baseline System Specification (BSS), A4WP-S-0001 v1.3.1, February 25, 2015.

For more on coil matching circuits: V. Muratov, “Multi-Mode Wireless Power Systems can be a Bridge to the Promised Land of Universal Contactless Charging,” Wireless Power Summit, Berkeley CA, U.S.A., November 2014.

For more on ZVS class D amplifiers: M. A. de Rooij, “Performance Comparison for A4WP Class-3 Wireless Power Compliance between eGaN FET and MOSFET in a ZVS Class D Amplifier,” International Exhibition and Conference for Power Electronics, Intelligent Motion, Renewable Energy and Energy Management (PCIM - Europe), May 2015.

M. A. de Rooij, “Wireless Power Handbook,” Second Edition, El Segundo, October 2015, ISBN 978-0-9966492-1-6.

A. Lidow, J. Strydom, M. de Rooij, D. Reusch, “GaN Transistors for Efficient Power Conversion,” Second Edition, Wiley, ISBN 978-1-118-84476-2.

For more on GaN transistors and the EPC2036 in particular: D. Reusch, J. Strydom, and A. Lidow, “Highly Efficient Gallium Nitride Transistors Designed for High Power Density and High Output Current DC-DC Converters,” IEEE International Power Electronics and Application Conference (PEAC), pp.456-461, 2014.

Efficient Power Conversion, “EPC2036—Enhancement Mode Power Transistor,” EPC2036 data sheet, April 2015, [Online] Available: epc-co.com/epc/Products/eGaNFETs/EPC2036.aspx

For more information on charging coils: NuCurrent. coil part number PNC21-T28L04E-152991R40, nucurrent.com

Würth Elektronik “WE-WPCC Wireless Power Charging Transmitter Coil,” 760308141 data sheet [Online] Available: katalog.we-online.com/pbs/datasheet/760308141.pdf

Würth Elektronik “WE-WPCC Wireless Power Charging Receiver Coil,” 760308102210 data sheet [Online] Available: katalog.we-online.com/pbs/datasheet/760308102210.pdf

SystemlevelefficiencycomparisonbetweentheA4WPclass2systemandtheA29Q1system.Thesystem-leveldcinputtodcoutputefficiencyresultsareshownandincludethegatedriveroperatingpower.



DC load current [mA]



Pow

er E

lect

roni

cs

Compensating for temperature in high-frequency rectifier diodes

P ulse-width-modulated converter and amplifier designs tend to spend much of their time handling significant levels

of electrical current. High currents tend to heat up the power semiconductors used in these circuits. Of course, temperature affects the operating parameters of these circuit devices. For example, the voltage across a forward-biased silicon diode drops by about 2 mV/°C rise in temperature. When a diode is reverse biased, the saturation current flowing through it is also a function of temperature, approximately doubling for each 10° C rise in temperature.

It can also be hard to predict temperature effects. For example, a physical diode that is reverse biased will have a current variation with temperature less than the values predicted by simple device models because there is a component of reverse-bias current, called a surface leakage current, that flows around the junction rather than through it. It varies with temperature at a slower rate than the reverse-bias current through the junction. A rule of thumb for discrete silicon diodes is that reverse-bias current approximately doubles for each 10° C rise in temperature.

In high-performance switching designs, it is customary to compensate for possible temperature rise and factors affected by

Pow

er E

lect

roni

cs

KOJI IKEDA K.C. NISHIOSanken Electric Co. Allegro Micro Systems

Over-temperature can be a problem in high-performance

switching supplies. New power rectifier diodes build in

safeguards to head off heat build-up.

Low High

Low

High

Res

ista

nce

(Ω)

Temperature (ºC)

As the temperature of an NTC thermistor rises, its resistance drops.

The FMKS series of diodes combine a 200-V ultra-fast recovery diode with a thermal-detect Schottky diode in a TO-220F package.

it, such as output-voltage error rate. The usual way of adjusting switching circuits for temperature is to add circuitry that reduces current through the diode as temperature rises above a certain point. The typical means of sensing temperature is a thermistor, which is usually situated in the detection circuit. The detection circuit reduces diode current as diode temperature rises above critical levels.

Of course, this kind of compensation works best if the temperature sensor is close to the diode. That is the idea behind new rectifier diodes that feature a Schottky barrier diode formed on the same die as the

Diodes_EE_2016_Power_Vs3.MD.indd 42 2/23/16 2:46 PM

Hirose 2-16 EE.indd 43 2/23/16 12:51 PM


Pow

er E

lect

roni

cs

1-800-269-6426 www.PioneerMagnetics.com

Need > 500 watts?

PMI REPLACES MOST:

REPLACE - REPLACE - REPLACE

PO W E R S UP P L I E S

AC-DCADVANCE POWERALPHA POWERATTBASLERC&D CHEROKEECOMPUTER POWER

COMPUTER PRODUCTSCOUTANTDATA POWERDELTRONFARNELLHC POWERINVENSYSJETA

LAMBDALH RESEARCHLUCENTMAGNETEKMARTEKOMEGAPOWER-MATEPOWER-ONE

POWERTECQUALIDYNESIERRACINSORENSONSTANDARD POWERTODDZYTECHAND MORE…

2.5 kW 5 kW 10 kW

DOUBLE YOUR POWER IN A 3U PACKAGESWITCHING POWER SUPPLIES

1-800-269-6426 | www.PioneerMagnetics.com

A typical temperature compensation technique for switching supplies is to use a thermistor to sense the temperature rise in a power diode that lies in the secondary of the switching circuit. The thermistor adjusts the detection voltage of the detection circuit in a way that turns on an optocoupler when the power diode temperature hits a critical point. The optocoupler turns on a circuit in the supply primary that reduces the diode current.

Pow

er E

lect

roni

csfast-recovery diode that is used as the secondary-side rectifier diode, making temperature-change detection in real time possible. This technique also reduces part count and design time.

Switched mode power supplies in applications such as audio systems and adapters generally detect an overload in rectifier diodes by monitoring the rectifier diode at the secondary side of the switched mode power supply. In a typical application circuit using an NTC thermistor, the NTC thermistor resistance drops when the temperature of the diode to which it is thermally connected goes up.

Because this temperature sensing circuit resides in the secondary of the power supply, it usually reduces current in the primary side through some kind of isolating element, generally an optocoupler, to avoid any possibility of a primary-secondary short circuit or inadvertent feedback. When the voltage at the REF pin of the shunt regulator

NTC Thermistor

Rectifier diode

Detection Circuit

Diodes_EE_2016_Power_Vs3.MD.indd 44 2/24/16 8:00 AM

1-800-269-6426 www.PioneerMagnetics.com

Need > 500 watts?

PMI REPLACES MOST:

REPLACE - REPLACE - REPLACE

PO W E R S UP P L I E S

AC-DCADVANCE POWERALPHA POWERATTBASLERC&D CHEROKEECOMPUTER POWER

COMPUTER PRODUCTSCOUTANTDATA POWERDELTRONFARNELLHC POWERINVENSYSJETA

LAMBDALH RESEARCHLUCENTMAGNETEKMARTEKOMEGAPOWER-MATEPOWER-ONE

POWERTECQUALIDYNESIERRACINSORENSONSTANDARD POWERTODDZYTECHAND MORE…

2.5 kW 5 kW 10 kW

DOUBLE YOUR POWER IN A 3U PACKAGESWITCHING POWER SUPPLIES

1-800-269-6426 | www.PioneerMagnetics.com

HIGH-FREQUENCY RECTIFIER DIODES

A Schottky barrier diode can serve as a temperature-sensing element instead of a thermistor. This approach has the advantage that the Schottky diode can be fabricated on the same die as the power diode, thus assuring a more intimate thermal connection between the two.

Rectifier diode

Detection circuit

IRIR

Schottky diode reaches the reference voltage, current flows through the optocoupler. The signal sent back to the primary side through the optocoupler is used to limit the supply of electric power. This eliminates the possibility of power supply destruction because of an overload.

When the intent is to put a temperature-sensing element on the same die as the power diode, there are advantages to using a Schottky barrier diode instead of a thermistor. Schottky barrier diodes have a leakage current (IR) that increases when temperature rises. When the temperature of the power diode rises, the leakage current from the Schottky barrier diode (which is thermally connected to it) will increase. The increase of this leakage current boosts the voltage at the REF pin of the shunt regulator.

Thus, in the same way as when using a thermistor, the current flows through an optocoupler. The signal sent back to the primary side through the optocoupler is used to limit the supply of electric power.

Diodes_EE_2016_Power_Vs3.MD.indd 45 2/24/16 8:00 AM

March 20 -24Long Beach Convention Center

Long Beach, California

The Premier Global Eventin Power ElectronicsTM

Visit the APEC 2016 websitefor the latest information:

www.apec-conf.orgwww.apec-conf.org

APEC 2016.indd 46 2/23/16 12:52 PM

HIGH-FREQUENCY RECTIFIER DIODES


In the ideal case, the temperature-detecting device is formed on the same die as the device of which temperature is detected as a means of forming a more intimate thermal connection. An example of the approach is the FMKS series diode, where a Schottky barrier diode is formed on the same die as the fast-recovery diode that is used as the secondary side rectifier diode. Thus, temperature change detection in real time is made possible, with ancillary benefits of lower part count and shorter design time.

REFERENCESAllegro MicroSystems, allegromicro.com

FMKS diode info, http://tinyurl.com/z8jy57r

Sanken Electric Co. diode info http://tinyurl.com/zxdj31b

The leakage current (IR) of a Schottky barrier diode rises with junction temperature, a quality that lets these diodes serve as temperature sensing elements.

150-300 cps

suitable for larger sizedelectrical encapsulations

+1.201.343.8983 ∙ www.masterbond.com

Cures in 75 minutes at 80°C

LOW EXOTHERM UPON CURING

Since 2001, clients have come to us for practical technical solutions in:

• Magnetic Materials

• Technical Training

• The Rare Earths

• Technical Writing

www.spontaneousmaterials.comNagging Problem, Thorny Question?

Please contact us.

Email: [email protected] • Phone: 317-614-5920Mail: 8595 E. Temple Dr., #474 • Denver, CO 80237, USA

Spontaneous Materialsa consultancy

Spontaneous Materials ad 2-16.indd 1 2/23/16 2:52 PM

Diodes_EE_2016_Power_Vs3.MD.indd 47 2/24/16 1:02 PM


Pow

er E

lect

roni

cs

How and whenMOSFETs blow upHigh temperatures and operating conditions outside the safe

operating area can sabotage MOSFETs used in switching circuits.

MAJEED AHMAD

New generations of MOSFETs incorporate features that include a low RDS(on) to minimize conduction losses and improve operational efficiency. Examples include NTMFS5C404NLT, NTMFS5C410NLT and NTMFS5C442NLT MOSFETs from ON Semiconductor, which have maximum RDS(on) values of 0.74, 0.9 and 2.8 mΩ respectively. These are complemented by NTMFS5C604NL, NTMFS5C612NL and NTMFS5C646NL models, which have breakdown voltage ratings of 60 V. Both 40- and 60-V devices are rated to operate at junction temperatures up to 175° C to facilitate greater thermal headroom for designs.

THE MOSFET (metal-oxide-semiconductor field-effect transistor) is a

primary component in power conversion and switching circuits for such applications as motor drives and switch-mode power supplies (SMPSs). MOSFETs boast a high input gate resistance while the current flowing through the channel between the source and drain is controlled by the gate voltage. However, if not appropriately handled and protected, the high input impedance and gain can also lead to MOSFET damage caused by over voltage or too-high current.

First a few basics about avoiding MOSFET damage. Obviously, Vgs and Vds must both be within limits. The same for current, Id. There is also a power limit given by

Power_Semiconductor_EE_2-16_Vs5.MD.indd 48 2/23/16 2:59 PM

MOSFETs


ONE WAY OF AVOIDING CURRENTS THAT ARE TOO HIGH IS TO PARALLEL MULTIPLE MOSFETS SO THEY SHARE LOAD CURRENT.

the maximum junction temperature. Basic values for the upper maximum on these parameters are given in the safe operating area (SOA) graph in the MOSFET datasheet. But it turns out, other thermal limits can apply. The SOA graph, for example, generally assumes an ambient temperature of 25° C with a specific junction temperature, usually below 150° C. But there are a variety of conditions that may cause high thermal gradients that may lead to expansion and cracking of the MOSFET die.

One factor to consider in this regard is that MOSFET thermal resistance is an average; it applies if the whole die is at a similar temperature. But MOSFETs designed for switch-mode power supplies can experience a wide temperature variation over different areas of their die. Optimized for on/off switching, they typically don’t work well in their linear region.

A typical failure mode for a MOSFET is a short between source and drain. In this case, only the source impedance of the power source limits the peak current. A common outcome

MOSFET power versus temperature graphs typically make assumptions about heat sinking and mounting, as is the case with this graph for an ON Semiconductor CPH3348 device.

A typical MOSFET safe-operating area graph, this one is for a CPH3348 MOSFET from ON Semiconductor. The SOA graph generally assumes an ambient temperature of 25° C, with a junction temperature below 150° C.

of a direct short is a melting of the die and metal, eventually opening the circuit. For example, a suitably high voltage applied between the gate and source (VGS) will break down the MOSFET gate oxide. Gates rated at 12 V will likely succumb at about 15 V or so; gates having a 20-V rating typically fail at around 25 V.

All in all, exceeding the MOSFET voltage rating for just a few nanoseconds can destroy it. Device

Power_Semiconductor_EE_2-16_Vs5.MD.indd 49 2/24/16 10:15 AM


Pow

er E

lect

roni

cs

manufacturers recommend selecting MOSFET devices conservatively for expected voltage levels and further suggest suppressing any voltage spikes or ringing.

TOO LITTLE GATE DRIVEMOSFET devices are designed to dissipate minimal power when turned on. And the MOSFET must be turned on hard to minimize dissipation during conduction, otherwise it will have a high resistance during conduction and will dissipate considerable power as heat.

Generally speaking, a MOSFET passing high current will heat up. Poor heat sinking can destroy the MOSFET from excessive temperature. One way of avoiding too-high current is to parallel multiple MOSFETs so they share load current.

Many P- and N-channel MOSFETs are used in topologies involving an H- or L-bridge configuration between voltage rails. Here, if the control signals to the MOSFETs overlap, the transistors will effectively short-circuit the supply. This is known as a shoot-through condition. When it arises, any supply decoupling capacitors discharge rapidly through both MOSFETs during every switching transition, causing short but large current pulses.

The way to avoid this condition is to provide a dead time between switching transitions, during which neither MOSFET is on.

The duration of MOSFET on time can greatly impact thermal resistance. This particular example graph is for a CPH3348 MOSFET from ON Semiconductor.

Over-currents even for a short duration can cause progressive damage to a MOSFET, often with little noticeable temperature rise before failure. MOSFETs often carry a high peak-current rating, but these typically assume peak currents only lasting 300 µsec or so. It is particularly important to over-rate MOSFETs for peak current when they switch inductive loads.

When switching inductive loads there must be a path for back EMF to freewheel when the MOSFET switches off. Freewheeling is the sudden voltage spike seen across an inductive load when its supply voltage is suddenly interrupted. Enhancement mode MOSFETs incorporate a diode that provides this protection.

High-Q resonant circuits can store considerable energy in their inductance and capacitance. Under certain conditions, this high energy causes the current to freewheel through the internal body diodes of the MOSFETs as one MOSFET turns off and the other turns on. (An intrinsic body diode is formed in the body-drain p-n junction connected between the drain and source. In N-channel devices, the body diode anode connects to the drain. The polarity is reversed in P-channel MOSFETs.) A problem can arise because of the slow turn-off (or reverse recovery) of the internal body diode when the opposing MOSFET tries to turn on.


MOSFETs


MOSFET body diodes generally have a long reverse recovery time compared to the performance of the MOSFETs themselves. If the body diode of one MOSFET conducts when the opposing device is on, a short circuit arises resembling the shoot-through condition. The solution to this problem involves a Schottky diode and a fast-recovery diode. The Schottky diode connects in series with the MOSFET source and prevents the MOSFET body diode from ever being forward biased by the freewheeling current. The high-speed (fast recovery) diode connects in parallel with the MOSFET/Schottky pair. It lets the freewheeling current bypass the MOSFET and Schottky completely. This ensures the MOSFET body diode is never driven into conduction.

TRANSITIONSA MOSFET dissipates little energy during its steady on and off states, but it dissipates considerable energy during times of a transition. Thus, it is desirable to switch as quickly as possible to minimize power dissipated. Because the MOSFET gate is basically capacitive, it requires appreciable current pulses to charge and discharge

the gate in a few tens of nanoseconds. Peak gate currents can be as high as an ampere.

The high impedance of MOSFET inputs can lead to stability problems. Under certain conditions, high-voltage MOSFETs can oscillate at high frequencies because of stray inductance and capacitance in the surrounding circuit (frequencies usually in the low megahertz range). Device manufacturers recommend that a low-impedance gate-drive circuit be used to prevent stray signals from coupling to the MOSFET gate.

REFERENCESON Semiconductor onsemi.com

Chassis mount solid state relays (SSRs) available in one, two or three pole switching designs. Single phase types up to 125 amps, two pole types up to 40 amps and three phase types up to 75 amps. Also offered in our new slim-line design (up to 90 amps) and compact fast-on type (up to 25 amps).

GavazziOnline.com • 847.456.6100 • [email protected]

USA Tel: 847.465.6100 Fax: 800.222.2659Canada Tel: 888.575.2275 Fax: 905.542.2248 Mexico Tel and Fax: 55.5373.7042www.GavazziOnline.com • [email protected]

twitter.com/CarloGavazziNA facebook.com/CarloGavazziNA

Visit our website for downloadable data sheets,brochures and pricing: www.GavazziOnline.com

CARLO GAVAZZI A u t o m a t i o n C o m p o n e n t s




DIN rail or chassis mount solid state contactors and SSRs which are UL508 rated for motor loads, and feature integrated heat sinks, fans and large load terminals. Designed for switching single phase loads up to 85 amps (15 Hp) and two or three pole types for switching up to 75 amps per phase (25 Hp).

Specialty SSRs for your growing demands: System Monitoring SSR for line /load voltage and load current, fused SSRs provide more protection, and 1, 2 and 3 pole proportional controllers with several switching modes: phase angle, distributed burst (1, 4 or 16 cycle), or soft start.

Carlo Gavazzi is one of the fastest growing SSR manufacturers worldwide! Contact us if you’re interested in a free evaluation sample (qualified OEMs only).

Solid State Relays and Contactors

WHAT DO YOU THINK?

Connect and discuss this and other design engineering issues with

thousands of professionals online



Pow

er E

lect

roni

cs

Measuring wireless charging efficiency in the real world

T here are a lot of claims made about the energy efficiency of wireless charging schemes. Engineers are understandably skeptical. After all, it is hard

to envision how two induction coils sitting some distance away from each other could engage in energy efficient charging. Fortunately, there is meaningful data available about wireless charging efficiency. It serves important purposes by helping decision makers assess different standards, and it enables informed choices regarding the deployment of a given standard. Additionally, efficiency data helps engineers and product designers determine which user benefits are worth the “cost” in terms of efficiency. That said, it is true that there’s limited public data comparing the real-world power-transfer efficiency of different wireless power standards. Wireless chargers are essentially power supplies, and power supply engineers generally assess the quality of a power supply by its efficiency over a load range. Consequently, wireless charging systems are often characterized by their efficiency at a given load current. This industry convention results in misleading and inaccurate characterizations of the performance of a power

supply that is designed specifically as a wireless charging system. It is possible, however, to specify the efficiency of a wireless charger more accurately by comparing the total energy used by the battery over a complete charge cycle, to the energy into the wireless power transmitter over the same charge cycle. Wireless charging for mobile consumer electronics has reached mainstream adoption, and a host of approaches in various stages of development and deployment have emerged that offer a range of benefits. Unfortunately, none of the various approaches are compatible nor are they interoperable. As designers consider incorporating wireless charging technology into their products, they must assess the benefits and tradeoffs of the available or emerging standards. The key tradeoff is efficiency versus a purported benefit. Without an industry consensus on how to measure efficiency, a basic design tradeoff cannot be assessed.

WHY DIFFERENT APPROACHESClosely-coupled configurations make up the lion’s share of the demand for wireless chargers. The classic example of this approach is that of a smartphone sitting in its wireless

Herearethetradeoffsofthevariousmagneticinduction-basedwirelesschargingapproaches.Asofthewritingofthisarticle,therewerenoRezencesystemsavailableinthemarket.ClaimsofRezencebenefitsarederivedfrompublically-availablemarketingmaterials.

Pow

er E

lect

roni

cs

JOHN PERZOWWireless Power Consortium

Whenitcomestoenergyefficiency,therearesignificantdifferences

incompetingwirelesschargingmethods.

WPC_EE_2016_Power_Vs6.MD.indd 52 2/23/16 5:06 PM

TLI Series Rechargeable batteries power the following applications:

• Energy Harversting

• Medical Devices

• Remote Sensors

• Military Equipment

• Safety Equipment

• Handheld Devices

Tadiran Batteries2001 Marcus Ave.Suite 125ELake Success, NY 110421-800-537-1368516-621-4980

www.tadiranbat.com

Consumer grade Lithium-ion

batteries are short lived.

From the moment they’re born, cell phones, laptops, and digital cameras are not long for this earth. So buried inside are the cheapest batteries possible: consumer grade rechargeable lithium cells that last up to 5 years and 500 full recharge cycles. Industrial grade products need the reliable long-term power provided by TLI Series industrial grade Li-ion rechargeable batteries. These rugged little workhorses deliver up to 20 years of operating life with 5,000 full recharge cycles, an unmatched temperature range of -40°to 85°C (storage up to 90°C) and the ability to deliver high pulses (5A for a AA cell).

Tadiran batteries: for when your device is not a toy.

Industrial grade TLI Series lithium ion batteries last up to 20 years.

Tadiran Batteries 2-16 EE.indd 53 2/23/16 4:17 PM


Pow

er E

lect

roni

cs

Measuringefficiencyintoadcloadattheoutputoftherectifiedreceiver(dcinputtodcoutAinthisfigure)canalsobehigh(>85%intheclosely-coupledsystemand75%intheloosely-coupledsystem).Butthisshouldnotbeconstruedasapredictorofoverallsystemefficiency.Measurementsat“dcoutB”canbegoodproxiesforreal-worldefficiencymeasurements,butoutputvoltagesandresistorsneedtobeselectedcarefullytomimicthebattery-chargecycle.

charging stand on a desktop. This mode results in maximum power transfer efficiency, lowest EMI/RFI/EMF and lowest cost. Loosely-coupled configurations are those characterized by a power transmitter mounting under, say, the surface of a desk, countertop or other furniture. The device-to-be-charged sits somewhere on the table surface. Loose coupling approaches are useful in after-market installations of existing furniture and represent about 5 to 10% of the volume demand for wireless chargers. Of course, a loosely-coupled system has a greater charging distance. The cost of the approach is lower efficiency and a higher bill of materials. It is also more difficult to contain the EMI/RFI/EMF in a loosely-coupled system that is operating at high frequency. With these two charging approaches in mind, consider the problem of useable efficiency data. One difficulty is the lack of an agreed measurement method. We have

seen claims of coil-to-coil or “dc-in” to “dc-out” efficiency numbers, but these do not predict overall system performance. For example, both closely-coupled (not operating in resonance) and loosely-coupled (operating in resonance) systems can hit coil-to-coil efficiencies in excess of 90%. This does not imply that the overall system efficiency is 90%, however, so this data point can be misleading. Measuring efficiency into a dc load at the output of the rectified receiver can also result in a high reading (greater than 85% in the closely-coupled system and 75% in the loosely-coupled system). But again, this should not be construed as a predictor of overall system efficiency. Measurements taken at the output of the power supply regulator can be a good proxy for real-world efficiency measurements, but output voltages and resistors used to simulate a real load must be selected to mimic the battery-charge cycle.


Pow

er E

lect

roni

cs

!

! ResearchersatColoradoStateUniversityusedthistopologyinmeasuringthetotalenergyefficiencyofloosely-andclosely-coupledwirelesschargingsystems.Thedc-to-dcpost-regulatorandthebatterychargerusedwerebothswitchingtopologiesandselectedfortheirbest-in-classefficiencyof90%orbetter.A2,100-mAhbatterymodelwasdeveloped,whichdeterminedtheloadprofileasdepictedatright.

!


WIRELESS CHARGING


The Wireless Power Consortium, the organization behind the primary wireless charging standard called Qi (pronounced

“chee”), commissioned a study at Colorado State University that examined the total energy efficiency between two types of wireless chargers: a loosely-coupled system with coils operating in mutual resonance at

ResearchersatColoradoStateUniversitymappeda3Dspatialdependenceofthetransferefficiencyforbothloosely(right)andclosely(left)coupledwirelesschargingsystems.Foreaseofviewing,onlyhalfofthechargingareaisdepictedinthemapoftightlycoupledefficiencies.

6.78 MHz, and a closely-coupled system operating out of resonance at a switching frequency of 110 to 205 kHz. Both cases used a typical cellphone battery-charging configuration. The dc-to-dc post-regulator and the battery charger used were both switching topologies and selected for their best-in-class efficiency of 90% or better. A

2,100-mAh battery model was developed, which determined the load profile. As wireless charging systems let the user randomly place the receiver (phone) on the charging surface, the test also mapped a 3D spatial dependence of the transfer efficiency for each system. In both cases, the receiver was placed at

800.323.2439 • 214.340.0265

. We’ll make YOU a fan.

AC Fans • DC Fans • Fan Trays • Accessoriesorionfans.com

Lowest Cost Louvered Filter Fan Kits

Protection for industrial, electrical and electronic cabinets•Kit includes louvered fan

guard, filter, fan, metal guard & hardware

•IP54, IP55, IP56 rated

•Maintains NEMA 12 rating when used with Orion’s harsh environment fans

Built for Harsh EnvironmentsOrion’s harsh environment AC fans, DC fans and accessories protect against dust, moisture, salt fog, salt spray, temperature changes, and humidity in applications including:

• Automation & instrumentation • Process control • HVAC equipment and blowers • Medical equipment • Military applications • Remote antenna installations • Wind power devices

With the industry’s shortest lead times, Orion rugged fans can be quickly integrated into your design.

Find it. Spec it. Buy it.

OF094 Materials for Power Supplement 1/2 page ad in Design World .indd 1 1/20/15 9:52 AM



Pow

er E

lect

roni

cs

the optimal X-Y position and at a 5-mm coil-to-coil Z-distance. The efficiency measurements were then taken over a typical charge cycle (5 to 95%) of the battery. The results of the study went into a proposal for a technique to accurately assess the power transfer-efficiency of a wireless charging system. The first proposal to come out of this work is that efficiency should be calculated as total energy into the battery divided by total energy into the transmitter over the battery charge cycle. When a wireless power receiver directly drives a resistive load, the power transfer from the input of the transmitter to the resistive load is quite efficient, in some cases exceeding 90% for low-frequency systems. However, tests show the total system efficiency can severely degrade when the receiver is configured for a typical battery-charging application. This is particularly true for high-frequency systems. We attribute this power-transfer degradation to two primary mechanisms: (a) The high-frequency

A plot of the energy used by the battery and energy consumed by the two different wireless charging transmitters provides a feel for relative amounts of energy lost in loosely- and closely-coupled charging systems. The closely-coupled system consumed 50% less energy than the loosely-coupled system over the course of a 2,100-mAh battery charge.

As you might expect, closely-coupled charging systems have a higher power transfer efficiency than loosely-coupled versions.

system requires a relatively high receiver antenna impedance. And the range of receiver output voltages forces the (time-averaged) input impedance of the dc regulator into a point where the wireless transfer process is less efficient. (b) Switching losses in the high-frequency transmitter output transistors. Integrating the energy delivered to the battery and energy put into the transmitter over the time of a charge-cycle enables a real-world assessment of system efficiency. The second proposal is that power-transfer efficiency measurements should be taken as a spatial average. Given the flux map characteristics of a given transmitter, the system efficiency can vary significantly over the charging area or charging volume. To mimic real-world use, efficiency measurements should be

taken in 2-mm increments over the load profile. If appropriate output impedances and voltages are used, it is easy to calculate the total energy for a charge-cycle for each spatial cell. Finally, the tests indicate it’s best to calculate total energy over a battery-charge cycle. The battery load profile describes the time-dependent battery voltage and current during a charge cycle, VB(t) and IB(t) and calculation of energy into the battery, EB, is as follows: At any time t in the charge-cycle, the battery receives an increment of energy from power P in time dt:

dEB (t) = P(t) n dt = VB (t) n IB (t) dt

Therefore, calculation of the energy delivered to the battery over a cycle is the integral of power, which for a discrete-time calculation, becomes a simple sum:

n

cycle

cycle

EB (t) = ∫ d E B (t) =

∑ VB (tn) n IB (tn) Δt

∫ VB (t) n

IB (t) dt =

where Δt is the incremental step and tn ranges over the duration of the charge cycle. In our calculations, Δt is one minute and tn ranges from 1 to 150 (the programmed charge time). In a similar manner we can use efficiency versus load (battery) current, η (IB), to calculate the incremental energy source input dES that provides battery energy dEB as a function of current draw by the battery:

dES (t) = = dEB (t)

η (IB (t))

VB (t) n IB (t)

η (IB (t))dt

Thus charge cycle efficiency is obtained from:

η cycle = = EB ES

∑n VB (tn) n IB (tn) n Δt

∑n [VB (tn) n IB (tn)/η (IB (tn))] n Δt

EFFICIENCY COMPARISONS The point of this study is to make public the tradeoffs between different wireless charging approaches. The closely-coupled Qi system was designed from the start to be as efficient and inexpensive as possible. This approach


WIRELESS CHARGING


Get On-Boardwith Power Resistors

from KOA SpeerMost of you know us as the world leader in thick film resistors. And many are aware ofthe world class standards our Quality 1st program has set for product quality, customersupport and on-time delivery. But did you know we offer a complete line of high performance power resistors that are supported by the same Quality 1st commitment? • Ceramic Fixed Power Leaded Resistors HPC/PCF- 1/2W~ 5W power ratings, resistance range: 3.3Ω ~ 390KΩ, ±10%, ±20% tolerance • High Power Rectangular Wirewound Resistors BGR, BWR, BSR - 1W ~ 40W power ratings, resistance range: 0.1Ω ~ 75KΩ, ±1% ~ ±10% tolerance, uses flame-retardant insulated ceramic case • Miniature Precision Power Wirewound Resistors RW - 0.5W ~ 10W power ratings, resistance range: 0.1 Ω ~ 62KΩ, ±0.5% ~ ±5% tolerance • Power Carbon Film Leaded Resistors SPR(X) - 0.25W ~ 5W power ratings, resistance range: 0.1 Ω ~ 110KΩ, ±1% ~ ±5% tolerance

Call 814-362-5536 or visit KOASpeer.com

Ask for your FREE Samples!

PowerResAd.qxp_Layout 1 2/5/16 11:46 AM Page 1

excels in most consumer applications where the transmitter and receiver lie in close proximity. Examples include charging stands for bedside tables, desktop charging pads, and through-hole chargers in public locations. There are examples where extended distance between the receiver and transmitter is necessary, and it may be appropriate to sacrifice some efficiency in these applications. After-market furniture installations are the typical example. The WPC approach has always been to give OEMs the ability to use whichever approach is most appropriate while maintaining full interoperability between Qi devices. When using the charge-cycle efficiency calculation described above, it is possible to plot total energy versus time over the battery-charge cycle for each spatial cell in the charging area or charging volume. The closely-coupled system consumed 50% less energy than the loosely-coupled system over the course of a 2,100-mAh battery charge. The high-frequency, loosely-coupled and the low-frequency, closely-coupled systems have different loss profiles. Although the loosely-coupled system we tested uses gallium-nitride output transistors and zero-voltage switching, the high operating frequency of the architecture results in significant transmitter switching loss (greater than 800 mW). The value of operating in resonance, however, becomes evident where even at 20 mm, the loosely-coupled system shows relatively little coupling loss. Another observed loss factor would best be described as a maximum power-point transfer (MPPT) problem. The characteristic impedance of the loosely-coupled receiver antenna (about 24 Ω) is not well-matched to the load impedance (3.5 to 32 Ω) of the battery. Most systems will optimize this power-point for the maximum load condition (4.2 V, 1. 2 A, 3.5 Ω), but the total energy used is impacted by the significant time spent in lower load, higher impedance output conditions. Armed with real data, it’s easy to see that it does not make sense to use a less efficient, high-frequency, loosely-coupled system when a closely-coupled approach

will do the job. Conversely, it is appropriate at times to sacrifice some efficiency when a larger Z-distance is needed. That is why a wireless power specification that meets all market requirements must support both closely- and loosely-coupled approaches.

WIRELESS CHARGING

REFERENCES

Wireless Power Consortium wirelesspowerconsortium.com

57DESIGN WORLD — EE Network



Pow

er E

lect

roni

csWHITE PAPER

Digital Inrush Current Controller Reference Design

T he Digital Inrush Current Controller reference design combines IXYS’ Digital Power Control technology with the capabilities of Zilog’s

8-bit Z8F3281 microcontroller, a member of the Z8 Encore! XP F6482 series of MCUs. It employs a unique approach to control inrush current in ac-to-dc rectifiers or ac-to-dc converters. The objective of this reference design is twofold: to highlight the advantages of digital control that overcomes many of the shortcomings of current technology, and to enhance interest in digital control of high power converters, potentially stimulating the development of next generation converters. Digital control allows for distinctive solutions to control inrush current in a typical ac-to-dc rectifier with capacitive load by limiting the capacitor pre-charge current to a predetermined value at each half sine wave cycle. This capacitor charge is spread over a number of cycles until the capacitor is charged to a peak value of ac voltage source. The capacitor is charged according to a time-dependent pulse train. Pulses are designed to provide substantially equal voltage increments applied to the capacitor to maintain peak charging current at approximately

the same value at each cycle. The number of cycles depends on capacitor value and charge current. For a given capacitor value, which is selected based on the desired amplitude of ripples, the charge current is a function of the number of pulses and their timing position with respect to the rectified sine wave. This reference design features programmable overload protection and the Power Good status signal. It is not sensitive to power outages, brownouts and ambient temperature variations. This reference design has the ability to operate within an input voltage range of 80 to 240 Vac and load current up to 3 A. The entire operating process and essential values are fully programmable. The controller may be programmed to 50 or 60 Hz, or any other line input frequency operation. This Digital Inrush Current Controller reference design is valuable for high-power loads with tens of amps of current in normal mode of operation. It allows users to optimize performance, maximize efficiency across the load range, and reduce the design time to market. IXYS power components handle the pre-charge of load capacitors at these values while limiting inrush current to controlled values.

Figure 1. Functional block diagram of the digital inrush controller


IXYS_EE_2016_Power_Vs4.MD.indd 58 2/23/16 3:10 PM

WHITE PAPER


WHITE PAPER

A basic example of the Digital Inrush Current Controller comprises components that include typical power components of an ac-to-dc rectifier (diode bridge, inductor and bulk capacitor), a switch to commutate capacitor pre-charging current, another switch to connect and disconnect the load, and a digital control module based on Zilog’s Z8F3281 MCU.

THEORY OF OPERATIONThe Digital Inrush Control approach aims to provide charge to a bulk capacitor in substantially equal increments. This is accomplished by providing control pulses to Sw1, as shown in Figure 1, resulting in equal increments of a voltage applied to the bulk capacitor. It is possible to apply this charge on a cycle-by-cycle basis considering a cycle is half of a line voltage sine wave. For example, we can assign N cycles for the inrush control operation and then split the normalized amplitude of the sine wave cycle to N segments with increment of 1/N, as shown in Figure 2. During Cycle 1, Sw1 is in the on state (conducting) from time t1 to T, as shown in Figure 2. The voltage across the capacitor increases to a voltage proportional to the normalized value 1/N. During this period, the charging current rise follows the LC resonant behavior, as shown in Figure 3 (green line). The current rises until the capacitor voltage reaches the input voltage, excluding voltage dropouts. The current continues its resonant behavior as long as Sw1 is on. No further oscillation occurs because the input voltage drops below the voltage on the capacitor, switching Sw1 to the off position (not conducting). The capacitor remains pre-charged to the voltage proportional to 1/N. In Cycle 2, capacitor C is pre-charged by another voltage increment 1/N, in a process similar to Cycle 1. Capacitor C is charged during N cycles to a voltage value proportional to the peak voltage value of the input power line.

COMPOSING TIMING FOR DIGITAL INRUSH CONTROL A simplified timing diagram for inrush control is shown in Figure 2, in which the voltage increment for each cycle is defined by the number of cycles (N). The capacitor’s charging current is proportional to the voltage increment, 1/N. Therefore, the number of cycles (N) is the variable used to control peak inrush current. Another variable to control inrush current is the LC time constant. The value of capacitor C depends on the desired ripple value. After selecting the value of capacitor C, the designer can reduce peak inrush current by increasing inductance L. If there are physical limits to the L value, the number of cycles (N) should be used to set the required peak current. Turn-on time for switch Sw1 should be defined for each active cycle. Assuming that delay from zero crossing (point 0 in Figure 2) to the beginning of turning Sw1 on, t4, is T_off, an active time to keep Sw1 on is T_on, and cycle duration is T.

T_on for each occurrence i is defined as geometrical transform:

T_on(i) = T

π/2asin (i/N), where i = 1 ... N (1)

where i = 1… N. The period (T) is measured by the MCU at initialization. Values for T_on are determined by (1) and stored in memory. Values for T_off are derived by firmware according to:

T_off = T – T_on (2)

Figure 4 illustrates how T_on values are defined for 16 cycles used in this reference design for 120 V at 60 Hz. In Figure 4, the blue line (rectified power line voltage) is shown for reference. The magenta line represents actual T_on time value in microseconds for each cycle, and the yellow line indicates T_on pulse positioned relatively to rectified power line voltage.

Figure 2. Digital inrush control timing


IXYS_EE_2016_Power_Vs4.MD.indd 59 2/24/16 10:24 AM


Pow

er E

lect

roni

csWHITE PAPER

Figure 4 conceptually illustrates the algorithm functionality as executed by the Z8F3281 MCU. The timing counter (red line) corresponds to time at any given moment of discrete time base provided by the internal clock. The counter first counts until the T_off value, represented by the green line. When the counter reaches T_off value, it initiates the T_on pulse (yellow line) which continues until the counter reaches T_on value (magenta line). Figure 5 displays the timing position and amplitude of the capacitor current (green line) with respect to T_on pulses. A single current pulse is produced by the inrush controller during a cycle because the input voltage drops below the capacitor voltage and the input power line is isolated from the rest of the circuitry through the diode bridge after the capacitor charge is completed. The inductor discharges into the capacitor, and Sw1 switch is turned off (not conducting) at the end of the cycle. This reference design also includes a load on/off switch, Sw2, overload protection, and power-good status output to display the capability of IXYS power components. Sw2 activation is programmable and is enabled after the capacitor C pre-charge completes. The Sw2 switch activates at zero crossing on the next cycle after the capacitor is pre-charged. In the simulation shown in Figure 5, the Sw2 switch is activated at time stamp 0.066 msec, when the capacitor current shows up as negative, because the current

Figure5.CapacitorCpre-charging:Blue—rectifiedpowerlinevoltage;Red—voltageoncapacitorCwithrespecttocommonground;Yellow—drivertoSw1;Green—capacitorcurrent

Figure4.T_ontiminggeneration:Blue—rectifiedpowerlinevoltage;Red—fullcycleperiodtimingcounter;Yellow—drivertoSw1;Green—timeofftoSw1;Magenta—timeontoSw1;White—periodT

Figure 3. Capacitor C pre-charging duringfirsttwocycles


WHITE PAPER


WHITE PAPER

is sourced from the capacitor. Sw2 switch activation can be programmed to any time stamp point, depending on customer requirements. The overload protection feature protects a device from damage in case of current overload. The overload threshold is programmable and is set to 3.5 A in this reference design. If overload is detected by a comparator, the MCU disconnects the load by turning the Sw2 and Sw1 switches off, preventing extra current from going into the load Overload protection can be programmed either to immediately shut down the device and wait for user input, or to allow the device to restart after the short circuit is removed. In this case, the device is allowed to restart for a predetermined number of short circuit occurrences if the short circuit occurrences repeat. In this mode of operation, the delay between restarts and the number of restarts is programmable. This reference design sets the delay time to 1.5 sec and the number of restarts to four. Power-good status activation is also programmable. This reference design delays activation by two cycles after the capacitor’s pre-charge completes. The power-good status is not set if an overload is detected. The Digital Inrush Current Controller implements the MCU module as an add-on device. The module consists of a connector for programming the microcontroller. The MCU module is powered by an auxiliary power supply of 3.3 V for the MCU and 12 V for the gate driver applied to the J4 connector on the

Figure 7. Digital inrush controller setup

Figure 6: (Left) MCU module (Right) Main power board with MCU module

main power board. The main power board is a two-layer surface-mount device that provides easy access to test points. Diode bridge BR1 and MOSFETs Q1 and Q2 mount on small heat sinks. Power dissipated on these heat sinks is less than 5 W at a 375-W power output. This board may be powered from a 50- or 60-Hz ac source.

TESTING THE REFERENCE DESIGNFor purposes of testing the design, the ac line input is fed through 0.5-kW isolation transformers. The load is designed to consume 2.5 A during normal operation. To test overload conditions, an additional load was used to provide 3.5 A. An instantaneous connection of additional load was enough to trigger overload protection. Continuous overload results in multiple attempts to restart the device with immediate interruption. Auxiliary power supply should be turned on only after the ac power is

on. Following a power-on reset and initialization, the MCU analyzes the power line, sets an appropriate timing, and begins pre-charging the bulk capacitor. Actual waveforms taken from a scope at normal operation are in Figure 8. The inrush current (top blue line) is limited to 10 A. The yellow line shows the signal at the Sw1 gate. After the inrush procedure is completed, the gate is set to a high level to allow Sw1 to continue conducting. One cycle later, the load is connected and load voltage rises from zero to the level on the pre-charged capacitor. After the load is connected, a slight drop in the rectified voltage occurs due to the limited power capability of the isolation transformer. Tests were run to highlight the performance of increasing number of inrush control cycles at the same power conditions, Here, inrush current measurements were taken with the controller reprogrammed to 16 cycles




Pow

er E

lect

roni

csWHITE PAPER

instead of the original 8 cycles. As a result, the inrush current dropped two times, as shown in Figure 9. Normal operation starts with pre-charging the bulk capacitor. A scope snapshot of the startup is shown in Figure 10 with a horizontal scale of one second. The yellow line indicates that the gate driver state is wide because of the low graphic resolution for 12 cycles of pre-charge. After the inrush control sequence completes, the load is connected, indicated by the red line going up. The power-good status (blue line) goes up as well. Continuous overload is applied at 2.3 sec. The load is disconnected and the power-good status goes low. An attempt

Figure 8. Scope snapshot of digital inrush current control: Blue—power line current (10 A/div); Red—load voltage (50 V/div); Green—rectified input voltage

(50 V/div); Yellow—Sw1 commutation signal

Figure 9. Scope snapshot of inrush pulses with N increased to 16: Blue—power line current (10 A/div); Red—load voltage (50 V/div); Green—rectified input voltage (50 V/div); Yellow—driver to Sw1

to restart the device is initiated in 1.5-sec intervals. The inrush sequence takes place again, and the load is connected. However, the overload is sensed right away, the load is disconnected, and the power-good status continues to stay low. Another attempt to restart takes place after 1.5 sec and the load stays connected for about one second while there is no overload condition. TheefficiencyoftheDigitalInrushController is estimated starting after the

dioderectifiertotheload.Ataloadpowerof375 W, the power loss is 2.1 W, which equates toanefficiencyof99.47%. Thisreferencedesigncanbeconfiguredfor different parameters, such as input voltage and frequency range; overload threshold; overload recovery time; number of overload events before shutdown; time position for power-good status; and time position to turnontheload.Thefirmwarethatcontrolsthe operations of this reference design was developedusingZilog’sdevelopmentstudioII

Figure 10. Scope snapshot to show performance during overload and restart: Blue—power-good status; Red—load voltage (50 V/div); Yellow—driver to Sw1


WHITE PAPER


WHITE PAPER


Figure 11. Schematic diagram of digital inrush controller main board

Figure 12. Schematic diagram of digital control module

(ZDS II – Z8 Encore! version 5.2.0). Testingofthisdesignconfirmedthat the inrush current is limited to a predefinedvalueandtheperformanceofthe limiter is quite close to the simulation results.Themeasuredefficiencyoftheinrush control path is 99.5%. This device is capable of working with a range of input voltages from 80 to 240 V. The tested powerlinefrequencyrangewas50to60Hz. A dedicated control pulse train was developedforeachpowerlinefrequency. To work with higher power line voltage, a longer control pulse train needs to be programmed. For instance, raising line voltage from 110 to 220 V required twice as much pre-charging time to have the same peak inrush current.

Overload protection is based on continuousmonitoringofdynamiccurrent from the bulk capacitor. In case of overload, the current drawn from the capacitorinstantlyincreasesandtriplesthecomparator,therebyinitiatingsystemoverload mode. The overload current threshold, number of overload instances, and period between overload events are all programmable. The option to turn the load on/off aids the overload mode of operation bydisconnectingtheload.Power-goodstatus is not available in overload conditions. The overload protection device is not sensitive to power interruptions, brownouts and temperature variations. All in all, an innovative current measurement algorithm lets this controller

REFERENCESThesourcecodefileassociatedwiththisreferencedesign,IXRD1001-SC01.zip,isavailablefreefordownloadfromtheIXYSPowerandZilog websites. This source code is tested with ZDS II – Z8 Encore! version 5.2.0. SubsequentreleasesofZDSIImayrequireyoutomodifythecodesuppliedwiththis reference design:ixyspower.com zilog.com

Documents associated with this reference design can be obtained from the IXYS or Zilog websites: Digital Inrush Controller Reference Design document, http://tinyurl.com/jxc4qox F6482 Series General-Purpose Flash Microcontroller Product Specification, http://tinyurl.com/oauvfvc F6482 Series Development Kit User Manual, http://tinyurl.com/jsrjkdn F6482 Series API Programmer’s Reference Manual, http://tinyurl.com/nqj7akg

use common input and load grounds. Users can optimize the device for a wide range of input voltages and frequencies. The Digital Inrush Controller can be used aspartofanac-to-dcrectifierorcanbeexpanded to higher-level devices such as a PFCconverter.Digitalcontrolcanbeusedto build a user interface that would allow users to change device parameters, gather statistics, add a communication interface, remotelymonitorperformance,orchangeparameters. EE


Applied Power Electronics Conference and Exposition ....................................................................... 46apec-conf.org

Behlman Electronics, Orbit Power Group .......................... 1vmevpx.com

Carlo Gavazzi ...................................................................... 51GavazziOnline.com

Chroma Systems Solutions ...............................................BCchromausa.com

Coilcraft ............................................................................... 11coilcraft.com

CTS Electronic Components ............................................... 9ctscorp.com

CUI, Inc .............................................................................. IBCcui.com

Digi-Key Corporation ............................................ IFC,Coverdigikey.com

Hirose Electric U.S.A. .......................................................... 43hirose-connectors.com

IXYS ...................................................................................... 34ixys.com

Keysight Technology, Inc. .................................................... 5keysight.com

Keystone Electronics Corp ................................................. 15keyelco.com

KOA Speer Electronics, Inc. ............................................... 57KOASpeer.com

Follow the whole team on twitter @DesignWorld


Pow

er E

lect

roni

cs AD INDEX

Master Bond ........................................................................ 47masterbond.com

Mean Well USA, Inc. ............................................................. 7meanwellusa.com/dw

Memory Protection Devices, Inc. ........................................ 3batteryholders.com

Microchip ............................................................................. 37microchip.com

Orion Fans ........................................................................... 55orionfans.com

Pico Electronics ................................................................... 27picoelectronics.com

Pioneer Magnetics ......................................................... 44,45pioneer-mag.com

Renco Electronics ............................................................... 31RencoUSA.com

Richardson Electronics ....................................................... 24Rellpower.com

Spontaneous Materials ...................................................... 47spontaneousmaterials.com

Tadiran Batteries ................................................................. 53tadiranbat.com

TDK Epcos ........................................................................... 35tdk.comepcos.com

Teledyne LeCroy ................................................................. 33teledynelecroy.com

www.powerelectronictips.com

Ad Index_EE_ 2-16_Vs1.indd 64 2/24/16 2:30 PM

www.cui.com/power-of-things

CONNECT

HEAL

BUILD

DELIVER

The Power of ThingsTechnology is transforming the world around us at a rapid pace. While you are designing the products that will save lives and bring global communities together, we are developing the power systems to support your most demanding applications. Our power expertise and collaborative approach are in place to support you while you shape the future of technology.

The power to .

TM

CUI Inc 2-16 EE.indd 1 2/23/16 12:53 PM

Instruments

AC Power SourcesRegenerative Grid SimulatorsProgrammable DC Power SuppliesAC & DC Electronic LoadsPower MetersMultimeters

Hipot Testers and AnalyzersWound Component TestersLCR MetersMilliohm MetersTEC ControllersThermal Data Loggers

Automated Test Systems

BatteryEV/EVSEPV InverterPower ConversionMedical DeviceLED Lighting and Driver

Chroma 2-16 EE.indd 1 2/23/16 12:54 PM