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1Topologies and OperatingPrinciples of Basic
Full-BridgeConverters
1.1 Introduction
1.1.1 Development Trends of Power Electronics Technology
High-frequency switching is one of the most important features
of power electronicstechnology, enabling power electronics
converters to meet the required specificationsand performance.
High-frequency power devices and components, including
magneticdevices and capacitors, are the basic elements of
high-frequency power electronics.Metal-oxide-semiconductor
field-effect transistors (MOSFETs) and insulated-gatebipolar
transistors (IGBTs) have become the dominant choices for the
implementationof power switches, and MOSFETs with low gate charges
and low junction capacitorsfurther boost the development of
high-frequency power electronics. Recently, therehas been
significant progress in the development of silicon carbide
(SiC)-based powerdevices, including SiC diodes [1], SiC MOSFETs,
and SiC IGBTs [2], and SiC-basedcommercial products have become
strong competitors to Si-based fast-recoverydiodes and MOSFETs in
medium power conversion applications. Gallium nitride(GaN) power
devices have drawn attention for achieving ultra-fast switching.
Fur-thermore, recent progress in the development of amorphous,
microcrystalline coresand high-frequency ferrites has been
significant.
Circuit topology is another important aspect of high-frequency
power electronics.Switching losses in switching devices have been
drastically reduced through thesystematic development of resonant
converters [3, 4], quasi-resonant converters[5], and multi-resonant
converters [5, 6], zero-voltage-switching (ZVS)
pulse-widthmodulation (PWM) and zero-current-switching (ZCS) PWM
converters [7, 8],
Soft-Switching PWM Full-Bridge Converters: Topologies, Control,
and Design, First Edition. Xinbo Ruan.© 2014 Science Press. All
rights reserved. Published 2014 by John Wiley & Sons Singapore
Pte. Ltd.
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2 Soft-Switching PWM Full-Bridge Converters
zero-voltage-transition (ZVT) and zero-current-transition (ZCT)
converters [9, 10],and resonant dc link inverters (RDCLIs) [11]
that partially or fully achieve ZVSand ZCS. As a result, the
operating switching frequency has increased by an orderof magnitude
and more. High-frequency switching is also the key contributor
tominiaturization and modularization, due to the significant gain
in efficiency of powerconversion that it can offer, in addition to
the high insulation and high thermalconductivity of the structure
employed. Therefore, the success of high-frequencypower electronics
can be attributed to the advent of high-frequency power devicesand
components, soft-switching technologies, mechanical structures,
materials, andrelated technologies.
1.1.2 Classification and Requirements of Power Electronics
Converters
Power electronics converters are a family of electrical circuits
that convert electricalenergy from one level of
voltage/current/frequency to another using semiconductordevices,
passive components, and advanced control methods. According to the
formof conversion, power electronics converters can be classified
into four different types[12]: (i) dc–dc converters, which convert
a dc input voltage into a dc output volt-age of a different
magnitude and possibly opposite polarity, or with galvanic
isolationof the input and output ground references; (ii) dc–ac
inverters, which transform adc input voltage into an ac output
voltage of controllable magnitude and frequency;(iii) ac–dc
rectifiers, which convert an ac input voltage into a dc output
voltage, andare capable of controlling the dc output voltage and/or
ac input current waveform;and (iv) ac–ac cycloconverters, which
convert an ac input voltage into an ac outputvoltage of
controllable magnitude and frequency. These four kinds of power
electron-ics converter can be unidirectional or bidirectional. The
unidirectional ones can onlyconvert electrical power from a defined
input terminal to a defined output terminal,while the bidirectional
ones can convert electrical power in either direction betweentwo
defined terminals.
The primary objective of power electronics converters is to meet
the correspond-ing electrical specifications and regulatory
requirements. While meeting the electricalspecifications, a power
electronics converter should idealy achieve high efficiency,high
power density, high reliability, and low cost. High efficiency not
only leadsto energy saving but also results in a reduced heat
dissipation requirement. Highpower density means compact size at
the required output power, which is very impor-tant for aerospace
applications and compact portable appliances. Power convertersof
high reliability are more competitive in the commercial market and
can be essen-tial for critical applications that require operation
under adverse working conditionsand long mean-time-between-failure
(MTBF). Moreover, the overall cost is a key fac-tor for commercial
power supply applications. Power electronics converters are
oftenrequired to have good maintainability, including reduced
technical requirements forrepair personnel and shortened repair
times.
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Topologies and Operating Principles of Basic Full-Bridge
Converters 3
1.1.3 Classification and Characterization of dc–dc
Converters
Dc–dc converters are an important kind of power electronics
converter. With thedevelopment of power electronics technology,
computer science and technology, andinformation technology, dc–dc
converter-based switching-mode power supplies havebeen widely used.
The dc–dc converter is thus the key basic building block in
powerelectronics and has attracted a great deal of research
attention in the past few decades.
According to the presence of galvanic isolation between input
and output,dc–dc converters can be divided into two classes:
non-isolated and isolated. Basicnon-isolated dc–dc converters
include buck, boost, buck–boost, Cuk, Zeta, andsingle-ended primary
inductor converter (SEPIC) converters. Other examples includethe
dual-switch buck–boost converter, the full-bridge converter, and so
on.
Isolated dc–dc converters are derived from non-isolated dc–dc
converters by incor-porating transformers and output rectifier
circuits. Isolated buck-derived convertersinclude forward,
push–pull, half-bridge, and full-bridge. The forward converters
canbe single-switch or dual-switch versions. Isolated boost-derived
converters includepush–pull, half-bridge, and full-bridge versions.
Isolated buck-boost converters areflyback converters, which can
also be single-switch or dual-switch versions. Cuk,Zata, and SEPIC
converters also have isolated versions.
With the power devices having the same voltage and current
ratings, the outputpower of the dc–dc converter is proportional to
the number of power switches.Thus, the isolated dc–dc converter
with two power switches (dual-switch forward,push–pull, and
half-bridge) can handle twice as much power as an isolated
dc–dcconverter with only one power switch, such as the
single-switch forward converter,and only half as much power as a
converter with four power switches, such as thefull-bridge
converter. Thus, the full-bridge converter can handle the largest
poweramong all the isolated buck-derived converters, and it has
been widely used inhigh-input-voltage and medium- to high-power
conversion applications.
Resonant converters, quasi-resonant converters, and
multi-resonant converters canachieve ZVS or ZCS for power switches
without using additional auxiliary powerswitches. However, such
soft-switching converters are different from the conventionalPWM
converters and they have the disadvantages of high voltage/current
stress, largecirculating energy, and variable switching frequency.
For ZVS-PWM and ZCS-PWMconverters, the operating frequency is
constant and additional auxiliary powerswitches are needed. Also,
the voltage/current stress of both the main and auxiliarypower
switches is relatively high. For ZVT and ZCT converters, the
operatingfrequency is also fixed. However, the additional auxiliary
power switches are usedonly for achieving ZVT or ZCT for the main
power switches and do not contribute topower processing. Among the
isolated buck-derived converters, the converters withtwo or four
power switches can achieve ZVT or ZCT without additional
auxiliaryswitches, provided an appropriate control scheme is
employed. Such simplicitymakes them popular choices in practical
applications. This book intends to describesystematically the
soft-switching techniques for isolated buck-derived full-bridge
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4 Soft-Switching PWM Full-Bridge Converters
converters. For brevity of illustration, the isolated
buck-derived full-bridge converteris simply called a “full-bridge
converter.”
1.2 Isolated Buck-Derived Converters
In order to give some insights into the characteristics of the
isolated buck-derivedconverters and to reveal the relationships
among them, this section begins by deriv-ing the forward converter
from the basic buck converter. It then goes on to derivethe
dual-switch forward converter, push–pull converter, half-bridge
converter, andfull-bridge converter.
1.2.1 Forward Converter
1.2.1.1 Derivation of a Single-Switch Forward Converter
The buck converter is the most basic of the dc–dc converters. It
is shown inFigure 1.1a, where Vin is the input dc voltage, Q is the
power switch, DFW is the free-wheeling diode, and Lf and Cf are the
output filter inductor and output filter capacitor,respectively. In
order to achieve galvanic isolation, the transformer Tr is
insertedbetween Q and DFW, as shown in Figure 1.1b. The primary and
secondary windingturns of Tr are Np and Ns, respectively, and the
corresponding turns ratio is K=Np/Ns.When Q conducts, the input
voltage Vin is applied on the transformer primarywinding. The
transformer is thus magnetized and the magnetizing flux 𝜙m
increases.
Vin
Vin
VinVinQ
DFW DFW
DFW
DFW
Cf
Cf
Cf
Cf
Lf Lf
Lf
Lf
DR
Dr
DR
RLd
RLd
RLd
RLd
Vo
Vo
Vo
VoQ * *vp vsec
vp
vp
vsec
vsec
vrect
vrect
vrect
Np
Np
Np
Ns
Ns
Nr
Ns
Tr
Tr
Tr
(a) Buck converter (b) Transformer inserted
Q * *Magnetic
Reset Circuit
(c) Magnetic core reset circuit and rectifier diode included
* *
Q
*
(d) Single-switch forward converter with magnetic reset
winding
–– –
+
–
+
–
+
–
+
–
+
–
+
–
+
–
+
–
+
+ +
–
+
–
+
Figure 1.1 Derivation of the single-switch forward converter
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Topologies and Operating Principles of Basic Full-Bridge
Converters 5
(a) Without magnetic core reset circuit (b) With magnetic core
reset circuit
ϕmt0
0 Ts 2Ts
Qt
0 t
vrect
vp
ϕm
Vin/K
0 Ts 2Ts
t0
Qt
0 t
vrect
vp
/KVin
Figure 1.2 Waveforms of the primary voltage and magnetizing flux
of the transformer
When Q is turned off, the filter inductor current is
freewheeling through diode DFW.The conducting diode short-circuits
the transformer secondary winding, forcing theprimary winding
voltage to zero and thus keeping 𝜙m unchanged. Therefore, in
aswitching period, 𝜙m has a net increase, and as time elapses, the
magnetic core willsaturate, leading to destruction of the power
switches. The waveforms of the primaryvoltage vp and magnetizing
flux 𝜙m of the transformer are sketched in Figure 1.2a.
In order to avoid saturating the transformer, the magnetizing
flux should be resetbefore the end of each switching period. Thus,
a magnetic core reset circuit is manda-tory. This circuit applies a
negative voltage across the primary winding of the trans-former
when Q is turned off, as shown in the shaded area in Figure 1.2b.
However, thisnegative voltage will be reflected to the secondary
winding and will force the free-wheeling diode DFW to conduct,
shorting the secondary winding. In order to avoidshort-circuiting
the transformer, a diode DR can be inserted in series with the
sec-ondary winding, as shown in Figure 1.1c. Thus, a magnetic core
reset circuit can beformed by a reset winding Nr and a reset diode
Dr. Exchanging the positions of powerswitch Q and the transformer
primary winding, the basic single-switch forward con-verter is
formed, as shown in Figure 1.1d. In practical applications, the
turns of thereset winding and primary winding are usually designed
to be equal, and thus the volt-age stress of the power switch is
2Vin and its maximum duty cycle is limited to 0.5, inorder to
achieve magnetic core reset of the transformer.
1.2.1.2 Derivation of a Dual-Switch Forward Converter
As the power switch sustains twice the input voltage, the
single-switch forwardconverter is suitable for low-input-voltage
applications. For high-input-voltage appli-cations, it may be
difficult to find an appropriate power switch with such a
highvoltage rating. For example, with a single-phase input ac
voltage of 220 V± 20%, therectified input voltage with power factor
correction is 380 V. The voltage stress of thepower switch is 760 V
and a power switch with voltage rating of 1000 V is
required.MOSFETs with such a high voltage rating have poor
performance at high frequencies,
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6 Soft-Switching PWM Full-Bridge Converters
and the drain-source conduction resistor Rds(on) is relatively
large. Although IGBTscan be adopted, the switching frequency is
limited to tens of kilohertz due to the pres-ence of the current
tail. At high frequencies, the turn-off loss will be relatively
high,causing degradation in efficiency.
For the sake of employing the available power switches with
better performance,it is desirable to reduce the voltage stress of
the power switch. As indicated before,when the turns of primary
winding and reset winding are equal, the power switch ofa
single-switch forward converter must withstand twice the input
voltage. If powerswitch Q is replaced by two power switches Q1 and
Q2, as shown in Figure 1.3a, thevoltage stress of Q1 and Q2 will be
the input voltage Vin. Exchanging the positionsof Q1 and the
primary winding leads to the circuit shown in Figure 1.3b. In order
toensure the voltage stress of Q1 and Q2 is Vin, a diode D2 is
inserted between point Aand the negative rail and a diode D1
between point B and the positive rail, as shownin Figure 1.3c. When
both Q1 and Q2 are turned off, the transformer is magneticallyreset
through the reset winding Nr and the transformer primary voltage
vAB is −Vin.In fact, the transformer can be magnetically reset
through the primary winding viaD1 and D2. Therefore, the path
constituting reset winding Nr and reset diode Dr isredundant and
can be removed. The simplified circuit is redrawn in Figure 1.3d,
givingthe well-known dual-switch forward converter. In this
circuit, the voltage stress of the
(a)
(c)
(b)
(d)
* *
Tr
Vin
DRQ1D1
D2 Q2
Lf RLd
Np NsDFW Cf
Vo–
+
* *
*
Vin
DR
Q1
Q2
Dr
Lf RLd
NpNr
NsDFW Cf
Vo–
+Tr
* **
NrA
B
Vin
DRQ1
D1
D2 Q2
Dr
Lf RLd
Np NsDFW Cf
Vo
–
+Tr
* **
A
B
Vin
DR
Dr
Q1
Q2
Lf RLd
Np
Nr
NsDFW Cf
Vo–
+Tr
Figure 1.3 Derivation of the dual-switch forward converter: (a)
using two power switchesin place of one power switch, (b)
exchanging the positions of Q1 and the transformer primarywinding,
(c) adding two diodes to ensure the two power switches sustain the
input voltage, and(d) final configuration of the dual-switch
forward converter
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Topologies and Operating Principles of Basic Full-Bridge
Converters 7
two power switches is Vin, which is half that of the
single-switch forward converter.Moreover, D1 and D2 are the reset
diodes, and also provide the path for regenerationof the leakage
inductor energy to the input voltage when Q1 and Q2 are turned
off.
1.2.2 Push–Pull Converter
For the single-switch forward converter, when the primary
winding and reset wind-ing have the same number of turns, the duty
cycle should be bounded below 0.5 toensure magnetic core reset.
Therefore, the magnitude of the secondary rectified volt-age should
be larger than twice the output voltage. This secondary rectified
voltagecontains a large amount of harmonics, necessitating the use
of a large output inductor.In order to reduce the magnitude of the
secondary rectified voltage and hence theoutput filter inductor, we
can use two single-switch forward converters connected inparallel
at the input side and the secondary rectifier, sharing the
freewheeling diodeand the output filter, as shown in Figure 1.4a.
Note that the two forward converters
V
(c)
(a) (b)
in
* *
*
* **
DR1 DFW
DR2
Dr1
Dr2
Ns1Np1
Ns2
Nr1
Nr2
Tr1
Tr2
Lf RLd
Cf
Q1
Q2
–
+
–
+
–
+
–
+
–
+
–
+
vp1
vp2
vsec1
vsec2
vrect Vo
Vin
* *Tr
*
* **
DFW
DR2
Dr1 D1
D2Dr2
Ns2Np2Np2
Nr1
Nr2
Cf
Q1
Q2
–
+
–
+
–
+
––
+
–
+
DR1Ns1Np1Lf RLd
+
vp1
vp2
vsec1
vsec2
vrect Vo
Vin
*
* *
*
iDR1 iLf
iDR2
DFW
DR2D1 D2
Ns1
Ns2
Np1 Np2
Tr
Cf
Q1 Q2
–
+
–
+
DR1 Lf RLd
vrect Vo
Figure 1.4 Derivation of the push–pull converter: (a) two
single-switch forward convertersconnected in parallel at the input
side and the secondary rectifier, sharing the freewheelingdiode and
the output filter, (b) the two transformers sharing a magnetic
core, and (c) finalconfiguration of the push–pull converter
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8 Soft-Switching PWM Full-Bridge Converters
0 Ts 2TsTs/2 3Ts/2
Vin/Kvrect
t0
ϕm2
vp2
0
Vin
t
vp1
ϕm1t0
Vin
Vin
Vin
tQ1 Q2 Q2Q1 Q1
Figure 1.5 Key waveforms of two interleaved forward
converters
are required to operate in an interleaving manner; that is,
power switches Q1 and Q2operate at the same switching frequency
with a time difference of half the switchingperiod, as shown in
Figure 1.5.
Referring to Figure 1.4a, if we let the two transformers share a
magnetic core andconnect an antiparallel diode D2 across Q2, as
shown in Figure 1.4b, the primary wind-ing Np2 and diode D2 can
provide the magnetic core reset function for the transformerwhen Q1
is turned off. Likewise, by connecting an antiparallel diode D1
across Q1,the transformer core can be magnetically reset through
the primary winding Np1 andD1 when Q2 is off. This makes the two
core reset circuits redundant and they can beremoved. The
simplified circuit is re-sketched in Figure 1.4c, showing what is
usuallyreferred to as the push–pull converter.
It should be noted that the transformer of the push–pull
converter cannot be magnet-ically reset when Q1 or Q2 is off
because the current of the output filter inductor will
befreewheeling through diode DFW, which clamps all of the
transformer winding at 0 V.When Q1 conducts, the transformer is
positively magnetized; when Q2 conducts, thetransformer is
negatively magnetized; and when both Q1 and Q2 are off, the
voltagesacross the transformer windings are zero and the magnetic
flux of the transformerremains unchanged. Figure 1.6 shows the key
waveforms of the push–pull converter.For the forward converter,
including the single-switch and dual-switch versions,
themagnetizing current can only flow in a single direction, as when
the magnetizing cur-rent decays to zero, it cannot flow in the
reverse direction. For the push–pull converter,
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Topologies and Operating Principles of Basic Full-Bridge
Converters 9
vp1, −vp2
ϕmt0
0 Ts 2TsTs/2 3Ts/2
Vin
Vin
Vin/Kvrect
t0
Q1 Q1 Q1Q2 Q2 t
Figure 1.6 Key waveforms of the push–pull converter
the magnetizing current of the transformer flows
bidirectionally. If the flux swing isconstrained by the core loss
rather than by the saturation flux density, the utilizationof the
transformer of the forward converter is the same as that of the
push–pullconverter.
As in the single-switch forward converter, the two switches of
the push–pull con-verter must withstand a voltage of 2Vin. Since
the push–pull converter is equivalent tothe interleaved parallel
connection of two single-switch forward converters, the
ripplefrequency of the secondary rectified voltage vrect is twice
the switching frequency andits duty cycle can reach unity, as shown
in Figure 1.6. With the same input and outputvoltages, the required
magnitude of vrect of the push–pull converter is only half thatof
the forward converter. Thus, the primary-to-secondary-winding-turns
ratio of thepush–pull converter is twice that of the forward
converter.
Referring to Figure 1.4c, when both switches are off, the output
filter inductor cur-rent can freewheel through DFW or flow through
the two secondary windings viarectifier diodes DR1 and DR2.
Therefore, DFW is redundant and can be removed. WhenDFW is removed,
we get:
iDR1 + iDR2 = iLf (1.1)
For an ideal transformer, the magnetizing current is zero; that
is:
iDR1 − iDR2 = 0 (1.2)
According to Equations 1.1 and 1.2, we have:
iDR1 = iDR2 = iLf∕2 (1.3)
Hence, when both the power switches are off, the output filter
inductor current isshared by the two rectifier diodes DR1 and DR2,
with DFW removed.
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10 Soft-Switching PWM Full-Bridge Converters
1.2.3 Half-Bridge Converter
Figure 1.7a shows two single-switch forward converters connected
in series at theinput side and in parallel at the secondary
rectifiers, sharing the freewheeling diodeand the output filter,
where Cd1 and Cd2 are two input dividing capacitors. The valuesof
the two capacitors are equal and quite large, and the voltage
across each is Vin/2.Power switches Q1 and Q2 operate at the same
switching frequency with a time dif-ference of half the switching
period Ts/2. If the two transformers share a magneticcore and an
antiparallel diode D2 is connected across Q2, the transformer core
canbe magnetically reset through primary winding Np2 and D2 when Q1
is off. Likewise,inserting an antiparallel diode D1 across Q1, the
transformer core can be magneticallyreset through primary winding
Np1 and D1 when Q2 is off. Thus, the two magnetic corereset
circuits are redundant and can be removed, as shown in Figure 1.7b.
Exchang-ing the positions of Q1(D1) and primary winding Np1 leads
to the circuit shown inFigure 1.7c. There is current flowing
through each of the two primary windings when
* *
Q1
DR1
* Tr1
DFW
Dr1
NP1 NS1
Nr1vp1
+
_
+
_vsec1
Lf
Cf
RLd
Vo
+
_vrect
+
_
* *
Q2
DR2
*
Tr2
Dr2
Np2 Ns2
Nr2vp2
+
_
+
_vsec2
Cd1
Cd2
Vin
* *
Q1
DR1
*
TrDFW
Dr1
Np1 Ns1
Nr1vp1
+
_
+
_vsec1
Lf
Cf
RLd
Vo
+
_vrect
+
_
* *
Q2
DR2
*
Dr2
Np2 Ns2
Nr2vp2
+
_
+
_vsec2
Cd1
Cd2
Vin
D1
D2
(a) (b)
*
Q1
Np1
*
Q2
Np2
D1
D2
Tr
Tr
DR1
DR2
*
*
Ns1
Ns2
Cd1
Cd2
VinVin
vrect+
Lf
Cf
RLd
Vo
+
__
DFW
*
Q1
Np
Q2
D1
D2
Tr
Tr
DR1
DR2
*
*
Ns1
Ns2
Cd1
Cd2
vrect+
Lf
Cf
RLd
Vo
+
__
DFW
(c) (d)
Figure 1.7 Derivation of the half-bridge converter: (a) two
single-switch forward convertersconnected in series at the input
side and in parallel at the secondary rectifiers, sharing
thefreewheeling diode and the output filter, (b) the two
transformers sharing a magnetic core,(c) the positions of Q1 and
primary winding Ns interchanged, and (d) the final configurationof
the half-bridge converter
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 11
Q1 and Q2 conduct in turn. Since the nonpolarity-marked
terminals of the two primarywindings are connected, the
polarity-marked terminals have the same voltage poten-tial and can
be connected. It is obvious that the two primary windings are in
parallel,and one of them can be removed. Figure 1.7d shows the
final configuration of the con-verter, which is the half-bridge
converter. As in the case of the push–pull converter,the
freewheeling diode DFW can be removed and DR1 and DR2 conduct
simultaneouslywhen Q1 and Q2 are off.
The half-bridge converter is equivalent to two single-switch
forward converters con-nected in series at the input side. Thus,
the voltage applied on the input side of eachforward converter is
half the input voltage Vin/2 and the magnitude of the
primarywinding voltage is Vin/2, which is half that of the
push–pull converter. Key waveformsof the half-bridge converter are
given in Figure 1.6, with Vin replaced by Vin/2. As inthe push-pull
converter, the transformer of the half-bridge converter is
bidirectionallymagnetized.
The voltage stress of the power switches in the half-bridge
converter is 2 ⋅Vin/2=Vin;in fact, it can be seen from Figure 1.7d
that when either switch is conducting, the othermust withstand
Vin.
1.2.4 Full-Bridge Converter
The dual-switch forward converter was derived in Section 1.2.1
and is redrawn inFigure 1.8a for convenience. This converter has an
alternative configuration, shownin Figure 1.8b. When both switches
conduct, the transformer is negatively magnetized;when the switches
are off, the transformer is magnetically reset through diodes D1
andD4. The two kinds of dual-switch forward converter can be
connected in parallel at theinput sides and the output rectifier,
as shown in Figure 1.8c, where Q1(Q4) and Q2(Q3)operate at the same
frequency and with a time difference of Ts/2. If the two
transform-ers share a magnetic core and an antiparallel diode is
connected to each switch, asshown in Figure 1.8d, then the
transformer can be magnetically reset through primarywinding Np2
and the antiparallel diodes of Q2 and Q3, or through primary
windingNp1 and the antiparallel diodes of Q1 and Q4. Consequently,
diodes D1 to D4 can beremoved. Since the two primary windings have
the same voltage waveforms, points A1and A2 and points B1 and B2
can be connected, respectively. Thus, the two primarywindings are
in parallel, and one can be removed, resulting in the circuit shown
inFigure 1.8e. This is the full-bridge converter. Similarly, the
freewheeling diode DFWcan be removed. For brevity of illustration,
the antiparallel diodes of Q1 to Q4 arelabeled D1 to D4.
The transformer of the full-bridge converter is bidirectionally
magnetized, and themagnitude of the primary winding voltage is Vin,
which is the same as that of thepush–pull converter and twice that
of the half-bridge converter. The voltage stress ofthe power
switches of the full-bridge converter is Vin, which is the same as
that of thedual-switch forward converter.
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12 Soft-Switching PWM Full-Bridge Converters
*Np
Tr
D2
Q4
Q1
Vin Vin
D3
*
DR
DFW
Lf
Cf
RLd
Vo
+
_Ns
*Np
Tr
D1
Q3
Q2
D4
*
DR
DFW
Lf
Cf
RLd
Vo
+
_
Ns
(a) (b)
*Np1
Tr1
D2
Q4
Q1
Vin
D3
*Np2
Tr2
D1
Q3
Q2
D4
*
*
Ns1
Ns2
DR1
DR2
DFWLf
Cf
RLd
Vo
+
_vrect+
_*Np1
Tr1
Q4
Q1
Vin
*Np2
Tr2
Q3
Q2
*
*
Ns1
Ns2
DR1
DR2
DFWLf
Cf
RLd
Vo
+
_vrect
+
_
A1
B1
A2
B2
(c) (d)
Q1
Vin
D1
Q3 Q4
Q2 D2
D4D3
*Np
DR1
DR2
*
*
Ns1
Ns2
vrect+
Lf
Cf
RLd
Vo
+
__
DFW
(e)
Figure 1.8 Derivation of the full-bridge converter: (a)
dual-switch forward converter,(b) alternative configuration of the
dual-switch forward converter, (c) two kinds of dual-switchforward
converter connected in parallel at the input sides and the output
rectifier, (d) thetwo transformers sharing a magnetic core, and (e)
final configuration of the full-bridgeconverter
1.2.5 Comparison of Isolated Buck-Derived Converters
From the derivation of the forward converter (including
single-switch and dual-switchversions), the push–pull converter,
the half-bridge converter, and the full-bridge con-verter, it can
be concluded that all of these isolated converters are originated
from thebuck converter. Table 1.1 provides a comparison.
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 13
Table 1.1 Comparison of isolated buck-derived converters
Convertertype
Voltagestress ofpower
switches
Primary-to–secondary-winding-turns ratio
Currentstress ofpower
switches
Numberof powerswitches
Totalpower
handlingcapacityof powerswitches
Ripplefrequency of
secondaryrectifiedvoltage
Maximumduty
cycle ofsecondaryrectifiedvoltage
Single-switchforward
2Vin K0 Io/K0 1 2VinIo/K0 fs 0.5
Dual-switchforward
Vin K0 Io/K0 2 2VinIo/K0 fs 0.5
Push–pull 2Vin 2K0 Io/(2K0) 2 2VinIo/K0 2fs 1Half-bridge Vin K0
Io/K0 2 2VinIo/K0 2fs 1Full-bridge Vin 2K0 Io/(2K0) 4 2VinIo/K0 2fs
1
1. Voltage stress of power switches: The power switches of the
single-switch for-ward converter and the push–pull converter have
to withstand twice the input volt-age, while the power switches of
the dual-switch forward converter, the half-bridgeconverter, and
the full-bridge converter are required to withstand the input
voltage.Thus, the single-switch forward converter and the push–pull
converter are suit-able for low-voltage applications, while the
dual-switch forward converter, thehalf-bridge converter, and the
full-bridge converter are suitable for
high-voltageapplications.
2. Transformer primary-to-secondary-winding-turns ratio: The
maximum dutycycle of the forward converter (including single-switch
and dual-switch versions)is limited to 0.5 and the maximum duty
cycle of the push–pull converter, thehalf-bridge converter, and the
full-bridge converter can reach unity. Under condi-tions of the
same input and output voltages, if the transformer winding turns
ratio ofthe forward converter is K0, the transformer-winding-turns
ratios of the push–pullconverter and of the full-bridge converter
should be 2K0. For the half-bridge con-verter, although its duty
cycle can reach unity, the magnitude of the primary voltageis half
of the input voltage, so its transformer-winding-turns ratio is
K0.
3. Current stress of power switches: Neglecting the output
filter inductor currentripple, the current stresses of the power
switches of the forward converter and ofthe half-bridge converter
are both Io/K0, while the current stresses of the powerswitches of
the push–pull converter and of the full-bridge converter are
bothIo/(2K0), where Io is the output current.
4. Total power handling capacity of power switches: The power
handling capacityof a power switch is defined as the product of the
voltage stress and current stressimposed on the power switch. From
the preceding analysis, it can readily beseen that the total power
handling capacity (i.e., the number of power switches
-
14 Soft-Switching PWM Full-Bridge Converters
multiplied by each power switch’s power handling capacity) of
the five isolatedbuck-derived converters is 2VinIo/K0. This means
that for the same input andoutput, the total power handling
capacities of switches of all five converters arethe same. In other
words, if the power switches have the same voltage and
currentratings, the output power the converter can handle is
proportional to the numberof power switches. Of the five isolated
buck-derived converters, the full-bridgeconverter has the most
power switches (four) and the highest power handling capa-bility.
Therefore, the full-bridge converter has been widely used in
medium-to-high-power-conversion applications.
5. Output filter: For the forward converter, the ripple
frequency of the secondaryrectified voltage is the switching
frequency fs and the maximum duty cycle islimited to 0.5. For the
push–pull converter, the half-bridge converter, and thefull-bridge
converter, the secondary rectified voltage has a ripple frequency
of 2fsand a maximum duty cycle of 1. Therefore, for the same output
voltage, the sec-ondary rectified voltages of the push–pull
converter, the half-bridge converter, andthe full-bridge converter
have smaller amounts of high-frequency harmonics thanthat of the
forward converter, and thus the required output filter is much
smaller.
1.3 Output Rectifier Circuits
Section 1.2 presented the derivations of the forward converter
(including single-switchand dual-switch versions), the push–pull
converter, the half-bridge converter, andthe full-bridge converter.
The output rectifier circuit of the forward converter is ahalf-wave
rectifier circuit, while those of the push–pull converter, the
half-bridgeconverter, and the full-bridge converter are full-wave
rectifier circuits. In fact, sincethe push–pull converter, the
half-bridge converter, and the full-bridge converterall transfer
energy from the input to the load during both the positive and the
neg-ative half-periods, they can all also adopt the full-bridge
rectifier circuit and thecurrent-doubler rectifier circuit. In this
section, the half-wave rectifier circuit, thefull-bridge rectifier
circuit, and the current-doubler rectifier circuit will be
derivedfrom the half-wave rectifier circuit. The purpose of this is
to reveal the relationshipamong these output rectifier
circuits.
1.3.1 Half-Wave Rectifier Circuit
Figure 1.9a and Figure 1.9b show the positive and negative
half-wave rectifier circuits,respectively. The two half-wave
rectifier circuits can only transfer energy to load in thepositive
or negative half-period of transformer primary voltage vp. The key
waveformsare depicted in Figure 1.10, from which it can be seen
that:
1. The output voltage Vo can be derived as:
Vo = DhVpm∕K (1.4)
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 15
Tr
DR1 DFW1
*
Lf
Cf
RLd
*Vo
+
_vp
+
_vsec1
+
_vrect
+
_
iLf
Tr
DR2
**DFW2
Lf
Cf
RLd
Vo
+
_
vp
+
_vsec2
+
_
vrect
+
_
iLf
(a) positive half-wave rectifier circuit (b) negative half-wave
rectifier circuit
Figure 1.9 Two kinds of half-wave rectifier circuit
t
t
vrect
Ts/2 Ts
0
0
iLf
vsec1
0
t
Vpm/K
Vo
Io
Vpm /K
0
Vpm /K
t
t
vrect
0
0
iLf
vsec2
0
t
Vpm/K
Vo
Io
0
Vpm/K
Ts/2 Ts
Vpm/K
(a) Positive half-wave rectifier circuit (b) Negative half-wave
rectifier circuit
Figure 1.10 Waveforms of two kinds of half-wave rectifier
circuit
where Vpm is the magnitude of the transformer primary voltage,
Dh is the dutycycle of the half-wave rectifier circuit, which is
the ratio of the width of thepositive (or negative) half-period to
the switching period, and K is the
primary-to-secondary-winding-turns ratio.
2. The ripple frequency of the rectified voltage and output
filter inductor current isthe switching frequency.
3. The voltage stress of both the rectifier diode and the
freewheeling diode is Vpm/K.
1.3.2 Full-Wave Rectifier Circuit
If the transformer is expected to transfer energy to the load
during both positive andnegative half-periods, the positive and
negative half-wave rectifier circuits should becombined, as shown
in Figure 1.11a. As illustrated in Section 1.2, when vp is equal
tozero, the output filter inductor can freewheel through
freewheeling diodes DFW1 andDFW2 or through two secondary windings
via DR1 and DR2. Therefore, DFW1 and DFW2
-
16 Soft-Switching PWM Full-Bridge Converters
DR2
DFW1DR1
DFW2
Lf
Cf
RLd
Vo
+
_vrect
+
_
Tr
*
*
* vsec1
+
_
vsec2
+
_
vp
+
_
Tr
DR2
*
*
Lf
Cf
RLd
*
DR1
Vo
+
_vsec1
+
_
vsec2
+
_
vp
+
_
vrect
+
_
(a) With freewheeling diodes (b) Without freewheeling diodes
Figure 1.11 Full-wave rectifier circuit
t
t
vrect
0
vsec1vsec2
0
Vpm/K
Vo
Vpm/K
Vpm/K
0
iLf
t
Io
0 Ts/2 Ts
0
iDR1
t
Io
iDR2
Figure 1.12 Key waveforms of the full-wave rectifier circuit
are redundant and can be removed, as shown in Figure 1.11b.
Thus, the full-waverectifier circuit is obtained. Figure 1.12 shows
the key waveforms of the full-waverectifier circuit, from which we
see that:
1. The output voltage Vo is given by:
Vo = 2DhVpm∕K = DyVpm∕K (1.5)
where Dy is the duty cycle of the secondary rectified voltage,
which is the ratioof the pulse width of the secondary rectified
voltage to half the switching period.Here, Dy is twice the duty
cycle of the half-wave rectifier circuit; that is, Dy = 2Dh.
2. The ripple frequency of the rectified voltage and the output
filter inductor currentis twice the switching frequency.
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 17
3. The voltage stress of the two rectifier diode is 2Vpm/K.
During the freewheelingperiod, the two rectifier diodes share the
output filter inductor current.
From Equations 1.4 and 1.5, it can be seen that the output
voltage of the full-waverectifier circuit is twice that of the
half-wave rectifier circuit when Dh is the same. Thisis because
with the full-wave rectifier circuit, voltage is applied on the
output filter inboth the positive and the negative half-periods. If
the output voltage stays the same, thetransformer-turns ratio of
the full-wave rectifier circuit is twice that of the
half-waverectifier circuit and the voltage stresses of the
rectifier diodes of the two kinds ofrectifier circuit are equal.
Compared with the half-wave rectifier circuit, the ripplefrequency
of the secondary rectified voltage is doubled and the
switching-frequencyharmonics are significantly reduced. Thus, the
output filter can be drastically reduced.
1.3.3 Full-Bridge Rectifier Circuit
In the full-wave rectifier circuit, the transformer has two
secondary windings, witheach one only conducting for a half-period.
If the secondary winding can conductcurrent during both positive
and negative half-periods, the utilization is increasedand one
secondary winding can be removed, leading to a simple configuration
of thetransformer. Figure 1.13a shows the rectifier circuit with
only one secondary winding
TrDR1
DFW1
*
Lf
Cf
RLd
*
DR2
DFW2
Vo
+
_vp
+
_vsec
+
_
TrDR1
DFW1
*
Lf
Cf
RLd
*
DR2
DFW2
DR3
DR4
Vo
+
_vp
+
_vsec
+
_
(a) (b)
(c) (d)
vp
+
_vsec
+
_
Tr
DFW1
*
Lf
Cf
RLd
*
DFW2
Vo
+
_
DR1 DR2
DR3 DR4
Tr**
DR1 DR2
DR3 DR4
Lf
Cf
RLd
Vo
+
_
vp
+
_vsec
+
_
iLf
vrect
+
_
Figure 1.13 Derivation of the full-bridge rectifier circuit: (a)
preliminary bidirectional rec-tifier circuit with one secondary
winding, (b) bidirectional rectifier circuit with one
secondarywinding, (c) full-bridge rectifier circuit with
freewheeling diodes, and (d) full-bridge rectifiercircuit
-
18 Soft-Switching PWM Full-Bridge Converters
t
t
vrect
0
vsec
0
Vpm/K
Vo
Vpm/K
Vpm/K
0
iLf
t
Io
0 Ts/2 Ts
0
iDR1
t
Io
iDR2iDR3 iDR4
Figure 1.14 Key waveforms of the full-bridge rectifier
circuit
conducting current bidirectionally, where DR1 and DFW1 are the
rectifier diode andfreewheeling diode of the positive half-wave
rectifier circuit, respectively, and DR2and DFW2 are the rectifier
diode and freewheeling diode of the negative half-wave rec-tifier
circuit, respectively. However, such an arrangement leads to
short-circuit of thesecondary winding; diodes DR4 and DR3 should be
inserted to avoid this, as shownin Figure 1.13b. Redrawing the
circuit in Figure 1.13b as Figure 1.13c, it can easilybe seen that
the output filter inductor current can freewheel through DFW1 or
DFW2,or through the branch consisting of DR1 and DR3, or through
the branch consisting ofDR2 and DR4. Therefore, DFW1 and DFW2 are
redundant and can be removed, leadingto the well-known full-bridge
rectifier circuit, as shown in Figure 1.13d.
Figure 1.14 shows the key waveforms of the full-bridge rectifier
circuit, from whichwe can conclude the following:
1. The expression of the output voltage is the same as Equation
1.5.2. As in the full-wave rectifier circuit, the ripple frequency
of the rectified voltage
and output filter inductor current is twice the switching
frequency.3. The voltage stress of all of the rectifier diodes is
Vpm/K. During the freewheeling
period, the rectifier diodes share the output filter inductor
current.
1.3.4 Current-Doubler Rectifier Circuit
The positive and negative half-wave rectifier circuits are
redrawn in the forms shownin Figure 1.15a and Figure 1.15b,
respectively. If the two rectifier circuits share the
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 19
DR1
DR1
DR2 DR2
DR1
RLd RLd RLdDFW1
DFW2
**Np Ns Np Ns Np Ns
Cf CfCf
Vo Vo Vo
io
iLf 2
iLf1Tr Tr Tr
Lf1
Lf 2
Lf 1
vrect
Lf 2
+
_ **
+
_ **
+
_
+
_
(a) (b) (c)
Figure 1.15 Derivation of the current-doubler rectifier circuit:
(a) positive half-wave rectifiercircuit, (b) negative half-wave
rectifier circuit, and (c) current-doubler rectifier circuit
t
t
0
0
Vo
Io
Io
Io/2
Ts/2 Ts
iDR2
iLf 2iLf 1
iDR1
0 t
00 t
vsecVpm/K
Vpm/K
Vpm/K
vrect
iLf 1+iLf 2
Figure 1.16 Key waveforms of the current-doubler rectifier
circuit
same set of transformer, rectifier diode, and freewheeling
diode, the current-doublerrectifier circuit is obtained, as shown
in Figure 1.15c, where two output filter inductorcurrents are
supplied to the output filter capacitor and load. Figure 1.16 shows
thekey waveforms of the current-doubler rectifier circuit, whose
characteristics are asfollows:
1. It can be treated as the parallel of the positive and
negative half-wave rectifiercircuits. The expression of the output
voltage is thus the same as that of the
-
20 Soft-Switching PWM Full-Bridge Converters
half-wave rectifier circuit; that is, Equation 1.4. In other
words, with the sametransformer-turns ratio, the output voltage of
the current-doubler rectifier circuitis half that of the full-wave
rectifier circuit or the full-bridge rectifier circuit.
2. The two output filter inductor currents pulsate at the
switching frequency with aphase shift of 180∘ and the output
current io is the sum of the two output filter induc-tor currents
and ripples at twice the switching frequency. Since two output
filterinductor currents have a phase shift of 180∘, the current
ripples at the switching fre-quency and its odd multiples are
cancelled in the output current, which means thatthe ripple of io
is smaller than the individual output filter inductor current
ripple.
3. The voltage stress of the two rectifier diodes is Vpm/K.
Under conditions of thesame input and output voltages, and with the
same duty cycle Dh, the transformer-winding-turns ratio of the
current-doubler rectifier circuit is half that of thefull-wave
rectifier circuit and the full-bridge rectifier circuit. Therefore,
the volt-age stress of the rectifier diodes is equal to that of the
full-wave rectifier circuit anddouble that of the full-bridge
rectifier circuit.
From this analysis, it can be concluded that the full-wave
rectifier circuit, thefull-bridge rectifier circuit, and the
current-doubler rectifier circuit can be derivedfrom the half-wave
rectifier circuit. Table 1.2 compares the three kinds of
rectifiercircuit, which can be described as follows:
1. Transformer primary-to-secondary-winding-turns ratio: The
full-wave recti-fier circuit and the full-bridge rectifier circuit
can be treated as a series connec-tion of positive and negative
half-wave rectifier circuits, while the current-doublerrectifier
circuit can be treated as a parallel connection of positive and
negativehalf-wave rectifier circuits. Therefore, when operating
with the same duty cycleDh and transformer
primary-to-secondary-winding-turns ratio, the output voltageof the
full-wave rectifier circuit and the full-bridge rectifier circuit
is twice thatof the current-doubler rectifier circuit. In other
words, if under the same input andoutput voltages, the
transformer-turns ratio of the full-wave rectifier circuit and
thefull-bridge rectifier circuit is K0, that of the current-doubler
rectifier circuit shouldbe K0/2.
Table 1.2 Comparison of three kinds of rectifier circuit
Rectifier circuit type Transformerprimary-to-secondary-turns
ratio
Voltagestress ofrectifierdiodes
Currentstress ofrectifierdiodes
Number ofrectifierdiodes
Total powerhandling
capacity ofrectifier diodes
Full-wave rectifier K0 2Vin/K0 Io 2 4VinIo/K0Full-bridge
rectifier K0 Vin/K0 Io 4 4VinIo/K0Current-doubler rectifier K0/2
2Vin/K0 Io 2 4VinIo/K0
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 21
2. Voltage stress of rectifier diodes: The voltage stress of the
rectifier diodes of thefull-wave rectifier circuit and the
current-doubler rectifier circuit is 2Vin/K0, whilethat of the
rectifier diodes of the full-bridge rectifier circuit is Vin/K0,
where Vin isthe input voltage.
3. Current stress of rectifier diodes: Neglecting the output
filter inductor currentripple, the current stress of the rectifier
diodes in all three rectifier circuits is theoutput current Io.
4. Total power handling capacity of rectifier diodes: The power
handling capacityof a rectifier diode is defined as the product of
the voltage stress and the currentstress of the rectifier diode.
From the preceding discussion, it can be seen thatthe total power
handling capacity of the rectifier diodes (i.e., the power
handlingcapacity of each rectifier diode multiplied by the number
of rectifier diodes in allthree rectifier circuits) is equal to
4VinIo/K0.
1.4 Basic Operating Principle of Full-Bridge Converters
1.4.1 Topologies of Full-Bridge Converters
Since the transformer of the full-bridge converter transfers
energy from the inputto the load during both the half-periods, the
rectifier circuit of the full-bridge con-verter can be either the
full-wave rectifier circuit, the full-bridge rectifier circuit,
orthe current-doubler rectifier, as shown in Figure 1.17.
1.4.2 Pulse-Width Modulation Strategies for Full-Bridge
Converters
Figure 1.18 shows the commonly used PWM strategies for the
full-bridge converter.In the basic PWM strategy, the two diagonal
switches of the two legs turn on and offsimultaneously, as shown in
Figure 1.18a. The two switches in one bridge leg canalso be
operated in a PWM fashion, and the two in the other bridge leg in a
comple-mentary manner with 50% duty cycle, as shown in Figure
1.18b. Figure 1.18c showsthe well-known phase-shift control, where
the switches of each bridge leg operatecomplementarily with 50%
duty cycle and a phase-shift is introduced between thetwo legs.
Regulation of the output voltage can thus be achieved by
controlling thephase-shift.
1.4.3 Basic Operating Principle of a Full-Bridge Converterwith a
Full-Wave Rectifier Circuit and a Full-BridgeRectifier Circuit
Regardless of the kind of PWM strategy adopted, the operating
principle of thefull-bridge converter with a full-wave rectifier
circuit is the same. In the following,the basic strategy is used
for illustration. Figure 1.19 shows the key waveforms of the
-
22 Soft-Switching PWM Full-Bridge Converters
*
*
*
+
_
+
_
A B
C
D
A B
*
*
DR2+
_
+
_
*
+
_
O
C
ip
A B
*
Vin
Vin
Vin
vrect
vrect
Tr
Tr
Np Ns
Tr
Np Ns
Lf
Lf
iLf
iLf
Lf1
iLf1
Lf 2
iLf2
Cf
Cf
ip
ip
io
Q2
Q4
Q2
Q4
Q1
Q1
D1 D2
D4
D2
D4
Q2
Q4
D2
D4
DR1
DR1
DR3
DR2
DR1
DR4
RLd
Cf
RLd
RLd
D3
D1
D3
Q3
Q3
Q1 D1
D3Q3
(a) With full-wave rectifier circuit
(b) With full-bridge rectifier circuit
(c) With current-doubler rectifier circuit
Vo
Vo
Vo
Figure 1.17 Full-bridge converter with different output
rectifier circuits
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 23
t
t0
t
t0
(b) One leg PWM operated and the other operatingin symmetrical
complementary manner
tQ1 Q3 Q1
t0
Q4 Q2 Q4
Q1 Q3
Q2
Q1
Q4Q4
Q1 Q3
Q2
Q1
Q4Q4
(a) Diagonal switches turning on and off simultaneously
(c) Phase-shifted control
Ts/2 TsTon
Ts/2 TsTon
Ts/2 TsTon
Figure 1.18 Widely used modulation strategies for the
full-bridge converter
full-bridge converter with a full-wave rectifier circuit, while
the equivalent circuits ofthe topological modes are depicted in
Figure 1.20.
When the diagonal switches Q1 and Q4 are conducting, as shown in
Figure 1.20a, thevoltage across the midpoints of the bridge legs
vAB is equal to Vin, the secondary rec-tifier diode DR1 conducts,
and the secondary rectified voltage vrect is equal to Vin/K,where K
is the transformer primary-to-secondary-winding-turns ratio. The
voltageapplied across the output filter inductor Lf is Vin/K−Vo,
which causes the inductorcurrent iLf to increase linearly. The
primary current ip is equal to the output filterinductor current
reflected to the primary side (i.e., ip = iLf/K), and
correspondinglyit increases linearly. Also, ip flows through Q1 and
Q4.
When the diagonal switches Q2 and Q3 are conducting, as shown in
Figure 1.20b,vAB =−Vin, DR2 conducts, vrect =Vin/K, and iLf
increases linearly.
When all of the power switches are off, the primary current ip
is zero and theoutput filter inductor current freewheels through
the two rectifier diodes as shown inFigure 1.20c. The two rectifier
diodes shares the output filter inductor current (i.e.,iDR1 = iDR2
= iLf/2). Since both rectifier diodes are conducting, the two
secondarywinding voltages are clamped to zero. Thus, vrect = 0 and
the voltage applied to theoutput filter inductor is −Vo. This
negative voltage makes iLf decay linearly. If the
-
24 Soft-Switching PWM Full-Bridge Converters
t
t
0
0
0 t
00 t
t0
0 t
t
t
0
0
0 t
00 t
t0
0 t
Q1&Q4 Q2&Q3 Q1&Q4 Q1&Q4 Q2&Q3 Q1&Q4
Ts/2 TsTs/2 Ts
VinVinvAB vAB
ip ip
Vin
Vo
Io
Io
Vo
Io
Vin/K Vin/K
Vin
vrect
iLf
iDR1 iDR4
iDR1iDR4
iDR2 iDR3
iDR2iDR3
iLf
vrect
(a) Continuous current mode (b) Discontinuous current mode
Figure 1.19 Key waveforms of the full-bridge converter with
full-wave rectifier circuit
load is light or the output filter inductor is small, iLf may
decay to zero before one pairof diagonal switches is turned on; it
is then kept at zero, as shown in Figure 1.20d.Under this
condition, the full-bridge converter operates in discontinuous
currentmode (DCM). The corresponding waveforms are sketched in
Figure 1.19b.
The operating principle of the full-bridge converter with a
full-bridge rectifier cir-cuit is similar to that of one with a
full-wave rectifier circuit. The key waveforms aregiven in Figure
1.19, where the rectifier diode current waveforms are shown. For
theequivalent circuits, the primary side is the same, while the
secondary side is shownin the dashed block in Figure 1.20. The
difference between the two rectifier circuitsis that, for the
full-bridge rectifier circuit, when all of the power switches are
off,the output filter inductor current freewheels through the four
rectifier diodes, whilethe secondary winding current is zero.
1.4.4 Basic Operating Principle of a Full-Bridge Converterwith a
Current-Doubler Rectifier Circuit
The full-bridge converter with a current-doubler rectifier
circuit is shown inFigure 1.17c. The operation of a converter
adopting the basic PWM strategy (see
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 25
*
*
*
+
_
+
_
B
B
*
*
+
_
+
_
A B
*
*
+
_
+
_
*
*
*
+
_
+
_
*
*
+
_
+
_
*
*
*
+
_
+
_
C
D *
*
*
+
_
+
_
D
C
*
*
+
_
+
_
Vin
ip
ip
ip
ip
Vin
Q1 D1
DR1
DR1
DR1 DR2
DR3 DR4
DR2 DR2
DR2
DR3 DR3DR4
D3Q3
A
Q1 D1
D3Q3
A
A
BA
Q2
Q4
D2
D4
BQ2
Q4
D2
D4
BVin
Q1 D1
D3Q3
AQ2
Q4
D2
D4
CTr
Tr
Tr
Tr
Np Ns
DR1
DR2
D
vrect
vrect
Lf
iLf
Lf
iLf
Lf
iLf
Lf
iLf
Cf
Cf
Cf
Cf
RLd
RLd
RLd
RLd
Vo
Vo
Vo
Vo
Vo
B
ip
DR1DR2
DR4
A
TrNp Ns
Np Ns Ns
vrect
vrect
vrectip
DR1 DR2
DR3 DR4
BA
Tr
Lf
iLf
Cf
RLd
VoNp vrect
ip
Vin
DR1
Q1 D1
D3Q3
A BQ2
Q4
D2
D4
Tr
Lf
iLfCf
RLd
vrect
Lf
iLf
Cf
RLd
Vo
ipC
TrDR1
DR2
D
vrect
Lf
iLfCf
RLd
Vo
(a) Q1 and Q4 are conducting (b) Q2 and Q3 are conducting
(c) All power switches are off (d) Output filter inductor
current is zero
Figure 1.20 Equivalent circuit of switching modes of the
full-bridge converter
-
26 Soft-Switching PWM Full-Bridge Converters
Figure 1.18a) is different from that of one adopting either of
the other two PWMstrategies (see Figure 1.18b,c).
1.4.4.1 Basic PWM Strategy
At Heavy LoadThe key waveforms of the full-bridge converter with
a current-doubler rectifier circuitunder heavy load condition are
shown in Figure 1.21a. When the diagonal powerswitches Q1 and Q4
conduct, the secondary rectifier diode DR1 also conducts, as
shownin Figure 1.22a. During this interval, vAB =Vin and vCO
=Vin/K. The voltage across Lf1is Vin/K−Vo and iLf1 increases
linearly, while the voltage across Lf2 is −Vo and iLf2decreases
linearly. The primary current ip is equal to iLf1 reflected to the
primary side(i.e., ip = iLf1/K) and increases linearly. The output
current io is the sum of the twooutput filter inductor currents
(i.e., io = iLf1 + iLf2) and also increases linearly.
t0
0 t
00 t
t0
0
0
t
t
t0
Vin
0 t
0 t
t0
0
0
t
t
0
Q1&Q4 Q2&Q3 Q1&Q4 Q1&Q4 Q2&Q3 Q1&Q4
Vin Vin
Vin
vABvAB
vCO
ip ip
vCO
Io
Io/2
Vo Vo
KVo
Vin/K
Vin/K
iDR1 iDR2
iDR1 iDR2
Ts/2 TsTon Ts/2 TsTon Tr
iLf 1
iLf 1 iLf 2
iLf 2
iLf 1+iLf 2
iLf 1+iLf 2
(a) At heavy load (b) At light load and iLf 1 and iLf 2 keep
positive
Figure 1.21 Key waveforms of the full-bridge converter with
current-doubler rectifier circuitunder the basic PWM strategy
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 27
t0
0 t
0 t
t0
0
0
t
t
0
t0
0 t
00 t
t0
0
0
t
t
Q1&Q4 Q2&Q3 Q1&Q4 Q1&Q4 Q2&Q3 Q1&Q4
Vin Vin
Vin
ip
KVo
Vo
vAB vAB
vCO
ip
vCOVin/K Vin/K
iLf 1 iLf 1iLf 2 iLf 2iLf 1+iLf 2 iLf 1+iLf 2
iDR1iDR1
iDR2iDR2
Ts/2 TsTon Ts/2 TsTon
(d) Discontinuous current mode (c) iLf 1 and iLf 2 are
bidirectional
Figure 1.21 (Continued)
When Q1 and Q4 are turned off, iLf1 and iLf2 flow through DR2
and DR1, respectively,and ip becomes zero, as shown in Figure
1.22b. At this point, the voltages acrossthe two output filter
inductors are all −Vo, making iLf1 and iLf2 decay linearly.
Sinceboth DR1 and DR2 are conducting, vCO = 0 and the secondary
winding voltage is zero.Correspondingly, the primary voltage vAB =
0.
The operation when Q2 and Q3 are conducting is similar to that
when Q1 and Q4 areconducting, which is omitted here.
At Light Load when Output Filter Inductor Currents are Positive
as DiagonalSwitches ConductAs just discussed, when Q1 and Q4 are
turned off, iLf1 and iLf2 freewheel through DR2and DR1,
respectively, and both decay linearly, as shown in Figure 1.22b. If
the load isvery light or the output filter inductor is not large
enough, iLf2 will decay to zero andDR1 will turn off naturally, as
shown in Figure 1.22c. Following this, iLf2 is kept at zerountil Q2
and Q3 are turned on, as shown in the dashed lines within interval
[Tr,Ts/2]in Figure 1.21b. During this interval, vAB =−KVo and vCO =
0. Similarly, when iLf1decays to zero, it stays at zero, as shown
in the dashed lines in Figure 1.21b. Duringthis interval, vAB =KVo
and vCO =Vo.
-
28 Soft-Switching PWM Full-Bridge Converters
*
+
_
* O **
*
*
+
_ *
+
_
*
C
ip
Vin
DR2
DR1
Q1 D1
D3Q3
A BQ2
Q4
D2
D4
Tr
Lf1
iLf1
Lf 2
iLf 2
Cf
RLd
VoNp Ns
io
Q
O
C
ip
Vin
DR2
DR1
Q1 D1
D3Q3
A BQ2 D2
D4
Tr
Lf 1
iLf1
Lf 2
iLf 2
Cf
RLd
VoNp Ns
io
O
C
ip
Vin
DR2
DR1
Q1 D1
D3Q3
A BQ2
Q4
D2
D4
Tr
Lf1
iLf1
Lf 2
iLf 2
Cf
RLd
VoNp Ns
io
Q4
+
_
O
C
ip
Vin
DR2
DR1
Q1 D1
D3Q3
A BQ2 D2
D4
Tr
Lf 1
iLf1
Lf 2
iLf 2
Cf
RLd
VoNp Ns
io
(a) Q1 and Q4 are conducting (b) All switches are off and iLf 2
>0
(c) All switches are off and iLf 2 = 0 (d) All switches are off
and iLf 2 < 0
Figure 1.22 Equivalent circuits of switching modes of the
full-bridge converter withcurrent-doubler rectifier circuit under
the basic PWM strategy
At Light Load when Output Filter Inductor Current becomes
Negativeas Diagonal Switches ConductWhen Q1 and Q4 are conducting,
iLf1 increases, iLf2 decays, and ip is equal to thereflected iLf1
(i.e., ip = iLf1/K). If the load becomes lighter, iLf2 will cross
zero and con-tinue flowing in the negative direction. When Q1 and
Q4 are turned off, iLf1 freewheelsthrough DR2 and decays linearly.
Moreover, iLf2 flows through the secondary windingand is reflected
to the primary side (i.e., ip =−iLf2/K). Thus, ip is positive and
flowsthrough D2 and D3, as shown in Figure 1.22d. During this
interval, vAB =−Vin and thevoltage across Lf2 is Vin/K−Vo, forcing
iLf2 to increase linearly. When iLf2 increasesto zero, ip reduces
to zero correspondingly. Since all the power switches are off,
ipcannot flow in the reverse direction and must be kept at zero,
which makes iLf2 stay atzero until Q2 and Q3 are turned on. It can
be seen from Figure 1.21c that, compared
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 29
with Figure 1.21b, there are additional portions in vAB and vCO,
shown as the dashedarea. Thus, Vo is equal to the average value of
vCO, and clearly Vo is dependent notonly on the duty cycle but also
on the load.
If the load continues to reduce, the hatched area shown in
Figure 1.21c increasesto reach the turn-on instant of diagonal two
switches, as shown in Figure 1.21d. Thismeans that at the turn-on
instant of Q2 and Q3, iLf2 is still negative. When iLf2 increasesto
zero, it can continue to flow in the positive direction, and it is
continuous. Underthis condition, vAB becomes an ac square voltage
with 180
∘ electrical degree, whilevCO becomes a pulse voltage with a
magnitude of Vin/K and a pulse width of Ts/2. Theaverage value of
vCO is equal to Vin/(2K). Since Vo is the average value of vCO, it
isequal to Vin/(2K) and is independent of the duty cycle.
This analysis illustrates that, employing the basic PWM
strategy, the output volt-age of the full-bridge converter with a
current-doubler rectifier circuit is dependantnot only on the duty
cycle but also on the load. Furthermore, the output voltage
willlose control at light load. Therefore, the basic PWM strategy
is not suitable for thefull-bridge converter with a current-doubler
rectifier circuit.
1.4.4.2 Phase-Shifted Control
Figure 1.23 shows the key waveforms of the full-bridge converter
with a current-doubler rectifier circuit when the phase-shifted
control is adopted. At heavy load,the operation is the same as that
adopting the basic PWM strategies, as shown inFigure 1.23a.
At light load, when Q1 and Q4 are conducting, iLf2 decays and
becomes negative.When the leading-leg switches Q1 is turned off and
Q3 is turned on, iLf1 is freewheelingthrough DR2 and decays
linearly and iLf2 flows through the secondary winding and
isreflected to the primary side (i.e., ip =−iLf2/K). Here, ip flows
through D3 and Q4, asshown in Figure 1.24a. At this time, vAB = 0
and the voltage across Lf2 is still −Vo,meaning iLf2 continues to
decay. Figure 1.23b shows the key waveforms under
thiscondition.
If the load becomes lighter, the two output filter inductor
currents decay linearlywhen vAB = 0. When iLf1 =−iLf2, DR2 is
turned off, and iLf1 and iLf2 are kept unchanged.Thus, io = 0. At
this time, the full-bridge converter operates in DCM. It should be
notedthat, for the current-doubler rectifier circuit, “DCM” refers
to the sum of the two out-put filter inductor currents flowing into
the load being in discontinuous conduction;the two output filter
inductor currents are not zero. Correspondingly, continuing
cur-rent mode (CCM) refers to when the sum of two output filter
inductor currents iscontinuous.
From this analysis, the following conclusions can be drawn:
1. When the phase-shifted control is adopted, regardless of
whether the operationis in CCM or DCM, the output voltage of the
full-bridge converter with a
-
30 Soft-Switching PWM Full-Bridge Converters
t0
Vin
0 t
00 t
t0
0
0
t
t
t
t0
0 t
00 t
t0
0
0
t
t
t
iDR1
iDR1
iDR2
iDR2
iLf 1iLf 1
iLf 2
iLf 2
iLf 1+iLf 2
iLf 1+iLf 2
Vin
Vin/K
Vin
Vin
Vin/K
Q1 Q3 Q1 Q1 Q3 Q1
Q2 Q4Q4 Q2 Q4Q4
vAB
vCO
ip
vAB
vCO
ip
Io
Io
Io/2
Ts/2 TsTonTs/2 TsTon
(a) At heavy load (b) At light load and iLf 1 and iLf 2 are
bidirectional
Figure 1.23 Key waveforms of the full-bridge converter with
current-doubler rectifier circuitunder phase-shifted control
current-doubler rectifier circuit can be regulated by
controlling the duty cycle.This is different from the basic PWM
strategy.
2. If the output filter inductor current becomes negative in
active mode (vAB =+Vin or−Vin), it will be reflected to the primary
side when vAB = 0. The induced primarycurrent increases linearly.
This can be used to achieve ZVS for the lagging leg,which will be
discussed in Chapter 8.
The operation of the full-bridge converter with a
current-doubler rectifier circuitemploying the PWM modulation
strategy shown in Figure 1.18b is the same as thatusing a
phase-shifted control. Details are omitted here.
-
Topologies and Operating Principles of Basic Full-Bridge
Converters 31
t0
0 t
00 t
t0
0
0
t
t
t
0 t
Q1 Q3
Q2
Q1
Q4Q4
Ts/2 TsTon
Vin
Vin
Vin/K
vAB
vCO
ip
iDR1
iDR2
iLf 1 iLf 2iLf 1+iLf 2
(c) DCM
Figure 1.23 (Continued)
*
*
+
*
*
ip
Vin
Q1 D1
D3Q3
A
+
_
C
DR2
DR1
BQ2
Q4
D2
D4
Tr
Lf1
iLf1
(a) iLf1 > _ iLf 2 (a) iLf1 =
_ iLf 2
Lf 2
iLf 2
Cf
RLd
VoNp Ns
io ip
Vin
Q1 D1
D3Q3
A
+
_
C
DR2
DR1
BQ2
Q4
D2
D4
Tr
Lf1
iLf1
Lf 2
iLf 2
Cf
RLd
VoNp Ns
io
Figure 1.24 Equivalent circuits when Q3 and Q4 are
conducting
-
32 Soft-Switching PWM Full-Bridge Converters
1.5 Summary
This chapter introduced some development trends in the switching
techniques,classifications, and requirements of power electronics
converters, as well as thetypes and characteristics of dc–dc
converters. The forward converter (includingsingle-switch and
dual-switch versions), push–pull converter, half-bridge
converter,and full-bridge converter were derived from the buck
converter, in order to help read-ers understand more clearly the
relationships among various isolated buck-derivedconverters.
Meanwhile, the full-wave rectifier circuit, the full-bridge
rectifier circuit,and the current-doubler rectifier circuit were
derived from the half-wave rectifiercircuit. Again, the emphasis
was on the relationships among these rectifier circuits.The basic
operation of the full-bridge converter with full-wave rectifier
circuit,full-bridge rectifier circuit, and current-doubler
rectifier circuit was analyzed. Thebasic background necessary for
study of the operation and design of soft-switchingPWM full-bridge
converters has thus been provided.
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