Speed Control of Dc Motor Using PWM

DC MOTOR

D c machines are characterized by their versatility. By means of various

combinations of shunt-, series-, and separately-excited field windings they can be

designed to display a wide variety of volt-ampere or speed-torque characteristics

for both dynamic and steady-state operation. Because of the ease with which

they can be controlled, systems of dc machines have been frequently used in

applications requiring a wide range of motor speeds or precise control of motor

output. In recent years, solid-state ac drive system technology has developed

sufficiently that these systems are replacing dc machines in applications previously

associated almost exclusively with dc machines. However, the versatility of dc

machines in combination with the relative simplicity of their drive systems will

insure their continued use in a wide variety of applications. Before the widespread

application of power-electronic drives to control ac machines, dc motors were by far

the machines of choice in applications requiring flexibility of control. Although in

recent years ac drives have become quite common, the ease of control of dc machines

insure their continued use in many applications.

Typical steady-state dc-motor speed-torque characteristics are shown in Fig. 7.6, in

which it is assumed that the motor terminals are supplied from a constant-voltage

source. In a motor the relation between the emf Ea generated in the armature and the

armature terminal voltage Va is

Va = Ea + IaR.

Where,

Ia is now the armature-current input to the machine. The generated emf Ea

is now smaller than the terminal voltage Va, the armature current is in the opposite

direction to that in a generator, and the electromagnetic torque is in the direction to

sustain rotation of the armature.

In shunt- and separately-excited motors, the field flux is nearly constant.

Consequently, increased torque must be accompanied by a very nearly proportional

increase in armature current and hence by a small decrease in counter emf Ea to

allow this increased current through the small armature resistance. Since counter emf

is determined by flux and speed (Eq. 7.4), the speed must drop slightly. Like the

squirrel-cage induction motor, the shunt motor is substantially a constant-speed

motor having about 6 percent drop in speed from no load to full load. A typical

speed-torque characteristic is shown by the solid curve in Fig. 7.6. Starting torque

and maximum torque are limited by the armature current that can be successfully

commutated. An outstanding advantage of the shunt motor is ease of speed control.

With a rheostat in the shunt-field circuit, the field current and flux per pole can be

varied at will, and variation of flux causes the inverse variation of speed to maintain

counter emf approximately equal to the impressed terminal voltage. A maximum

speed range of about 4 or 6 to 1 can be obtained by this method, the limitation again

being commutating conditions. By variation of the impressed armature voltage, very

wide speed ranges can be obtained.

In the series motor, increase in load is accompanied by increases in the armature

current and mmf and the stator field flux (provided the iron is not completely

saturated). Because flux increases with load, speed must drop in order to maintain

the balance between impressed voltage and counter emf; moreover, the increase in

armature current caused by increased torque is smaller than in the shunt motor

because of the increased flux. The series motor is therefore a varying-speed motor

with a markedly drooping speed-torque characteristic of the type shown in Fig. 7.6.

For applications requiting heavy torque overloads, this characteristic is particularly

advantageous because the corresponding power overloads are held to more

reasonable values by the associated speed drops. Very favorable starting

characteristics also result from the increase in flux with increased armature current.

In the compound motor, the series field may be connected either cumulatively, so

that its mmf adds to that of the shunt field, or differentially, so that it opposes. The

differential connection is rarely used. As shown by the broken-dash curve in Fig. 7.6,

a cumulatively-compounded motor has speed-load characteristics intermediate

between those of a shunt and a series motor, with the drop of speed with load

depending on the relative number of ampere-tums in the shunt and series fields. It

does not have the disadvantage of very high light-load speed associated with a series

motor, but it retains to a considerable degree the advantages of series excitation.

The application advantages of dc machines lie in the variety of performance

characteristics offered by the possibilities of shunt, series, and compound excitation.

Some of these characteristics have been touched upon briefly in this section. Still

greater possibilities exist if additional sets of brushes are added so that other voltages

can be obtained from the commutator. Thus the versatility of dc-machine systems and

their adaptability to control, both manual and automatic, are their outstanding

features.

Speed Control

The three most common speed-control methods for dc motors are adjustment of the

flux, usually by means of field-current control, adjustment of the resistance

associated with the armature circuit, and adjustment of the armature terminal voltage.

Field-Current Control In part because it involves control at a relatively low power level (the power into the

field winding is typically a small fraction of the power into the armature of a dc

machine), field-current control is frequently used to control the speed of a dc motor

with separately excited or shunt field windings. The equivalent circuit for a

separately excited dc machine is shown in Fig.1. The method is, of course, also

applicable to compound motors. The shunt field current can be adjusted by means of

a variable resistance in series with the shunt field. Alternatively, the field current can

be supplied by power-electronic circuits which can be used to rapidly change the

field current in response to a wide variety of control signals.

Figure 11.2a shows in schematic form a switching scheme for pulse-width

modulation of the field voltage. This system closely resembles the pulse-width

modulation system discussed in Section 10.3.2. It consists of a rectifier which

rectifies the ac input voltage, a dc-link capacitor which filters the rectified voltage,

producing a dc voltage Vdc, and a pulse-width modulator. In this system, because

only a unidirectional field current is required, the pulsewidth modulator consists of a

single switch and a free-wheeling diode rather than the more complex four-switch

arrangement of Fig. 10.45. Assuming both the switch and diode to be ideal, the

average voltage across the field winding will be equal to

Vf = DVdc

where D is the duty cycle of the switching waveform; i.e., D is the fraction of time

that the switch S is on.

Figure 11.2b shows the resultant field current. Because in the steady-state the

average voltage across the inductor must equal zero, the average field current If will

thus be equal to

Thus, the field current can be controlled simply by controlling the duty cycle of the

pulse-width modulator. If the field-winding time constant Lf /Rf is long compared

to the switching time, the ripple current if will be small compared to the average

current If.

Armature-Circuit Resistance Control Armature-circuit resistance control provides a means of obtaining reduced speed by

the insertion of external series resistance in the armature circuit. It can be used with

series, shunt, and compound motors; for the last two types, the series resistor must be

connected between the shunt field and the armature, not between the line and the

motor. It is a common method of speed control for series motors and is generally

analogous in action to wound-rotor-induction-motor control by the addition of

external series rotor resistance.

Depending upon the value of the series armature resistance, the speed may vary

significantly with load, since the speed depends on the voltage drop in this resistance

and hence on the armature current demanded by the load. For example, a 1200-r/min

shunt motor whose speed under load is reduced to 750 r/min by series armature

resistance will return to almost 1200-r/min operation if the load is removed because

the no-load current produces a voltage drop across the series resistance which is

insignificant. The disadvantage of poor speed regulation may not be important in a

series motor, which is used only where varying-speed service is required or can be

tolerated.

A significant disadvantage of this method of speed control is that the power loss

in the external resistor is large, especially when the speed is greatly reduced. In fact,

for a constant-torque load, the power input to the motor plus resistor remains

constant,

while the power output to the load decreases in proportion to the speed. Operating

costs are therefore comparatively high for lengthy operation at reduced speeds.

Because

of its low initial cost however, the series-resistance method (or the variation of it

discussed in the next paragraph) will often be attractive economically for applications

which require only short-time or intermittent speed reduction. Unlike field-current

control, armature-resistance control results in a constant-torque drive because both

the field-flux and, to a first approximation, the allowable armature current remain

constant as speed changes.

A variation of this control scheme is given by the shunted-armature method,

which may be applied to a series motor, as in Fig. 11.3a, or a shunt motor, as in

Fig. 11.3b. In effect, resistors R1 and R2 act as a voltage divider applying a reduced

voltage to the armature. Greater flexibility is possible because two resistors can now

be adjusted to provide the desired performance. For series motors, the no-load speed

can be adjusted to a finite, reasonable value, and the scheme is therefore applicable to

the production of slow speeds at light loads. For shunt motors, the speed regulation in

the low-speed range is appreciably improved because the no-load speed is definitely

lower than the value with no controlling resistors.

Armature-Terminal Voltage Control Armature-terminal voltage control can be readily accomplished with the use of

power-electronic systems. Figure 11.4 shows in somewhat schematic form three

possible configurations. In Fig. 11.4a, a phase-controlled rectifier in combination

with a dc link filter capacitor can be used to produce a variable dc-link voltage which

can be applied directly to the armature terminals of the dc motor.

In Fig. 11.4b, a constant dc-link voltage is produced by a diode rectifier in

combination with a dc-link filter capacitor. The armature terminal voltage is then

varied by a pulse-width modulation scheme in which switch S is alternately opened

and closed.

When switch S is closed, the armature voltage is equal to the dc-link voltage Vdc,

and when the switch is opened, current transfers to the freewheeling diode,

essentially setting the armature voltage to zero. Thus the average armature voltage

under this condition is equal to

Va = D Vdc

where

Va = average armature voltage (V)

Vdc = dc-link voltage (V)

D = PWM duty cycle (fraction of time that switch S is closed)

Figure 11.4c shows an H-bridge configuration as is discussed in the context of

inverters in Section 10.3.3. Note that if switch $3 is held closed while switch $4

remains open, this configuration reduces to that of Fig. 11.4b. However, the H-bridge

configuration is more flexible because it can produce both positive- and negative

polarity armature voltage. For example, with switches S 1 and $3 closed, the

armature voltage is equal to V~c while with switches $2 and $4 closed, the armature

voltage is equal to -Vdc. Clearly, using such an H-bridge configuration in

combination with an appropriate choice of control signals to the switches allows this

PWM system to achieve any desired armature voltage in the range -V~c < Va < V~c.

Armature-voltage control takes advantage of the fact that, because the voltage

drop across the armature resistance is relatively small, a change in the armature

terminal voltage of a shunt motor is accompanied in the steady state by a

substantially

equal change in the speed voltage. With constant shunt field current and hence field

flux, this change in speed voltage must be accompanied by a proportional change in

motor speed. Thus, motor speed can be controlled directly by means of the armature

terminal voltage.

Frequently the control of motor voltage is combined with field-current control

in order to achieve the widest possible speed range. With such dual control, base

speed can be defined as the normal-armature-voltage, full-field speed of the motor.

Speeds above base speed are obtained by reducing the field current; speeds below

base speed are obtained by armature-voltage control. As discussed in connection with

field-current control, the range above base speed is that of a constant-power drive.

The range below base speed is that of a constant-torque drive because, as in

armatureresistance control, the flux and the allowable armature current remain

approximately constant. The overall output limitations are therefore as shown in Fig.

11.6a for approximate allowable torque and in Fig. 11.6b for approximate allowable

power. The constant-torque characteristic is well suited to many applications in the

machine tool industry, where many loads consist largely of overcoming the friction

of moving parts and hence have essentially constant torque requirements.

The speed regulation and the limitations on the speed range above base speed are

those already presented with reference to field-current control; the maximum speed

thus does not ordinarily exceed four times base speed and preferably not twice base

about one-tenth of base speed, corresponding to a total maximum-to-minimum range

not exceeding 40:1.

With armature reaction ignored, the decrease in speed from no-load to full-load

torque is caused entirely by the full-load armature-resistance voltage drop in the

dc generator and motor. This full-load armature-resistance voltage drop is constant

over the voltage-control range, since full-load torque and hence full-load current

are usually regarded as constant in that range. When measured in r/min, therefore,

the speed decrease from no-load to full-load torque is a constant, independent of the

no-load speed, as we saw in Example 11.3. The torque-speed curves accordingly are

closely approximated by a series of parallel straight lines for the various motor-field

adjustments. Note that a speed decrease of, say, 40 r/min from a no-load speed of

1200 r/min is often of little importance; a decrease of 40 r/min from a no-load speed

of 120 r/min, however, may at times be of critical importance and require corrective

steps in the layout of the system.

Figure 11.7 shows a block diagram of a feedback-control system that can be used

to regulate the speed of a separately excited or shunt-connected dc motor. The inputs

to the dc-motor block include the armature voltage and the field current as well as

the load torque Tload. The resultant motor speed Wm is fed back to a controller block

which represents both the control logic and power electronics and which controls the

armature voltage and field current applied to the dc motor, based upon a reference

speed signal Wref. Depending upon the design of the controller, with such a scheme

it is possible to control the steady-state motor speed to a high degree of accuracy

independent of the variations in the load torque.

In the case of permanent-magnet dc motors, the field flux is, of course, fixed

by the permanent magnet (with the possible exception of any effects of temperature

changes on the magnet properties as the motor heats up). The voltage generated

voltage can be written in the form,

Ea = Kmwm

and that the electromagnetic torque can be written as

Tmech = Km Ia

Hence it can be observed that the analysis of a permanent-magnet dc motor is

identical to that of a shunt or separately excited

dc motor with the exception that the torque-constant Km must be substituted for

the term Kf If .

Pulse Width Modulation (PWM):

Pulse-width modulation (PWM), or pulse-duration modulation (PDM), is a commonly

used technique for controlling power to inertial electrical devices, made practical by

modern electronic power switches.

The average value of voltage (and current) fed to the load is controlled by turning the

switch between supply and load on and off at a fast pace. The longer the switch is on

compared to the off periods, the higher the power supplied to the load is.

The PWM switching frequency has to be much faster than what would affect the load,

which is to say the device that uses the power. Typically switching’s have to be done

several times a minute in an electric stove, 120 Hz in a lamp dimmer, from few

kilohertz (kHz) to tens of kHz for a motor drive and well into the tens or hundreds of

kHz in audio amplifiers and computer power supplies.

The term duty cycle describes the proportion of 'on' time to the regular interval or

'period' of time; a low duty cycle corresponds to low power, because the power is off

for most of the time. Duty cycle is expressed in percent, 100% being fully on.

The main advantage of PWM is that power loss in the switching devices is very low.

When a switch is off there is practically no current, and when it is on, there is almost no

voltage drop across the switch. Power loss, being the product of voltage and current, is

thus in both cases close to zero. PWM also works well with digital controls, which,

because of their on/off nature, can easily set the needed duty cycle.

PWM has also been used in certain communication systems where its duty cycle has

been used to convey information over a communications channel.

Principle:

Pulse-width modulation uses a rectangular pulse wave whose pulse width is modulated

resulting in the variation of the average value of the waveform. If we consider a pulse

waveform with a low value , a high value and a duty cycle D (see

figure 1), the average value of the waveform is given by:

As is a pulse wave, its value is for and

for . The above expression then becomes:

This latter expression can be fairly simplified in many cases where

as . From this, it is obvious that the average value of the signal ( ) is

directly dependent on the duty cycle D.

The simplest way to generate a PWM signal is the intersective method, which requires

only a saw tooth or a triangle waveform (easily generated using a simple oscillator) and

a comparator. When the value of the reference signal (the red sine wave in figure 2) is

more than the modulation waveform (blue), the PWM signal (magenta) is in the high

state, otherwise it is in the low state.

There are many forms of modulation used for communicating information. When a high

frequency signal has amplitude varied in response to a lower frequency signal we have

AM (amplitude modulation). When the signal frequency is varied in response to the

modulating signal we have FM (frequency modulation). These signals are used for radio

modulation because the high frequency carrier signal is needs for efficient radiation of

the signal. When communication by pulses was introduced, the amplitude, frequency

and pulse width become possible modulation options. In many power electronic

converters where the output voltage can be one of two values the only option is

modulation of average conduction time. 1 . Linear Modulation

The simplest modulation to interpret is where the average ON time

of the pulses varies proportionally with the modulating signal. The advantage of

linear processing for this application lies in the ease of de-modulation. The

modulating signal can be recovered from the PWM by low pass filtering. For a

single low frequency sine wave as modulating signal modulating the width of a

fixed frequency (fs) pulse train the spectra is as shown in Fig 1.2. Clearly a low

pass filter can extract the modulating component fm.

2. Sawtooth PWM

The simplest analog form of generating fixed frequency PWM is by

comparison with a linear slope waveform such as a saw tooth. As seen in Fig 1.2

the output signal goes high when the sine wave is higher than the saw tooth. This

is implemented using a comparitor whose output voltage goes to logic HIGH when

ne input is greater than the other. Other signals with straight edges can be used

for modulation a rising ramp carrier will generate PWM with Trailing Edge

Modulation.

Fig.1.4 Trailing Edge

It is easier to have an integrator with a reset to generate the ramp in Fig1.4 but the

modulation is inferior to double edge modulation.

3. Regular Sampled PWM

The scheme illustrated above generates a switching edge at the instant of crossing of the

sine wave and the triangle. This is an easy scheme to implement using analog

electronics but suffers the imprecision and drift of all analog computation as well as

having difficulties of generating multiple edges when the signal has even a small added

noise. Many modulators are now implemented digitally but there is difficulty is

computing the precise intercept of the modulating wave and the carrier. Regular

sampled PWM makes the width of the pulse proportional to the value of the modulating

signal at the beginning of the carrier period. In Fig 1.5 the intercept of the sample

values with the triangle determine the edges of the Pulses. For a saw tooth wave of

frequency fs the samples are at 2fs.

There are many ways to generate a Pulse Width Modulated signal

other than fixed frequency sine sawtooth. For three phase systems the modulation

of a Voltage Source Inverter can generate a PWM signal for each phase leg by

comparison of the desired output voltage waveform for each phase with the same

sawtooth. One alternative which is easier to implement in a computer and gives a

larger modulation depth is using space vector modulation.

‘4. Modulation Depth

Fig.1.6 Saturated Pulse Width Modulation

For a single phase inverter modulated by a sine – saw tooth

Comparison , if we compare a sine wave of magnitude from -2 to +2 with a triangle

From -1 to +1 the linear relation between the input signal and the average output

signal will be lost. Once the sine wave reaches the peak of the triangle the pulses

will be of maximum width and the modulation will then saturate. The Modulation

depth is the ratio of the current signal to the case when saturation is just starting.

Thus sine wave of peak 1.2 compared with a triangle with peak 2.0 will have a

modulation depth of m=0.6.

TIMER

A timer is a specialized type of clock. A timer can be used to control the sequence of an

event or process. Whereas a stopwatch counts upwards from zero for measuring elapsed

time, a timer counts down from a specified time interval, like an hourglass. Timers can

be mechanical, electro mechanical ,electronic (quartz), or even software as all

modern computers include digital timers of one kind or another. When the set period

expires some timers simply indicate so (e.g., by an audible signal), while others operate

electrical switches, such as a time switch, which cuts electrical power.

Electronic timers

Electronic timers are essentially quartz clocks with special electronics, and can achieve

higher precision than mechanical timers. Electronic timers have digital electronics, but

may have an analog or digital display. Integrated circuits have made digital logic so

inexpensive that an electronic timer is now less expensive than many mechanical and

electromechanical timers. Individual timers are implemented as a simple single-

chip computer system, similar to a watch and usually using the same, mass-produced,

technology.

Many timers are now implemented in software. Modern controllers use a programmable

logic controller rather than a box full of electromechanical parts. The logic is usually

designed as if it were relays, using a special computer language called ladder logic. In

PLCs, timers are usually simulated by the software built into the controller. Each timer

is just an entry in a table maintained by the software.

Digital timers are used in safety devices such as a gas timer

Computer timers

Computer systems usually have at least one timer. These are typically

digital counters that either increment or decrement at a fixed frequency, which is often

configurable, and which interrupt the processor when reaching zero, or alternatively a

counter with a sufficiently large word size that it will not reach its counter limit before

the end of life of the system.

More sophisticated timers may have comparison logic to compare the timer value

against a specific value, set by software , that triggers some action when the timer value

matches the preset value. This might be used, for example, to measure events or

generate pulse width modulated waveforms to control the speed of motors (using a class

D digital electronic amplifier).

As the number of hardware timers in a computer system or processor is finite and

limited, operating systems and embedded systems often use a single hardware timer to

implement an extensible set of software timers. In this scenario, the hardware timer's

interrupt service routine would handle house-keeping and management of as many

software timers as are required, and the hardware timer would be set to expire when the

next software timer is due to expire. At expiry, the interrupt routine would update the

hardware timer to expire when the next software timer is due, and any actions would be

triggered for the software timers that had just expired. Expired timers that are

continuous would also be reset to a new expiry time based on their timer interval, and

one-shot timers would be disabled or removed from the set of timers. While simple in

concept, care must be taken with software timer implementation if issues such as timer

drift and delayed interrupts is to be minimized.

555 TIMER IC The 555 timer IC is an integrated circuit (chip) used in a variety of timer, pulse

generation, and oscillator applications. The 555 can be used to provide time delays, as

an oscillator, and as a flip-flop element. Derivatives provide up to four timing circuits in

one package.

Introduced in 1971 by Signetics , the 555 is still in widespread use, thanks to its ease of

use, low price, and good stability. It is now made by many companies in the original

bipolar and also in low-power CMOS types. As of 2003, it was estimated that 1

billion units are manufactured every year

Design

The IC was designed in 1971 by Hans R. Camenzind under contract to Signetics, which

was later acquired by Philips.

Depending on the manufacturer, the standard 555 package includes 25 transistors,

2 diodes and 15 resistors on a silicon chip installed in an 8-pin mini dual-in-line

package (DIP-8).[2]

Variants available include the 556 (a 14-pin DIP combining two

555s on one chip), and the two 558 & 559s (both a 16-pin DIP combining four slightly

modified 555s with DIS & THR connected internally, and TR is falling edge sensitive

instead of level sensitive). There is no 557.

The NE555 parts were commercial temperature range, 0 °C to +70 °C, and the SE555

part number designated the military temperature range, −55 °C to +125 °C. These were

available in both high-reliability metal can (T package) and inexpensive epoxy plastic

(V package) packages. Thus the full part numbers were NE555V, NE555T, SE555V,

and SE555T. It has been hypothesized that the 555 got its name from the three

5 kΩ resistors used within,[3]

but Hans Camenzind has stated that the number was

arbitrary.[1]

Low-power versions of the 555 are also available, such as the 7555 and CMOS

TLC555.[4]

The 7555 is designed to cause less supply noise than the classic 555 and the

manufacturer claims that it usually does not require a "control" capacitor and in many

cases does not require a decoupling capacitor on the power supply. Such a practice

should nevertheless be avoided, because noise produced by the timer or variation in

power supply voltage might interfere with other parts of a circuit or influence its

threshold voltages.

Usage

The 555 monolithic timing circuit is a highly stable controller capable of producing

accurate time delays, or oscillation. In the time delay mode of operation, the time is

precisely controlled by one external resistor and capacitor. For a stable operation as an

oscillator, the free running frequency and the duty cycle are both accurately controlled

with two external resistors and one capacitor. The circuit may be triggered and reset on

falling waveforms, and the output structure can source or sink up to 200 mA.

Pin 1: Ground. All voltages are measured with respect to this terminal.

Pin 2: Trigger. The output of the timer depends on the amplitude of

the external trigger pulse applied to this pin. The output is low if the voltage at

this pin is greater than 2/3 VCC. When a negative going pulse of amplitude

greater than 1/3 VCC is applied to this pin, comparator 2 output goes low, which

in turn switches the output of the timer high. The output remains high as long as

the trigger terminal is held at a low voltage.

Pin 3: Output. There are two ways by which a load can be

connected to the output terminal: either between pin 3 and ground or between

pin3 and supply voltage +VCC. When the output is low the load current flows

through the load connected between pin3 and +VCC into the output terminal and

is called sink current. The current through the grounded load is zero when the

output is low. For this reason the load connected between pin 3 and +VCC is

called the normally on load and that connected between pin 3 and ground is

called normally off-load. On the other hand, when the output is high the current

through the load connected between pin 3 and +VCC is zero. The output terminal

supplies current to the normally off load. This current is called source current.

The maximum value of sink or source current is 200mA.

Pin 4: Reset. The 555 timer can be reset (disabled) by applying a

negative pulse to this pin. When the reset function is not in use, the reset

terminal should be connected to +VCC to avoid any possibility of false triggering.

Pin 5: Control. An external voltage applied to this terminal changes

the threshold as well as trigger voltage. Thus by imposing a voltage on this pin

or by connecting a pot between this pin and ground, the pulse width of the

output waveform can be varied. When not used, the control pin should be

bypassed to ground with a 0.01μF Capacitor to prevent any noise problems.

Pin 6: Threshold. This is the non-inverting input of comparator 1,

which monitors the voltage across the external capacitor. When the voltage at

this pin is greater than or equal to the threshold voltage 2/3 VCC, the output of

comparator 1 goes high, which inturn switches the output of the timer low.

Pin 7: Discharge. This pin is connected internally to the collector of

transistor Q1. When the output is high Q1 is OFF and acts as an open circuit to

external capacitor C connected across it. On the other hand, when the output is

low, Q1 is saturated and acts as a short circuit, shorting out the external

capacitor C to ground.

Pin 8: +VCC. The supply voltage of +5V to + 18V is applied to this pin

with respect to ground.