Six-Step Twelve-Step Current-Source Inverters DC Side ...koasas.kaist.ac.kr/bitstream/10203/6838/1/[1981]Novel Six-Step and... · CHOANDPARK:CURRENT-SOURCEINVERTERS Line F i e i Fig.

IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. IA-17, NO. 5, SEPTEMBER/OCTOBER 1981

Novel Six-Step and Twelve-Step Current-Source Inverterswith DC Side Commutation and Energy Rebound

GYU H. CHO, MEMBER, IEEE, AND SONG B. PARK, MEMBER, IEEE

Abstract-Novel six-step and twelve-step current-source inverters(CSI) with dc side commutation and energy rebound capability arepresented with detailed explanation of the circuit operation. Theproposed inverters can operate in a very wide range of frequency andload variation by employing dc side commutation. Also, the energyrebound makes the use of low voltage silicon controlled rectifiers(SCR's) possible and increases the inverter efficiency. Unlike the dualcurrent-source inverter, one auxiliary inverter with a power re-quirement of about one-half that of the main inverter is simply addedto the six-step CSI in order to obtain a twelve-step CSI. Motoroperation is possible in four quadrants in both six- and twelve-stepinverters. The advantages of the proposed CSI's over conventionalones are described, and experimental results are given in oscillo-grams.

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I. INTRODUCTION

THE ADVANTAGES of the current-source inverter (CSI)over the voltage inverter are well known. The most widely

used CSI is probably the autosequential commutated inverter(ASCI) [1] -[3]. Recently, another type of CSI, namely, thethird harmonic auxiliary commutated inverter (THACI), waspublished. Their configurations are shown in Fig. 1(a) and(b), respectively.

In THACI, which has a simpler configuration than ASCI,stable operation is secured in both the motoring and thegenerating regions by employing the delayed gating method[4]. However, many factors such as the torque loss, harmonicpower, and spike voltage depend on the amount of delay andthe capacitance value; also, the optimum control of the delayunder various operating conditions seems to be not so simple.In addition, the allowable range of load variation and its oper-ating frequency in THACI are about one-third those of ASCI.In a specific or limited range operation, however, THACI maybe a good choice for its simplicity in configuration.

The wide difference in the operating range between THACIand ASCI is caused mainly by the difference of the paths ofthe dc current flowing through the capacitor during commuta-tion; namely, this current passes through the load in ASCIwhile it bypasses the load in THACI. Therefore, the operatingrange becomes narrow in THACI due to the capacitor chargingtime in addition to the delayed gating. This may be the com-mon problem in all circuit configurations where the currentthrough the commutating capacitor bypasses the load.

Paper IPCSD 80-8, approved by the Static Power Converter Com-mittee of the IEEE Industry Applications Society for publication inthis TRANSACTIONS. Manuscript released for publication May 19,1981.

The authors are with the Department of Science, Korea AdvancedInstitute of Science, P. 0. Box 150, Chongryangni, Seoul, Korea.

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In contrast, the current through the commutating capacitorin ASCI delivers power to the load, which occurs smoothlyeven in the consecutive commutation where a new commuta-tion starts at the end of one. A wide range of operation isthus possible in ASCI. However, in this case, the operatingrange as well as the spike voltage are also functions of loadand capacitance value, since commutation and phase-to-phasecurrent changes occur at the same time while the commutatingcapacitors are discharged and charged by the dc current. Thesesituations necessitate some compromise in the choice of thecapacitance value among the various factors involved. Anothershortcoming is that it requires a number of commutatingcapacitors and high power diodes and/or high voltage siliconcontrolled rectifiers (SCR's).

In the inverter scheme proposed in this paper, the operatingrange, which is limited in ASCI and THACI for the reasondescribed above, is extended by employing a different com-mutating scheme, namely the dc side commutating (dc-sc)

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CHO AND PARK: CURRENT-SOURCE INVERTERS

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Fig. 2. Block diagram of proposed six-step current-source inverter.

method. In the new dc-sc circuit, not only the polarity of thecommutating capacitor reverses at high speed for turning offthe main SCR's and eliminating the time of capacitor recharg-ing, but also the regenerative capability of the inverter is pre-served. Also employed in the new inverter is a novel energyrebounding method, the main purpose of which is to limit thepeak voltage, making possible the use of low voltage SCR's.

Combining the dc-sc and the energy rebound circuits insuch a way that their respective functions be kept independ-ently, a novel six-step CSI is obtained, which allows bothmotoring and regenerating operations. A new and economictwelve-step CSI will then be presented in which one auxiliaryinverter with a power requirement of about one-half that ofthe main inverter is added while the dc-sc circuit and the en-ergy rebound (ER) circuit operate for both inverters. Proto-types of both six-step and twelve-step inverters have been de-signed, constructed, and tested; oscillograms of the experimen-tal results are shown.

II. A NEW SIX-STEP CURRENT SOURCE INVERTER

A. General

It is not a new idea in commutating circuits to reverse acapacitor voltage at high speed and thereby turn off associatedSCR's; actually this method is employed in the McMurrayinverter, chopper, and other circuits. However, several con-siderations must be made to apply this scheme to CSI's with-out loss of their advantages. Fig. 2 shows the block diagram ofthe new six-step CSI, while Fig. 3 shows its complete circuitdiagram. The distinctive features of the new inverter are, ascan be seen from these figures, employment of the dc-sc, andaddition of the peak voltage limit and ER circuit. The dc-scmethod has relative merits and demerits as compared with thebranch commutating method [5] -[6]. In the conventionaldc-sc inverters a free-wheeling diode is simply connected back-to-back to each main SCR. This simple connection may failwhen applied directly to the CSI, since regenerative capability,one of its advantages, will then be lost. However, the free-wheeling diodes can not be eliminated, since they play a cru-cial role in the commutating transient, during which phase-to-phase change of the motor current occurs right after themain SCR's are blocked by the operation of the commutatingcircuit. Also, the associated high spike voltage is inevitable.This necessitates some means to limit the spike voltage when

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rectifiers and energy rebound circuit. (b) DC side commutating cir-cuit and inverter.

the dc-sc method is applied to CSI's. In order to limit the spikevoltage and at the same time increase the inverter efficiency,the spike energy must be fed back to the source. This and theconnection of free-wheeling diodes have to be done withoutaffecting the regenerative mode operation of the motor. Forthis purpose a new ER circuit has been devised. With thiscircuit and a simple control, it is possible to increase the in-verter efficiency using low voltage SCR's.

B. OperationFig. 4 shows the sequence of topology modes starting with

a state where SCR's T1 and T2 are conducting. Typical wave-forms of voltages across capacitors C1, C2, and C4, as wellas currents through C2, diode D 1 , and phase currents areshown in Fig. 5, in which the various intervals correspond to

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Fig. 4. Sequence of topology modes of six-step CSI.

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Fig. 5. Voltage and current waveforms for each mode. Intervals A, B,* -, I corresponds to modes (a), (b), , (i) in Fig. 4.

respective topology modes. (The operation of the ER circuit,which is included in Fig. 4 as a block, will be described later inconnection with Fig. 8.)

The initial state with T1 and T2 conducting is shown as

interval A. Interval B starts with SI turned on. The voltageacross C2 reverses polarity and SI turns off before the end ofinterval B. S2 is turned on at the start of interval C, whichplaces C1 and C2 in series with L1 and S2. When voltageacross C1 becomes zero, interval D begins. The new conduc-ting paths from the main and commutating buses are now

formed with diodes D4, D5, D7 -D10 turned on.

The forward drops of diodes D7 and D8 determine thecutoff of T1 and T2 which is the start of interval E when alldiodes D1 to D6 conduct through the commutating bus andD7 and D8. The current flowing through L1 in interval Econsists of the dc current from the main bus and the currentfrom the commutating bus which contains the load current.Therefore, the commutating circuit must be capable of hand-ling twice the dc current. The voltage across C2 becomes posi-tive again by the inductive current through L1, which then de-creases with the commutating bus current. As the bus currentdecreases to the load current, T2 turns on again and intervalF starts. In interval F, the current through the commutatingbus continues to decrease. After it becomes zero, Cl beginscharging as the L1 current decreases below the load current,when interval G starts. In interval G, T3 is newly turned on

and the sum of the current through C1 and L1 is almost equalto Id, where the maximum rate of dv/dt is given by Id/Cl.This value can be large as long as it does not exceed thedv/dt rating of the main SCR's at full load. Actually, C1 can

be much smaller than C2 but the value must be determinedconsidering some other factors involved. When the voltage on

Cl equals the voltage on C4, D8 and Dg stop conducting, thenD3 and D1 start conducting and charging capacitor C4 ininterval H.

Specifically, the phase-to-phase current, which has keptalmost constant until the end of interval G, starts changing asit charges capacitor C4. Thus, the capacitor voltage continuesto rise very slowly, if C4 is sufficiently large, until the phaseAcurrent becomes zero and the phase B current reaches Id.After the end of the transition state, T3 maintains a constantconduction current as T2 does; this state continues until thenext commutation begins.

For convenience, S3 is turned on during interval H to sup-ply any energy loss (from the dc source consisting of D1 6,D1 7, D1 8, and C3) to capacitor C2 in order to insure stableand continuous operation. The control sequence for the abovecycle is S1 turned on (interval B); S2 turned on (interval C);T2 turned on again (interval F); T3 turned on (interval G);and S3 turned on in any interval G to I.

Figs. 6 and 7 show the sketches and oscillograms, respec-tively, of the voltage and current waveforms during the com-plete cycle of operation for the six-step inverter. As is seenfrom Fig. 7, the voltage spikes are limited but still relativelyhigh because the experiment was done at low voltage. Thisfact will further be examined later in connection with the ERcircuit operation.

C. Distinctive Features of the DC-SC Circuit

The dc-sc circuit as described in the above has severaladvantages over the conventional ones. First, a wide range ofoperating frequency is possible due to the fact that the com-mutating interval is determined mainly by the SCR turn-offtime and is almost independent of the load current, since thevoltage on C2 reverses at high speed and a small capacitancesuffices for C1 .

Second, a very wide range of load variation is allowed.This can be explained as follows. The voltage on C2, which isnecessary for commutation, is determined by the voltage onC3 due to the role of S3. The voltage on C3 is maintained al-most always at a maximum level, with a slight variation de-pending on the converter output voltage, since it is suppliedby the incomplete bridge rectifier consisting of diodes D1 6-D1 8 and SCR's P2, P4, and P6. Thus for simplicity, C3 is re-placed by a dc source in Fig. 4. As a result, stable operationis secured even at low voltage and high current such as in themotor starting. Even in the extreme load conditions, open orshorted load, the commutating circuit operates independently,and temporary failure such as caused by noise turn-on of themain SCR's can be immediately recovered.

The third distinctive feature of the proposed dc-sc circuitis that it does not affect the regenerative operation of the in-verter. Specifically, the commutating circuit is isolated fromthe rest of the system when the motor operates as a generator,since then the voltage polarity of the p side in Fig. 3 becomesminus while that of the n side becomes plus. Hence, diodes Dgand D1 0 are turned off. In other words, by the presence ofDgand D10, both motoring and generating operations are pos-sible; at commutating operation they are conducted to turnoff the main SCR's, and at regenerative operation they are

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(c)waveform sketches for new six-step in-(b) Line-to-line current. (c) Line-to-line

Fig. 7. Oscillograms of line-to-line voltage (top) and phase current(bottom). Vertical: 50 V/div (top), 5 A/div (bottom). Horizontal:10 Ms/div (Load: Induction motor of S hp, 15 A, 220 V (34, 60 Hz),17S0 r/min).

blocked to isolate the commutating circuit from the rest ofthe system.

III. VOLTAGE LIMIT AND ENERGY REBOUND CIRCUIT

A. Operation

In the proposed system, the phase currents do not changeappreciably until interval G; they change only after the turn-off of the main SCR's. A desirable situation in this case isthat the line-to-line voltage reaches a high value in as short a

time as possible, and at the same time the peak voltage belimited to an appropriate value in relation to the input voltagein order to make a rapid and safe phase-to-phase currentchange. The purpose of the ER circuit is to store temporarilyand then feed back to the source, the excess energy resultingfrom the peak voltage limiting action at each commutation.This feedback loop is initiated by an excess voltage level,therefore the circuit 'provides voltage limiting as well as energy

feedback.

Fig. 8 shows the sequence of topology modes of operationsfor the ER circuit. A large capacitor C4 is used for temporarystorage of the excess energy, which is connected to the com-mutating system through D, 1. The behavior of C4 was de-scribed previously in connection with interval H of Fig. 4. Inthe following we will examine the operation of the ER circuitin more detail, referring to the topology modes.

In interval Xl, the capacitors are charged with the polaritiesshown. When the voltage on C4 becomes sufficiently high, S4and S5 are turned on simultaneously, which reverses the volt-ages on C5 and C6 very rapidly through L3 and L4, respec-tively (interval X2). As the sum of the voltages on C4, C5, andC6 exceeds the source voltage, interval X3 begins conductingthrough Dl 4 and DI s, feeding back the excess energy to thesource while discharging C4. The turn-off time of54 and SS ismade sufficiently long by using an inductance Lr (=L,l +Lr2) much larger than L3 (=L4). When C5 and C6 are chargedin the same direction as in interval Xl and each of their volt-ages reaches that of C4, the magnetic energy stored in inductorLr, which has not been discharged fully, recharges C4 throughD1 2 and DI 3; in this way rebounding continues (interval X4).After all the stored energy in L4 has been discharged, the cycleis in the state of interval XI and the process repeats. Again ininterval Xl, DI14 and D1 5 are blocked, preventing the dis-charge of C5 and C6 for the continuation of stable operationof the next cycle. Faster energy rebounding can be carried out,if necessary, by turning on S4 and S5 before the currentthrough Lr becomes zero in interval X4, in which case, how-ever, turn-off of the SCR's must be taken into consideration.

Energy rebounding occurs discretely, and the voltage dropof C4 per pumping depends on its voltage and the size of C5and C6. A possible control reference voltage Vref to limit thespikes is shown in Fig. 9, where Vi-f changes linearly from VIto V2 (Vl < Vyn < V2; Vm = maximum source voltage) inthe full range of the source voltage V8. For a given sourcevoltage V81, if the instantaneous voltage on C4 exceeds Vrj,S5 and S6 are fired; otherwise not. VI, the reference voltagefor V8 = 0, should be high enough, say above 50 V, in orderto get a sharp rise and fall in the load current waveform duringthe commutation transient at low source voltage. This resultsin a relative amplitude of the spike voltages being high at lowsource voltage, while low at high source voltage. In this simplecontrol there is no need of taking the motor operating condi-tion into consideration. Fig. 10 shows the oscillogram of theexperimental ER circuit. We see that the start of energy re-bounding and rapid inversion of the voltage on C6 (and C5)occur simultaneously as soon as the voltage on C4 exceeds areference voltage due to the entering of the load current ateach commutating instant. We also aware that the voltage onC4 discharges through the source without accompaning anycurrent spikes.

B. Distinctive Features

As mentioned earlier, the ER circuit is essential in the pro-posed system in order to obtain a wide range of operationsby having commutation and phase-to-phase changes of theload current occur in different intervals. The addition of theER circuit, however, does not increase the overall system cost;it may actually reduce it. First, the main SCR's, which are

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perhaps the most expensive components of the system, can

have low voltage ratings since the peak voltages across themare always limited to a predetermined level irrespective of theload condition, full load or no load. Second, the ER circuitcan be built with low-power components, since the power tobe handled in this circuit is estimated to be less than 10 per-

cent of the total input power of the system. Specifically, a

small size inductor sufficies for 4, since the current flowingthrough it is small; also S4, Ss, and all diodes in this circuitmay have very small ratings compared with the main SCR's.C4 is the only exception; it should have a relatively largecapacitance. A high ripple electrolytic capacitor (or paral-lel connection of several such capacitors), which is ofrelatively small size and low cost, can be used for C4 since it is

always charged with the same polarity. However, some specialconsiderations must be taken for full load and high frequencyoperation when the ripple current flowing through the capaci-tor becomes quite high. Finally, gating control of the SCR'sin the energy rebound circuit is achieved in a simple manneras stated earlier. All this accounts for the overall economy ofthe proposed system.

The regenerative mode of operation is made possible bymerely interrupting the firing of S4 and Ss, in which case C5and C6 play an excellent role as the snubbers with a stableblocking condition of SCR's being secured. The mode changebetween motoring and regenerating operations occurs veryrapidly and smoothly.

IV. NOVEL TWELVE-STEP CURRENT SOURCEINVERTER

A. General

We have described a six-step CSI designed with a newapproach. One disadvantage of such a single current-sourceinverter (SCSI) is that it is rich in harmonics. One method forharmonics reduction is to synthesize a twelve-step waveformby a phase-shifted addition of the output currents of twoinverters. The inverter using this method is called a dual cur-rent-source inverter (DCSI). We will introduce a new twelve-step CSI, which does not employ such a DCSI method.

Fig. 11 shows two typical DCSI's [7] -[8]. In both, theoutput currents of two complete SCSI's, such as ASCI's areshifted in phase by 30° and added in order to obtain a twelve-step waveform. In this scheme, reactors with considerablylarge inductance must be inserted between the two invertersin order to suppress the circulating current generated by themotor electromotive force (EMF). In Fig. ll(a), two pairs ofindependent reactors are inserted, each between the controlledrectifier and one of the inverters. The reactor size can be con-siderably reduced in Fig. I l(b) by using two current balancinginductors. The balancing inductor, which has two identicalwindings wound on the same core with no air gap, may be ofa small size and yet is very effective in suppressing the circulat-ing current, since the fluxes generated by the bisected dccurrent components flowing through inverters I and II canceleach other while it exhibits a large impedance for the circulat-ing current component. These systems may be useful for largepower applications, but are expensive because two independ-ent inverters are used, each having its own commutating capa-bility.

In the following we present a novel twelve-step CSI inwhich the size of current balancing inductors is further re-duced and which is obtained by adding one auxiliary inverterto the proposed six-step CSI described in the foregoing sec-tions. Fig. 12 shows the block diagram of the proposed twelve-step CSI, from which we see that one auxiliary inverter isadded, and only one commutating circuit is used effectively incommon for the main and auxiliary inverters.

B. Operation

Fig. 13 shows the detailed circuit diagram of the proposedtwelve-step CSI; only the inverter part is shown since thefront part in the block is identical to that in Fig. 3. The in-

529

IEEE TRANSACTIONS ON INDUSTRY APPLICATILONS, VOL. IA-17, NO. 5, SEPT[EMBER/OCTOBER 1981

31'AC -

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(a)

Fig. 11. Two typical configurations of twelve-step CSI.

Fig. 12. Block diagram of new twelve-step CSI.

verter part consists of a main inverter, which has the same con-

figuration as in the six-step CSI, and an auxiliary inverter,which has quite a different configuration. Fig. 14 shows thefiring sequence of the proposed twelve-step CSI, from whichwe see that the main SCR's and the auxiliary SCR's are turnedon and off altemately and a half-current step is formed in theline current in the conducting state of each auxiliary SCR.

Fig. 15 shows the conducting state at the positive half-current step for one branch of the auxiliary inverter. Eachbranch of the auxiliary inverter consists of a set of bridgediodes and a pair of inductors. As can be seen from Fig. 15,the role of the bridge diodes is twofold; it makes a currentpath through B2 I and B22 when A1 is ON and through B2 3

and B24 when A4 is ON, while is isolates points A and B,causing no magnetization of L2 1 and L2 2, when both AI andA4 are OFF. In this way L21 and L22 play the same role as

the current balancing inductor in Fig. 11 (b) when one of theauxiliary SCR's is ON. L21 and L22 are wound on the same

core with no air gap, as are (L1 1, L 2) and (L31 ,L32). Thusone more core is required compared with Fig. 1 1(b). However,the core size can be considerably reduced, since in Fig. 14current flows through the current balancing inductor duringonly one-sixth of the period, while in Fig. 1l(b) current flowcontinues through the whole period. The three sets of induc-

tors can even be wound on the same core due to the actioni ofthe bridge diodes, as described above.

Both the main and the auxiliary SCR's, which have beenin the conducting state, are turned off simultanieously by theoperation of tlhe commutatinig circuit. Diodes Da1-DDa6 canbe of a very small capacity, since they are used only to securestable turn-off of the auxiliary SCR's. They do this by forminga demagnetizing current path during the commutating initervalfor the current balancing inductor, which has been nmagnetizedby the motor EMF during the operation of the auxiliary in-verter. Also, the power rating required for the auxiliary SCR'sis about one-half that for the main SCR's, since as can be seenfrom Fig. 14, the ratio of conduction intervals is one to three,while the conduction current magnitudes are the same in thetwo sets of SCR's. Finally, the power rating required for thebridge diodes is one-half that for the auxiliary SCR's, since thecurrent through the auxiliary SCR is bisected in the bridgearms. Fig. 16 shows the oscillograms of the phase voltage andthe line current obtained with an experimental twelve-stepCsI.

C. Distinctive Features

The auxiliary inverter, which is added to the six-step CSIin order to obtain a twelve-step waveform, does not requirea separate commutating circuit; one dc side commutating cir-cuit controls simultaneously both the main and the auxiliaryinverters. The current balancing inductor in the auxiliary in-verter can be of a very small size compared with the one ofDSCI. Power rating required for the auxiliary SCR's is aboutone-half that for the main SCR's, and that for the bridgediodes is further reduced. Furthermore, all components canbe low voltage devices due to the particular peak voltagelimit and energy rebound method employed. Firing control ofthe SCR's in the proposed twelve-step CSI is simple, since onlythe alternating ON time durations for the main and auxiliarySCR's need to be considered in timing the firing pulses. Thisis different from the situation in DCSI where a rather elabo-rate firing control is needed, taking into account the differenceof the power delivered by two CSI's.

In all, the proposed twelve-step CSI has an economic ad-vantage and can be used for low and medium power appli-cations as well as for higher power applications, where theharmonics content is the main concern.

By slightly modifying the firing sequence, namely, by send-ing the gating pulses only to the main inverters, the twelve-stepCSI can be smoothly converted to a six-step CSI. This is ad-vantageous at high frequency operation of the motor, sincethe commutation frequency is halved. Fig. 17 shows smoothtransition of twelve-step to six-step conversion and vice versa.

V. CONCLUSIONNovel six-step and twelve-step inverters with dc side com-

mutation and energy rebound features were designed, con-structed, and tested in the laboratory with the expectedresults obtained. The proposed inverters have improved re-liability and capability of very wide range operations in termsof load variation and operating frequency, and they can be im-plemented with low voltage SCR's and hence at a lower over-

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LoadFig. 13. Circuit diagram of new twelve-step CSI.

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Fig. 14. Firing sequence of new twelve-step CSI.

all system cost. The twelve-step inverter is obtained simplyby adding an auxiliary inverter with a power requirementsmaller than that of the main inverter to the new six-step CSI.Also, both inverters are capable of regenerative operation.

On the other hand, there are some minor disadvantages ofthe new CSI's. Due to the presence of dc side commutationand energy rebound circuits, firing control is rather compli-cate compared with ASCI, but comparable to other inverters.Some redundant commutating loss occurs in the proposedinverters at light load, since the commutating circuit is de-signed considering the full load condition. Also, the torqueloss is expected to be considerable at higher frequencies, asthe commutation interval becomes a major part of the period.This is due to the reduction of the root-mean-square (rms)current supplied to the motor, since the voltage on the ERcircuit is higher than the controlled rectifier output.

In conclusion, the new six- and twelve-step inverters may beexcellent candidates for many ac adjustable speed drives dueto their many advantages. Detailed quantitive analyses andextensive experimental results will be published elsewhere inthe near future.

(a)

(b)Fig. 15. One branch of auxiliary inverter. (a) 'IdI2 step. (b) -IdI2

step.

ACKNOWLEDGMENTThe authors wish to thank Mr. W. C. Song for helping the

design of the control part and experimental works, and Dr.J. D. Cumming, Visiting Professor from University of Cali-fornia, Berkeley, for careful reading of the manuscript andhelpful comments.

REFERENCES[11 K. Phillips, "Current source converter for AC motor drives,"

IEEE Trans. Ind. Appi., vol. IA-8, Nov./Dec. 1972.

53


Fig. 16. Oscillograms of line-to-line voltage (top) and phase current(bottom) for new twelve-step CSI. Vertical: 50 V/div (top), 5 A/div(bottom). Horizontal: 10 ,s/div.

Fig. 17. Oscillogram of waveform resulting from periodical switchingbetween twelve-step and six-step. Vertical: 50 V/div (top), 5 A/div(bottom). Horizontal: 10 jLs/div.

[2] W. Farrer and J. D. Miskin, "Quasi-sine wave fully regenerativeinverter," Proc. IEEE, vol. 120, Sept. 1973.

[3] R. H. Nelson and T. A. Radomski, "Design methods for currentsource inverter/induction motor drive systems," IEEE Trans.Ind. Electron. Contr. Instrum., vol. IECI-22, no. 2, May 1975.

[4] R. L. Steigerwald, "Characteristics of a current-fed inverter withcommutation applied through load neutral point," IEEE Trans.Ind. Appl., vol. IA-15, Sept./Oct. 1979.

[5] S. Martinez and F. Aldana, "Current-source double DC-side forcedcommutated inverter," IEEE Trans. Ind. App!., vol. IA-14, Nov./Dec. 1978.

[6] S. B. Dewan and D. L. Duff, "Optimum design of an input-commutated inverter for AC motor control," IEEE Trans. Ind.Gen. Appl., vol. IGA-5, Nov./Dec. 1969.

[7] R. Palanippan and J. Vithayathil, "The current-fed twelve-stepcurrent-source inverter," IEEE Trans. Ind. Electr. Contr. In-strum., vol. IECI-25, Nov. 1978.

[8] R. Palanippan, J. Vithayathil, and S. K. Data, "Principle of a dualcurrent-source converter for AC motor drives,' IEEE Trans. Ind.Appl., vol. IA-15, July/Aug. 1979.

WXXfj.} Gyu H. Cho (S'76-S'78-M'80) was born inKorea on April 19, 1953. He received the M.S.

Xl and Ph.D. degrees in electrical engineering fromthe Korea Advanced Institute of Science andTechnology, Seoul, in 1977 and 1981, respec-

Since 1977, he has been working as a Researchand Teaching Assistant in the Department ofElectrical Engineering of the Korea Advanced

X d l l d | Institute of Science and Technology. His currentresearch area of interest is solid-state power

electronics, including switching power converters, inverters, anduninterruptible power supplies. In particular, he has been engaged in thedevelopment of variable frequency ac motor drives.

Dr. Cho holds two patents and has four patents pending.

Song B. Park (S'66-M'68) was born ing mChongjin, Korea, on May 18, 1924. He received

the M.S. and Ph.D. degrees from the UniversityoMinnesota, Minneapolis, MN, in 1962 and

1968, respectively.From 1965 to 1968 he was engaged in research

and teaching in circuits and systems at OregonState University, Corvallis, OR, and then joinedthe Korea Advanced Institute of Science andTechnology, Seoul, where he is currently a VicePresident and Professor at the Department of

Electrical Engineering. His research areas of interest include computer-aided circuit design, power electronics, and microprocessor applications.

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Six-Step Twelve-Step Current-Source Inverters DC Side ...koasas.kaist.ac.kr/bitstream/10203/6838/1/[1981]Novel Six-Step and... · CHOANDPARK:CURRENT-SOURCEINVERTERS Line F i e i Fig.

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