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WP411 (v1.0) January 30, 2012 www.xilinx.com 1
© Copyright 2012 Xilinx, Inc. Xilinx, the Xilinx logo, Artix,
ISE, Kintex, Spartan, Virtex, Zynq, and other designated brands
included herein are trademarks of Xilinx in the United States and
other countries.
The purpose of a Power Distribution Network (PDN) is toprovide
power to electrical devices in a system. Each devicein a system not
only has its own power requirements for itsinternal operation, but
also a requirement for the inputvoltage fluctuation of that power
rail. For Xilinx Kintex™-7and Virtex®-7 FPGAs, the analog power
rails have an inputvoltage fluctuation requirement of not more than
10 mVpeak-to-peak from the 10 kHz to the 80 MHz frequencyrange. The
self- generated voltage fluctuation on the powerrails is a function
of frequency and can be described byOhm's Law: Voltage (frequency)
= Current (frequency) *Self-Impedance (frequency).
Thus, if the user determines the self-impedance (frequency)and
knows the current (frequency) of the PDN, then thevoltage
(frequency) can be determined. The self-impedance(frequency) can
easily be determined by simulating thefrequency domain
self-impedance profile of the PDN andis, thus, the subject of this
white paper.
White Paper: Kintex-7 and Virtex-7 FPGAs
WP411 (v1.0) January 30, 2012
Simulating FPGA Power Integrity Using S-Parameter Models
By: Hany Fahmy and Colin Warwick of Agilent Technologies,
Inc.and Jack Carrel, Ray Anderson, Harry Fu, and Romi Mayder of
Xilinx, Inc.
http://www.xilinx.com
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2 www.xilinx.com WP411 (v1.0) January 30, 2012
Overview
OverviewBefore simulating the frequency domain self-impedance
profiles of a PDN, it is important to establish expectations for
the simulation results. To do this, an understanding of the
fundamental concepts must be attained:• Series-Resonance Circuit
and Impedance Minimums • Parallel-Resonance Circuit and Impedance
Maximums • Frequency Components of Electrical Signals • S-Parameter
Model vs. Lumped RLC Model for Decoupling Capacitors
Series-Resonance Circuit and Impedance MinimumsA series-resonant
circuit is defined by a capacitor (C) and inductor (L) that are
connected in series. When the XC (capacitive reactance) and XL
(inductive reactance) are equal in magnitude and opposite in phase,
the current is at maximum. This condition gives rise to an
impedance minimum. The frequency at which this equality occurs is
called the series-resonant frequency and is described by Equation
1:
Equation 1
A common series-resonant circuit is formed by the capacitance
(C) and the parasitic inductance (L) of a given capacitor mounted
on a printed circuit board. Figure 1 shows the schematic circuit
representation while Figure 2 shows the frequency domain impedance
profile.X-Ref Target - Figure 1
Figure 1: Series-Resonant Components of a PCB-Mounted
Capacitor
X-Ref Target - Figure 2
Figure 2: Frequency-Domain Impedance Profile of PCB-Mounted
Capacitor
f 12π LC------------------=
Series-Resonant Circuit
P3Num = 3
P4Num = 4
CC2
C=C_PCB_Capacitance
RR3
R = R_PCB_Capacitance
LL2
L=L_PCB_Cap_InductanceWP411_01_012412
0.0
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
0.1 0.2 0.3 0.4 0.5
Freq, GHz
ImpedanceMinimum
Series Resonant Circuit
Ser
ies_
Res
onan
ce
0.6
WP411_02_111011
0.7 0.8 0.9 1.0
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-
Overview
WP411 (v1.0) January 30, 2012 www.xilinx.com 3
Parallel-Resonance Circuit and Impedance MaximumsA parallel
anti-resonant circuit is defined by a capacitor (C) and inductor
(L) that are connected in parallel. When the XC (capacitive
reactance) and XL (inductive reactance) are equal in magnitude and
opposite in phase, the reactive branch currents are also equal in
magnitude and opposite in phase. This gives rise to a minimum total
current and thus, a maximum total impedance is created. The
frequency at which this condition occurs is called the parallel
anti-resonant frequency and is described by Equation 2:
Equation 2
A common parallel anti-resonant circuit is one formed by the die
capacitance and package inductance. Figure 3 shows a schematic
circuit representation while Figure 4 shows the frequency domain
impedance profile.X-Ref Target - Figure 3
Figure 3: Parallel Anti-Resonant Components of Die and Package
Properties
X-Ref Target - Figure 4
Figure 4: Frequency-Domain Impedance Profile of Die and Package
Properties
f 12π LC------------------=
Parallel Anti-Resonant Circuit
P1Num = 1
P2Num = 2
WP411_03_112411
RR1R = R_Die_Capacitance
RR2R = R_Package_Inductance
LL1L = L_Package_Inductance
CC1C = C_Die_Capacitance
0.0
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.1 0.2 0.3 0.4 0.5
Freq, GHz
ImpedanceMaximum
Parallel Anti-resonant Circuit
Par
alle
l_A
nti_
Res
onan
ce
0.6
WP411_04_111011
0.7 0.8 0.9 1.0
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4 www.xilinx.com WP411 (v1.0) January 30, 2012
Overview
Frequency Components of Electrical SignalsThe frequency domain
current profile of VCCO(f) is shown in Figure 5 and Figure 6 as
simulated at the BGA power balls of the Xilinx Virtex-7 XC7VX485T
FPGA in the FFG1761 pin package.
In the example, the simulation is running a memory interface at
1.866 Gb/s with a PRBS15 data pattern. The power spectral density
of VCCO(t) is wide-banded, extending from 10 MHz up to the 10 GHz.
As the data traffic pattern and activity change, the simulations
demonstrate that the dominant frequency components of the power
spectral density also change. Therefore, the simulations show that
PDN noise is a wide-band phenomenon; PDN simulations must,
therefore, be run over a wide-band frequency range.
X-Ref Target - Figure 5
Figure 5: Memory Interface Simulation Activity Patterns
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
Freq, Hz
Frequency Spectrum SSTL15 CurrentSSTL15 HPIO VCCO Current
mag
(FS
_Kir
tex_
7_S
ST
L15_
Cur
rent
), m
A
WP411_05_011112
IE9IE8IE7 IE10
0
5
10
15
20
25
30
Freq, Hz
Frequency Spectrum DDR3 Die Input
mag
(FS
_Kin
tex_
7_S
ST
L15)
, mV
-0.5
0.0
0.5
1.0
1.5
2.0
DD
R3_
Die
_Inp
ut_E
ye_D
iagr
am, V
olts
IE9IE8IE7 IE10
-10
-5
0
5
10
15
20
25
30
Time, nsec
Kin
tex_
7_S
ST
L15_
Cur
rent
, mA
70 80 9040 50 600 01 20 30 100
DDR3 Die Input 1,866 Mb/s
Time, nsec
0.7 0.8 0.90.4 0.5 0.60.0 0.1 0.2 0.3 1.0 1.1
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Overview
WP411 (v1.0) January 30, 2012 www.xilinx.com 5
Because the power spectral density is of a wide band, the
frequency domain self-impedance profile must be simulated over a
wide range. Below 1 kHz, the voltage regulator module (VRM)
dominates the frequency domain self-impedance profile. Above 10
GHz, the on-die capacitance dominates the impedance profile. Thus,
Xilinx recommends running the simulations from 1 kHz to 10 GHz.
S-Parameter Model vs. Lumped RLC Model for Decoupling
CapacitorsAs a comparison between using lumped RLC circuits and
S-parameters to run PDN simulations, the decoupling capacitors
portion of the PDN circuit is examined first.
In this simulation, an attempt is made to curve-fit an
S-parameter model for common X5R capacitors in the following EIA
case sizes: 0201, 0402, 0603, 0805, 1206, and 1610. After matching
the capacitive reactance and the series-resonant frequency given in
Equation 1,the percentage error of the inductive reactance at 100
MHz is measured.
These simulations are done at room temperature (25°C) with no
applied DC bias. Figure 7 through Figure 9 show the circuit
schematic representations. Figure 10 and Figure 11 show the
simulation results.
X-Ref Target - Figure 6
Figure 6: Simulation Test Setup
T T
L
PCPC
GCGC
PU
PD PD
PUVIN
I/OO
DigO
PRBS15K7325TFF900SSTL15 HP IO
DDR31.86 Gb/s
DQ Pin
40ΩImpedance3 Inch Trace
DDR3PackageParasitics
WP411_06_011112
VtPRBSVPRBS1
PRBS
IBIS_OIBIS1
IBIS_IOIBIS4
SLINTL1Subst = ”SSub1”W = 5.0 milL = 3000.0 mil
LL1L = 1.38 nHR = 0.25656
CC1C = 0.33 pF
DDR3 Die Input
Eye_ProbeEye_Probe1
V_DCSRC1
VDC = 1.5V
V_DCSRC2VDC = 1.5V
Power
Output+ +
+
+
–
–
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6 www.xilinx.com WP411 (v1.0) January 30, 2012
Overview
X-Ref Target - Figure 7
Figure 7: Decoupling Capacitors Simulation, Schematic
Representation 1
X-Ref Target - Figure 8
Figure 8: Decoupling Capacitors Simulation, Schematic
Representation 2
X-Ref Target - Figure 9
Figure 9: Decoupling Capacitors Simulation, Schematic
Representation 3
Data SheetCeramic0201 1uF4V X5R
WP411_07_013012
TermTerm1Num = 1Z = 50Ω
+
–
TermTerm2Num = 2Z = 50Ω
S2PSNP1
LL2L= 320 pHR= 10 mΩ
CC2C = 0.69 uF
+
–
TermTerm3Num = 3Z = 50Ω
+
–
TermTerm4Num = 4Z = 50Ω
+
–
Data SheetCeramic0402 4.7uF4V X5R
TermTerm16Num =16Z = 50Ω
+
–
TermTerm15Num = 15Z = 50Ω
S2PSNP4
LL3L = 500 pHR = 8 mΩ
CC3C = 3.7 uF
+
–
TermTerm14Num = 14Z = 50Ω
+
–
TermTerm13Num = 13Z = 50Ω
+
–
Data SheetCeramic0603 22uF4V X5R
WP411_08_013012
TermTerm12Num = 12Z = 50Ω
+
–
TermTerm11Num = 11Z = 50Ω
S2PSNP3
LL4L = 800 pHR = 4 mΩ
CC4C = 14 uF
+
–
TermTerm10Num = 10Z = 50Ω
+
–
TermTerm9Num = 9Z = 50Ω
+
–
Data SheetCeramic0805 22uF4V X5R
TermTerm17Num = 17Z = 50Ω
+
–
TermTerm18Num = 18Z = 50Ω
S2PSNP5
LL6L = 800 pHR = 3.5 mΩ
CC6C = 14 uF
+
–
TermTerm19Num = 19Z = 50Ω
+
–
TermTerm20Num = 20Z = 50Ω
+
–
Data SheetCeramic1206 100uF4V X5R
WP411_09_013012
TermTerm21Num = 21Z = 50Ω
+
–
TermTerm22Num = 22Z = 50Ω
S2PSNP6
LL5L = 1,360 pHR = 2.75 mΩ
CC5C = 50 uF
+
–
TermTerm23Num = 23Z = 50Ω
+
–
TermTerm24Num = 24Z = 50Ω
+
–
Data SheetCeramic1210 100uF4V X5R
TermTerm32Num = 25Z = 50Ω
+
–
TermTerm31Num = 26Z = 50Ω
S2PSNP8
LL7L = 1,200 pHR = 1.9 mΩ
CC7C = 60 uF
+
–
TermTerm30Num = 27Z = 50Ω
+
–
TermTerm29Num = 28Z = 50Ω
+
–
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Overview
WP411 (v1.0) January 30, 2012 www.xilinx.com 7
X-Ref Target - Figure 10
Figure 10: Simulation Results (EIA Case Sizes 0201 / 0603 /
1206)
X-Ref Target - Figure 11
Figure 11: Simulation Results (EIA Case Sizes 0402 / 0805 /
1210)
1E-21E5 1E6 1E7 1E8 1E9 6E9
1E-1
1
1E1
Freq, Hz
Impedance Magnitude 0201m
ag(C
eram
ic_0
201_
1uF
_RLC
)m
ag(C
eram
ic_0
201_
1uF
_S)
1E5 1E6 1E7 1E8 1E9
958065503520
0-10-25-40-55-70-85
-100
Freq, Hz
Impedance Phase 0201
phas
e(C
eram
ic_0
201_
1uF
_RLC
), d
eg
phas
e(C
eram
ic_0
201_
1uF
_S),
deg
1E-2
1E-31E5 1E6 1E7 1E8 1E9 6E9
1E-1
1
1E1
Freq, Hz
Impedance Magnitude 0603
mag
(Cer
amic
_060
3_1u
F_R
LC)
mag
(Cer
amic
_060
3_1u
F_S
)
1E5 1E6 1E7 1E8 1E9
958065503520
0-10-25-40-55-70-85
-100
Freq, Hz
Impedance Phase 0603
phas
e(C
eram
ic_0
603_
1uF
_RLC
), d
eg
phas
e(C
eram
ic_0
603_
1uF
_S),
deg
1E-2
1E-31E5 1E6 1E7 1E8 1E9 6E9
1E-1
1
1E1
Freq, Hz
Impedance Magnitude 1206
mag
(Cer
amic
_120
6_1u
F_R
LC)
mag
(Cer
amic
_120
6_1u
F_S
)
1E5 1E6 1E7 1E8 1E9
958065503520
0-10-25-40-55-70-85
-100
Freq, Hz
Impedance Phase 1206
phas
e(C
eram
ic_1
206_
1uF
_RLC
), d
eg
phas
e(C
eram
ic_1
206_
1uF
_S),
deg
WP411_10_011612
1E-3
1E-2
1E5 1E6 1E7 1E8 1E9 6E9
1E-1
1
1E1
Freq, Hz
Impedance Magnitude 0402
mag
(Cer
amic
_040
2_4_
7uF
_RLC
)m
ag(C
eram
ic_0
402_
4_7u
F_R
LC)
1E5 1E6 1E7 1E8 1E9
958065503520
0-10-25-40-55-70-85
-100
Freq, Hz
Impedance Phase 0402
phas
e(C
eram
ic_0
402_
4_7u
F_R
LC),
deg
ph
ase(
Cer
amic
_040
2_4_
7uF
_S),
deg
1E-2
1E-31E5 1E6 1E7 1E8 1E9 6E9
1E-1
1
1E1
Freq, Hz
Impedance Magnitude 0805
mag
(Cer
amic
_080
5_22
uF_R
LC)
mag
(Cer
amic
_080
5_22
uF_S
)
1E5 1E6 1E7 1E8 1E9
958065503520
0-10-25-40-55-70-85
-100
Freq, Hz
Impedance Phase 0805
phas
e(C
eram
ic_0
805_
22uF
_RLC
), d
eg
phas
e(C
eram
ic_0
805_
22uF
_S),
deg
1E-2
1E-31E5 1E6 1E7 1E8 1E9 6E9
1E-1
1
1E1
Freq, Hz
Impedance Magnitude 1210
mag
(Cer
amic
_121
0_10
0uF
_RLC
)m
ag(C
eram
ic_1
210_
100u
F_S
)
1E5 1E6 1E7 1E8 1E9
958065503520
0-10-25-40-55-70-85
-100
Freq, Hz
Impedance Phase 1210
phas
e(C
eram
ic_1
210_
100u
F_R
LC),
deg
ph
ase(
Cer
amic
_121
0_10
0uF
_S),
deg
WP411_11_011612
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8 www.xilinx.com WP411 (v1.0) January 30, 2012
Overview
A summary of the data is shown below in Table 1:
It is known that the typical capacitor manufacturer specifies
the capacitance of a capacitor with zero DC bias and 0.5 Vrms AC
voltage, while the s-parameter models are typically measured with a
0 dbm AC signal.
In S-Parameter Models for Decoupling Capacitors section, the
various methods for generating the S-parameter model of a capacitor
are examined.
S-Parameter Models for Decoupling CapacitorsAt first glance, the
measurement of the capacitor's PDN impedance profile (the impedance
with respect to frequency) seems to be a simple task, but several
subtle details are required to ensure the measured data is
accurate.
The frequency domain measurement is usually accomplished by
utilizing a Vector Network Analyzer (VNA). The obvious method is to
probe the PDN making an S11 measurement, and then convert the
measured s-parameters to impedance by means of the Equation 3
relationship:
Equation 3
An impedance measurement using this method, however, has
inherent inaccuracies due to the fact that the instrument typically
has a 50Ω input impedance and the PDN has a very low impedance
(typically in the milliohm range). The accuracy of the measured VNA
data inherently has errors because the typical uncertainty of S11
(when rho, the reflection coefficient, is near 1) can be in the
1%–2% range.
This equates to an impedance uncertainty in the 0.3Ω-to-0.4Ω
range. If PDN impedances in the milliohm range are being measured,
it quickly becomes obvious that the desired impedance measurement
is lost in the measurement uncertainty.
A second factor to consider is that the inductive parasitics of
the probing arrangement can easily exceed the value of the DUT
inductance. There is no easy way to back out the probe parasitics
from the measured data.
Fortunately, an S21 measurement is a good alternative to an S11
measurement to determine the PDN impedance. In this method, it is
found that Zdut = 25(S21). With this measurement technique, the
solder of the decoupling capacitor is included in the measurement.
By utilizing the S21 measurement, the impedance uncertainty is
reduced into the 10s-of-milliohms range. In addition, the probe
parasitics are in series with 50Ω as opposed to being in series
with the DUT impedance, which reduces their effects to near
negligible levels. For a more complete discussion of this topic,
see Accuracy Improvements of PDN Impedance Measurements in the Low
to Middle Frequency
Table 1: Summary of Result Data, Decoupling Capacitors
Simulation
Size Capacitance (µF) Impedance Magnitude @ 100
MHzSeries-Resonant
FrequencyEIA Code S-Parameter
Data Sheet % Error S-Parameter
RLC Model % Error
1210 60 100 66.7 0.209 0.751 259.3 600 KHz
1206 50 100 100.0 0.255 0.845 231.4 700 KHz
805 14 22 57.1 0.18 0.501 178.3 1.5 MHz
603 14 22 57.1 0.178 0.501 181.5 1.5 MHz
402 3.7 4.7 27.0 0.15 0.313 108.7 3.5 MHz
201 0.69 1 44.9 0.129 0.198 53.5 10 MHz
Zdut 501 S11+1 S11–------------------=
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-
Overview
WP411 (v1.0) January 30, 2012 www.xilinx.com 9
Range presented at DesignCon 2010 by Istvan Novak of SUN
Microsystems and Yasuhiro Mori and Mike Resso of Agilent
Technologies
(http://www.home.agilent.com/upload/cmc_upload/All/DC10_ID2696_Novak-Mori-Resso.pdf).
RLC Models for Decoupling CapacitorDecoupling capacitors are
often characterized by vendors by means of three parameters: R
(resistance), L (inductance), and C (capacitance). The C parameter
is the decap's intrinsic capacitance; the L is its intrinsic
inductance; and the R is the ESR of the decoupling capacitor. When
this simple RLC model for a decoupling capacitor is utilized in a
simulation along with a good PDN model, the mounting inductance and
spreading inductance associated with the package or PCB combines
with the decap's intrinsic inductance to effectively model the loop
inductance. This loop inductance plus the package inductance
resonates with the die capacitance to form a parallel anti-resonant
circuit with a unique impedance profile.
Series RLC models of decoupling capacitors are easy to
understand, and they simulate quickly as both frequency domain and
transient simulations with a minimum of issues. As noted
previously, the RLC values for the model can come from a vendor's
data sheet; alternatively, they can be derived from measured
s-parameter data by fitting the values of a simple series RLC
circuit to the response of the s-parameters. In some cases,
particularly at low frequencies, the simple series RLC circuit
works adequately. However, when it is required to determine the
impedance profile of a PDN accurately over a wide bandwidth of DC
to several gigahertz, things usually do not work out so simply.
Two main issues make simple series RLC models inadequate for
accurate PDN simulations. Due to the stacked layers of the
decoupling capacitor construction, there is distributed inductance
and resistance in the Z axis of the plate stack. This causes the L
parameter of the series RLC representation to be frequency
dependent. In most simulators, there is no frequency-dependent L
element. First, a reasonably accurate series RLC model can be
constructed at either low frequencies or high frequencies, but
cannot model both simultaneously. Second, while a complex
multi-element model can be constructed to more accurately model the
frequency-dependent L effect, such models are very difficult to
design and manage.
Therefore, rather than use a simple series RLC circuit that is
known to be inaccurate over a wide bandwidth, or attempt to
synthesize a more complex multi-element model, the simulation work
done at Xilinx suggests that it is much easier and more accurate to
utilize a measured wideband s-parameter decoupling capacitor model
when simulating PDNs.
Note: Ceramic decoupling capacitor models are strongly voltage
dependent. Therefore, it is important to obtain the s-parameter
model from the capacitor manufacturer that has been measured at the
operating voltage of interest—for both DC and AC voltages.
http://www.xilinx.comhttp://www.home.agilent.com/upload/cmc_upload/All/DC10_ID2696_Novak-Mori-Resso.pdfhttp://www.home.agilent.com/upload/cmc_upload/All/DC10_ID2696_Novak-Mori-Resso.pdf
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10 www.xilinx.com WP411 (v1.0) January 30, 2012
Running the PDN Simulations with the Agilent ADS 2011.10
Running the PDN Simulations with the Agilent ADS 2011.10To
simulate the frequency domain self-impedance profile of a Power
Distribution Network, Xilinx recommends using the Agilent ADS 2011
software bundle. This software bundle provides the
high-speed-digital (HSD) designer with a wide range of tools. Every
aspect of the power integrity problem requires a specific technique
for solving it. For example, PDN analysis requires the
following:
1. True frequency-domain simulation of the PDN parallel
anti-resonances and series resonances with solid S-parameter
handling and assurance of “Passivity and Causality”
2. Patented convolution (Kramers-Kronig) to bring
frequency-domain models (measurement-based models and EM-based
models) into the time domain (eye diagrams, BER contours, and
jitter decomposition)
3. Using an extraction technique, such as Method-of-Moment,
which has excellent accuracy from DC to GHz range
PDN Simulation Example In this simulation example, the
simulation performed is the PDN of the MGTAVCC and MGTAVTT analog
power rails for the Xilinx 7 series XC7VX485T FPGA in the FFG1761
pin package.
Two cases are simulated here. Case 1 uses the PCB capacitors
listed in Table 2, which are similar to the recommended PCB caps
for the Xilinx Virtex-6 devices.
Case 2 uses the PCB capacitors described in Table 3.
Figure 12 is the schematic for both cases (1) and (2) listed
above for the MGTAVCC and MGTAVTT power rails. For case 2 (with no
PCB capacitors), there is still one bulk capacitor mounted on the
PCB specified by the manufacturer of the voltage regulator
module.
Table 2: Case 1 Capacitors
QTY per Group Capacitance (µF)MGTAVCC MGTAVTT MGTVCCAUX
4 4 2 0.022
4 4 0 0.47
2 2 1 1
2 2 1 4.7
Table 3: Case 2 Capacitors
QTY per Group Capacitance (µF)MGTAVCC MGTAVTT MGTVCCAUX
0 0 0 0.022
0 0 0 0.47
0 0 0 1
0 0 0 4.7
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-
Running the PDN Simulations with the Agilent ADS 2011.10
WP411 (v1.0) January 30, 2012 www.xilinx.com 11
X-Ref Target - Figure 12
Figure 12: Power Rails Simulation Schematic Representation
TermTerm13Num=13Z=50 Ohm
TermTerm14Num=14Z=50 Ohm
CC30C=22.16 nF
S2PSNP105File=“GRM155R61C223KA01_022uF_0402 S2P”
S2PSNP49File=“T520V337M2R5ATE025.s2p”
S2PSNP111
File=“GRM152R60J474ME15_047_0402.S2P”
S2PSNP108
File=“GRM152R60J474ME15_047_0402.S2P”
S2PSNP107
File=“GRM152R60J474ME15_047_0402.S2P”
S2PSNP106
File=“GRM152R60J474ME15_047_0402.S2P”
S2PSNP109
File=“GRM188R61C105KA93_1uF_0603.S2P”
S2PSNP53File=“GRM033C80G104KE19series(for_Fuji2_AVTT_AVCC).s2p”
S2PSNP52File=“GRM033C80G104KE19series(for_Fuji2_AVTT_AVCC).s2p”
S2PSNP51File=“GRM033C80G104KE19series(for_Fuji2_AVTT_AVCC).s2p”
S2PSNP50File=“GRM033C80G104KE19series(for_Fuji2_AVTT_AVCC).s2p”
LL28L=25 nHR=1 mOhm
V_DCSRC7Vdc=1.2 V
+
–
S6PSNP48File=“fga2034_485t_ff1761_031411_Avcc_G10.s6p”
S2PSNP104File=“GRM155R61C223KA01_022uF_0402.S2P”
S2PSNP105File=“GRM155R61C223KA01_022uF_0402.S2P”
S2PSNP105File=“GRM155R61C223KA01_022uF_0402.S2P”
RR8R=10 mOhm
WP411_12_012212
++
––
PackageCapacitors
VRM
Die Capacitors
S2PSNP112File=“GRM188R60J475KE19_47uF_0603.S2P”
S2PSNP113File=“GRM188R60J475KE19_47uF_0603.S2P”
S2PSNP110File=“GRM188R61C105KA93_1uF_0603.S2P”
S20PSNP47
File=“VC7203_MGTAVCC_092611_175001_S.s20p”
http://www.xilinx.com
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12 www.xilinx.com WP411 (v1.0) January 30, 2012
Running the PDN Simulations with the Agilent ADS 2011.10
Figure 13 show the simulations results.
Figure 14 shows the complete simulation time using a typical
laptop computer running the Windows-7 64-bit operating system is
only 11.68 seconds!
Because the simulation results for both cases result in almost
identical frequency domain self-impedance profiles for the MGTAVCC
and MGTAVTT power rails, and because the MGTVCCAUX power rail has
an internal low drop out regulator integrated on the die, similar
performance between the two cases should be expected. As a simple
reference, the impedance profiles were simulated on a competitive
device with 0 PCB capacitors beyond the 1 bulk PCB capacitor
typically required by the voltage regulator manufacturer. Profiles
representing the VCCH_GXBL0, VCCT_GXBL0, and VCCR_GXBL0 power rails
were run.
X-Ref Target - Figure 13
Figure 13: Power Rails Simulation Results
1E3 1E4 1E5 1E6 1E7 1E8 1E9 1E10
freq, Hz
MGTAVCC - Magnitude
Typical Competitor
mag
(Com
petit
or_P
LL_S
uppl
y_W
ith_P
CB
_Cap
s)m
ag(M
GTA
VC
C_D
ie_W
ithou
t_P
CB
_Cap
s)m
ag(M
GTA
VC
C_D
ie_W
ith_P
CB
_Cap
s)
1300
1200
1100
1000
900
800
700
600
500
400
300
200
100
01E3 1E4 1E5 1E6 1E7 1E8 1E9 1E10
freq, Hz
MGTAVCC - Magnitude
Typical Competitor
mag
(Com
petit
or_T
x_R
x_S
uppl
y_W
ith_P
CB
_Cap
s)m
ag(M
GTA
VT
T_D
ie_W
ithou
t_P
CB
_Cap
s)m
ag(M
GTA
VT
T_D
ie_W
ith_P
CB
_Cap
s)
1300
1200
1100
1000
900
800
700
600
500
400
300
200
100
0
WP411_13_012212
X-Ref Target - Figure 14
Figure 14: Complete Simulation Time, Windows-7 64-bit OS
WP411_14_012212
http://www.xilinx.com
-
Running the PDN Simulations with the Agilent ADS 2011.10
WP411 (v1.0) January 30, 2012 www.xilinx.com 13
As can easily be seen in the PDN profiles of a typical
competitive device, the analog rails would have a peak impedance of
well over 2Ω if the PCB caps were removed!
Transmitter Hardware MeasurementsFigure 15 through Figure 18
contain a series of eye diagrams at 10.3125 Gb/s using the QPLL and
6.25 Gb/s using the CPLL with PRBS15 data pattern measured on the
Agilent Infiniium DCA-J Wide-Bandwidth Oscilloscope. This Agilent
86100C with the 86108A precision waveform analyzer has been
selected to make these hardware measurements because of the
following key attributes:
1. High bandwidth, low noise, and ultra-low residual jitter
2. Simple one connection “triggerless” operation
3. PLL characterization including loop BW/jitter transfer
4. Integrated hardware clock recover with adjustable loop
BW/Peaking—exceeds industry standards
Figure 15 shows the eye diagram and associated jitter
decomposition when using the CPLL running at 6.25 Gb/s for case
1.
Figure 16 shows the eye diagram and associated jitter
decomposition when using the CPLL running at 6.25 Gb/s for case 2
(no PCB caps).
X-Ref Target - Figure 15
Figure 15: Case 1 Eye Diagram, 6.25 Gb/s
WP411_15_011612
http://www.xilinx.com
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14 www.xilinx.com WP411 (v1.0) January 30, 2012
Running the PDN Simulations with the Agilent ADS 2011.10
Figure 17 shows the eye diagram and associated jitter
decomposition when using the QPLL running at 10.3125 Gb/s for case
1.
Figure 18 shows the eye diagram and associated jitter
decomposition when using the QPLL running at 10.3125 Gb/s for case
2 (no PCB caps).
X-Ref Target - Figure 16
Figure 16: Case 2 Eye Diagram, 6.25 Gb/s
WP411_16_011612
X-Ref Target - Figure 17
Figure 17: Case 1 Eye Diagram, 10.3125 Gb/s
WP411_17_011612
http://www.xilinx.com
-
Running the PDN Simulations with the Agilent ADS 2011.10
WP411 (v1.0) January 30, 2012 www.xilinx.com 15
As seen in the scope screenshots in Figure 15 through Figure 18,
the total jitter is both cases 1 and 2 is within the measurement
tolerance of the setup. Thus, hardware measurements have confirmed
the simulation results showing that 0 PCB caps are required for
proper operation of the transmitter.
Receiver MeasurementsTable 4 is a summary of the receiver
hardware measurements based on a loopback test using eyescan. The
data recorded in Table 4 is the voltage amplitude noise with all
transceivers in the package running asynchronously. As shown by the
data, the voltage amplitude noise is the same or less after all the
PCB caps have been removed when using either the CPLL or the
QPLL.
Figure 19 is a summary of the receiver's jitter tolerance
analysis with all transceivers in the package running
asynchronously for both cases 1 and 2.
As shown by the data, the jitter tolerance is the same or less
after all the PCB caps have been removed. The jitter tolerance
analysis was done at 10-12 BER threshold and a data rate of 10.3125
Gb/s.
X-Ref Target - Figure 18
Figure 18: Case 2 Eye Diagram, 10.3125 Gb/s
WP411_18_011612
Table 4: Comparison of Voltage Amplitude Noise with/without
Decoupling Caps
PLL CPLL QPLL
Bit Rate 6.25 Gb/s 10.3125 Gb/s
MGTAVCC All Caps No Caps All Caps No Caps
MGTAVTT All Caps No Caps All Caps No Caps
MGTVCCAUX All Caps No Caps All Caps No Caps
% Full Scale 3.6% 3.3% 5.0% 4.5%
http://www.xilinx.com
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16 www.xilinx.com WP411 (v1.0) January 30, 2012
Summary
SummaryPDN simulations, confirmed by hardware measurements, have
shown that no PCB caps beyond that recommended by the voltage
regulator manufacturer are required for the MGTAVTT, MGTAVCC, and
MGTVCCAUX power rails for proper operation of the transceivers in
Xilinx's Kintex-7 and Virtex-7 devices.
While the PCB capacitors are not needed for proper operation of
the transceivers, however, proper filtering can be required on the
PCB to achieve the input voltage ripple noise specification of 10
mV peak-to-peak (10 kHz to 80 MHz) when measured at the BGA ball of
the package.
Currently, Xilinx has several Agilent ADS Power Integrity Design
Kits available for 7 series FPGAs that support all device power
supplies (digital and analog). Contact your local Xilinx field
application engineer to obtain these Agilent ADS Design Kits.
To obtain a 30-day free license of Agilent ADS2011, please visit
the following link:
https://software.business.agilent.com/TrialLicense/TrialLicenseRequest.aspx?ProdNum=W2200F-1U1-TRL
X-Ref Target - Figure 19
Figure 19: Comparison of Jitter Tolerance with/without
Decoupling Caps
100.00
10.00
1.00
0.100
All Caps BER12
No Caps BER12
0.01 0.10 1.00
Frequency, MHz
SI,
UI
10.00 100.00
WP411_xx_012212
http://www.xilinx.comhttps://software.business.agilent.com/TrialLicense/TrialLicenseRequest.aspx?ProdNum=W2200F-1U1-TRLhttps://software.business.agilent.com/TrialLicense/TrialLicenseRequest.aspx?ProdNum=W2200F-1U1-TRL
-
Revision History
WP411 (v1.0) January 30, 2012 www.xilinx.com 17
Revision HistoryThe following table shows the revision history
for this document:
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(the “Materials”) is provided solely for the selection and useof
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http://www.xilinx.com/warranty.htm#critapps.
Date Version Description of Revisions
01/30/12 1.0 Initial Xilinx release.
http://www.xilinx.comhttp://www.xilinx.com/warranty.htmhttp://www.xilinx.com/warranty.htm#critapps
Simulating FPGA Power Integrity Using S-Parameter
ModelsOverviewSeries-Resonance Circuit and Impedance
MinimumsParallel-Resonance Circuit and Impedance MaximumsFrequency
Components of Electrical SignalsS-Parameter Model vs. Lumped RLC
Model for Decoupling Capacitors
Running the PDN Simulations with the Agilent ADS 2011.10PDN
Simulation ExampleTransmitter Hardware MeasurementsReceiver
Measurements
SummaryRevision HistoryNotice of Disclaimer