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Naval Command,Control and Ocean San Diego. CASurveillance Center
RDT&E Division 92152-5001
ADaA28 2 369
Technical Report 1646
May 1994
UltrawidebandShipboard ElectroopticElectromagneticEnvironment
Monitoring
S. A. Pappert DTICM. H. Berry A ELECTES. M. Hart JUL25199 4
R. J. Orazi m G DL. B. Koyama NJS.T.Li
94-22723
A ii M l Aproved for public rele ase; distribuion is un
blled
Ivord
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Technical Report 1646May 1994
Ultrawideband Shipboard ElectroopticElectromagnetic Environment
Monitoring
Accesion For
S. A. Pappert NTIS CRA&MDTIC TAB
M. H. Berry Unannounced 0S. M . Hart Justification
.................... .
R. J. OraziL.B .K ya aBy ......................... .L. B. Koyama
Dist. ibution I
Availability Codes
Avail and orDist Special
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NAVAL COMMAND, CONTROL ANDOCEAN SURVEILLANCE CENTER
RDT&E DIVISIONSan Diego, California 92152-5001
K. E. EVANS, CAFiT USN R. T. SHEARERCommanding Officer Executive
Director
ADMINISTRATIVE INFORMATION
This report is submitted in partial fulfillment of Milestones 3,
4, 5, and 6, Task Num-ber 3 (Electrooptic Electromagnetic
Monitoring) of the Electromagnetic CompatibilityProject (RH21C13)
of the Surface Ship Technology Area Program (SC1A/PE0602121N).The
work described herein was sponsored by the Office of Naval Research
(ONR 334)under contract N0016791WX10040. The work was performed by
the Naval Command,Control and Ocean Surveillance Center (NRaD Codes
555, 822, 843, and 753).
Released by Under authority ofMatthew N. McLandrich, Head Dr.
Howard E. Rast, HeadOptical Electronics Branch Solid State
Electronics
Division
SM
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EXECUTIVE SUMMARY
OBJECTIVE
1. Design, construct, package, test, and evaluate the best
available wideband photonicelectromagnetic field probe for
shipboard electromagnetic environment (EME) moni-toring.
2. Develop engineering solutions to problems encountered in
Phase I testing of a first-generation breadboarded sensor.
Investigate and develop alternative sensor configura-tions that can
lead to improved system performance.
3. Develop requirements and a test plan for the shipboard
testing of a 2-MHz to 18-GHzprototype system.
RESULTS
1. Broadband electromagnetic (EM) field detection using
antenna-coupled fiber opticlinks has been successfully demonstrated
in the 2-MHz to 18-GHz frequency range.This remote sensing system
is potentially operable from 2 MHz to 50 GHz and can bepackaged
into small, lightweight units. Both amplitude and frequency
information ofmultiple radio frequency and microwave signals have
been simultaneously monitoredwith this wideband optical system. A
root mean square electric field sensitivity ofapproximately 1 gV/m
and a spurious free dynamic range of 109 dB/Hz213 have
beendemonstrated with the 18-GHz externally modulated system.
2. Semiconductor optical waveguide modulators have been
developed as an alternativeto lithium niobate Mach-Zehnder
modulators for this application. Systems employingthe less mature
semiconductor optical modulators have shown greater
modulationefficiency for a given bandwidth at the expense of added
optical insertion loss thanlithium niobate-based systems.
3. Remote modulator bias control techniques have been
investigated and a system basedon the active control of the
modulator direct current bias to compensate for bias posi-tion
drift has been developed. This system utilizes a remote
power-by-light approachin which a low-frequency optical control
signal is sent to the modulator via an opticalfiber and is remotely
monitored to properly bias the optical modulator. This systemhas
been successfully demonstrated in the 10'C to 90°C temperature
range.
4. A remote optical polarization controller has been
successfully demonstrated, whicheliminated the need for expensive
polarization-maintaining fiber. The controller hasan optical
insertion loss of 4.5-dB, resulting in a 9-dB penalty in detection
sensitivity.Recent improvements in liquid crystal technology have
resulted in polarization con-troller units with
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high-dielectric-constant substrate. A third approach has been to
use a sectoralizedloaded monopole antenna. A fourth approach is to
suffer the gain penalty and use anelectrically short dipole
antenna. Each technique has its advantages and disadvan-tages,
although the use of an electrically short dipole antenna is the
simplestapproach, and results indicate that acceptable 2-MHz to
500-MHz electric field sensi-tivities can be obtained using this
approach.
6. A preliminary test plan for the FY 94 shipboard demonstration
aboard an LSD-41class surface ship is outlined. Preliminary results
indicate that as few as three topsidemonitoring locations are
required to obtain the electromagnetic signature of the ship.
RECOMMENDATIONS
I. Prepare the 2-MHz to 18-GHz prototype electrooptic EME system
for a shipboarddemonstration test to occur in FY 94. Continue
developing the semiconductor opticalwaveguide modulators to
eventually replace the presently used lithium niobate modu-lators.
Extend fiber optic link operation to 50 GHz using both
semiconductor andlithium niobate modulators and compare their
performance.
2. Develop and optimize radio frequency and microwave probe
designs for continuous2-MHz to 50-GHz frequency coverage with
compact, lightweight antennas.
3. Demonstrate EM field detection from 2 MHz to 50 GHz with a
single fiber optic linkand as few antennas as possible.
iv
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CONTENTS
EXECUTIVE SUM MARY
................................................... iii
ABBREVIATIONS
......................................................... ix
1.0 INTRODUCTION
..................................................... 1
1.1 TECHNICAL OBJECTIVE AND APPROACH ........................
1
1.2 BACKGROUND .................................................
2
2.0 SHIPBOARD EO ENE MONITORING
.................................... 4
2.1 DESCRIPTION OF EXTERNALLY MODULATED EO EMEMONITORING SYSTEM
.......................................... 4
2.1.1 Lithium Niobate Optical Waveguide Modulators
................. 6
2.1.2 1H-V Semiconductor Optical Waveguide Modulators
.............. 7
2.2 ANALYSIS OF EO EME MONITORING SYSTEM ....................
10
2.3 PERFORMANCE OF FIBER OPTIC LINK ...........................
17
2.4 ELECTROMAGNETIC FIELD MEASUREMENTS IN THE2- to 18-GHz
FREQUENCY RANGE USING AN EO EMEMONITORING SYSTEM
.......................................... 20
2.4.1 System Configuration
....................................... 21
2.4.2 Results ..................................................
22
3.0 IMPROVEMENTS TO EO EME MONITORING SYSTEM
.................... 28
3.1 IMPROVEMENTS TO FIBER OPTIC LINK ..........................
28
3.1.1 Ultrawideband Optical Modulators
............................ 28
3.1.2 Optically Powered and Controlled Remote Fiber Optic Links
....... 30
3.1.3 Remote Optical Polarization Control of EM Field Sensor
........... 37
3.2 IMPROVEMENTS TO WJDBAND RF PROBE ........................
40
3.2.1 Electrically Short Dipole or Monopole Antenna
.................. 41
3.2.2 Enlarged Spiral Antenna
.................................... 42
3.2.3 Sectorialized Monopole Antenna
.............................. 42
3.2.4 High-Dielectric Spiral Antenna
............................... 42
v
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4.0 SHIPBOARD DEMONSTRATION TEST PLAN
............................ 47
4.1 SHIP CLASS SELECTION
........................................ 47
4.2 SHIPBOARD EME DETERMINATION ..............................
52
4.3 SHIPBOARD TEST AND EVALUATION PROCEDURES ...............
55
5.0 CONCLUSIONS
...................................................... 56
6.0 RECOMMENDATIONS
................................................ 58
7.0 REFERENCES
........................................................ 59
FIGURES
2-1. Schematic diagram of an antenna-coupled externally
modulated EO EMEm onitoring system
..................................................... 5
2-2. Optical powering scheme using GaAs photovoltaic cells for
the EO EMEm onitoring system
..................................................... 5
2-3. Triplexing scheme to combine RF information from 2 MHz
through 50 GHz ....... 6
2-4. Schematic of lithium niobate MZ interferometric optical
modulator .............. 7
2-5. Structure of LPE-grown 1.32-pm InGaAsP FKE optical
modulator ............... 9
2-6. Structure of MOCVD-grown 1.52-1&m InGaAs/InP multiple
quantum wellQCSE optical modulator
................................................ 9
2-7. Relative transmission versus applied bias curves for the
1.32-pam FKE waveguidemodulator, the 1.52-pm QCSE waveguide
modulator, and the 1.32-tpm lithiumniobate MZ waveguide modulator
......................................... 11
2-8. Experimental values for Aa versus applied bias for the
1.32-pm FKE and1.52-pm QCSE waveguide modulators
..................................... 12
2-9. Optical receiver output power versus modulation index,
including harmonicsand intermodulation products for the 1.32 -pm MZ
modulator ................... 13
2-10. Optical receiver output power versus modulation index,
including harmonicsand intermodulation products for the 1.32-pm FKE
modulator ................... 13
2-11. Optical receiver output power versus modulation index,
including harmonicsand intermodulation products for the 1.32-prm
QCSE modulator ................. 14
2-12. Frequency response curve for 1.32-pm LPE-grown FKE
modulator .............. 16
2-13. Plot of m2/(S/N), where m is the modulation index, in a
1-Hz bandwidthversus optical detector photocurrent for different
laser RIN values ............... 17
2-14. Schematic diagram of best available DC to 18-GHz
externally modulated fiberoptic link
.............................................................
18
vi
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2-15. Modulation frequency response of wideband lithium niobate
optical modulator ..... 18
2-16. Externally modulated fiber optic link RF insertion loss
versus frequencyin the 50-MHz to 18-GHz range
.......................................... 19
2-17. Diagram of the externally modulated fiber optic link loss
budget ................. 19
2-18. Total noise power (solid line) and experimental data
(circles) versus opticaldetector photocurrent for the externally
modulated fiber optic link ............... 20
2-19. RF insertion loss versus frequency for a DC to 500-MHz
fiber optic link usinga Va = 4 V optical modulator
............................................. 21
2-20. Schematic diagram of the antenna-coupled externally
modulated fiber opticlink for EM field detection
............................................... 22
2-21. Modulation frequency response from 2 to 18 GHz of the
externally modulatedfiber optic link and spiral antenna
......................................... 23
2-22. Plot of m2/(S/N), where m is the modulation index, due to
total link noise(solid line), along with experimental data points
(circles) ....................... 24
2-23. Plot of experimental (circles) and theoretical detected
power for fundamental(solid line) and third-order intermodulation
products (broken line) versuspower applied to the IOM
............................................... 24
2-24. EM field detection system output versus incident RMS
electric field level ......... 26
2-25. Measured RMS electric field levels versus calculated RMS
electric field levels ..... 27
3-1. Fiber optic link transfer function
.......................................... 29
3-2. Schematic of optical powering system and modulator bias
control circuitry ........ 31
3-3. Photocell output electrical voltage as a function of series
resistance for 50-mWinput optical power
..................................................... 31
3-4. Photocell output voltage versus input optical power using a
2.7-kQ seriesresistance
............................................................ 32
3-5. Fundamental and spurious signal levels as a function of
phase bias error for theminimum detectable modulation index
..................................... 33
3-6. Effect of 1-degree modulator phase bias error on the
30-500-MHz fiberoptic link
.............................................................
34
3-7. Modulator phase bias tolerance as a function of receiver
bandwidth for the500-M Hz fiber optic link
................................................ 35
3-8. Modulator phase bias drift as a function of temperature for
proton-exchangedand titanium-indiffused lithium niobate optical
modulators ..................... 35
3-9. Computer-controlled modulator phase bias stabilization
results as a function oftemperature for both low- and
high-frequency fiber optic links .................. 36
vii
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3-10. Remote polarization control setup using liquid-crystal
phase retarders ............ 38
3-11. Fiber optic link RF insertion loss with optimum
polarization alignment ........... 39
3-12. Fiber optic link RF insertion loss with worst-case
polarization alignment .......... 39
3-13. Fiber optic link RF insertion loss with optimum
liquid-crystal phase-retarderalignm ent
............................................................ 40
3-14. Gain versus frequency for the Watkins-Johnson two-arm
spiral antenna ........... 41
3-15. Antenna output power into 50-9 load versus frequency for a
bare andresistively loaded 15-cm dipole with 0.7-V/m RMS electric
field excitation ........ 43
3-16. Minimum detectable RMS electric field level versus
frequency for a baredipole, assuming a 40-dB system noise figure
and a 30-kHz receier bandwidth ...... 43
3-17. Return loss (S11) versus frequency for 3-inch, 1.5-turn
spiral antennasfabricated on relative dielectric constant 2.2
(squares) and 80 (crosses)substrate m aterial
...................................................... 45
3-18. Return loss (S11) versus frequency for 6-inch, 1.5-tumm
spiral antennasfabricated on relative dielectric constant 2.2
(squares) and 80 (crosses)substrate m aterial
...................................................... 45
3-19. Electrical return loss (S11) for 6-inch spiral antennas on
2.2 dielectricconstant substrates with 1 (crosses), 1.5 (squares),
and 2 (diamonds) turns ......... 46
3-20. Electrical return loss (S11) versus frequency for a
12-inch, 6-turn spiralantenna on 2.2 dielectric constant substrate
.................................. 46
4-1. Bow of USS Germantown (LSD-42)
....................................... 48
4-2. Stern of USS Germantown (LSD-42)
...................................... 48
4-3. Forty-five degrees from bow, port view of USS Germantown
(LSD-42) ........... 49
4-4. Forty-five degrees for stem, port view of USS Germantown
(LSD-42) ............ 49
4-5. Broadside, port view of USS Germantown (LSD-42)
.......................... 50
4-6. Broadside, starboard view of USS Germantown (LSD-42)
...................... 50
4-7. Candidate topside measurement sites aboard LSD 41 class
ship .................. 54
TABLES
2-1. Externally modulated EO EME monitoring system electric
field level detectionranges at 2.25, 9.52, and 16.0 GHz
........................................ 27
4-1. LSD-41 class antenna characteristics
....................................... 51
viii
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ABBREVIATIONS
BW bandwidthEA electroabsorptionEHF extremely high frequencyEM
electromagneticEMC electromagnetic compatibilityEMCON
electromagnetic controlENE electromagnetic environmentEMI
electromagnetic interferenceEO electroopticFKE Franz-Keldysh
effectHF high frequencyIOM integrated optical modulatorLNA
low-noise amplifierLPE liquid phase epitaxyMF multimode fiberMOCVD
metalorganic chemical vapor depositionMZ Mach-ZehnderNF noise
figurePBL power-by-lightPMF polarization-maintaining fiberQCSE
quantum-confined Stark effectQMS quality monitoring systemQW
quantum wellRF radio frequencyRIN relative intensity noiseRMS root
mean squareSATCOM satellite communicationSFDR spurious free dynamic
rangeSL superlatticeSMF single-mode fiberS/N signal-to-noise
ratioUHF ultrahigh frequencyVHF very high frequencyVSWR voltage
standing wave ratio
ix
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1.0 INTRODUCTION
The shipboard electromagnetic environment (EME) routinely
consists of high-level on-shipemissions extending from the
high-frequency (HF) band into the extremely high frequency(EHF)
band. Rapid advances in the areas of radar, electronic warfare
(EW), and communica-tions technologies are making this EME more
complex and at the same time more difficult tomanage. The Navy has
an increasing need to monitor these emissions and verify
electromag-netic control (EMCON) status as well as alert personnel
of hazardous emission levels. Toaddress this need, the Navy is
seeking affordable, broadband shipboard EME monitoring probesand
systems that are as nonperturbing as possible.
In response to this shipboard EME monitoring requirement,
photonic techniques are beingdeveloped in association with wideband
radio frequency (RF) probes for use in EME monitoringsystems. The
emerging photonic technologies will enable the Navy to collect and
transmitbroadband shipboard EME information for processing at a
remote site while minimizing theoverall system intrusiveness.
Recent advances of high-speed fiber optic and electrooptic
(EO)components has made this EME monitoring system approach viable
and will provide shipoperators with a valuable new command,
control, communication, computer, and intelligence(C41) capability.
This status report reviews past EO EME monitoring system work and
presentsresults from current efforts geared toward making this
technology feasible. Requirements aswell as test and evaluation
plans for the shipboard demonstration to be conducted under
thisprogram in FY 94 are also presented.
1.1 TECHNICAL OBJECTIVE AND APPROACH
The principal objective of the Office of Naval
Research-sponsored Electromagnetic Compat-ibility (EMC) Project is
to minimize exterior electromagnetic interference (EMI)
problemsduring the entire life cycle of Navy ships. An additional
objective is the reduction or control ofthe RF emission signature
of Navy surface combatants. This involves the task of
controllingshipboard RF emissions by remote monitoring of the
entire shipboard RF spectrum to determine,verify, and enforce the
EMCON status.
Task Number 3 of the EMC Project develops a broadband,
large-dynamic-range EM fieldprobe that can be packaged into a
small, lightweight system. The Task couples EO devices towideband
RF probes to provide a low-cost system that can monitor EM emission
from the ship.This system is potentially useful from 2 MHz to 50
GHz, where most shipboard emission occurs.In addition to EMCON
monitoring, the potential benefits or usages of this system will
includethe detection of RF hazards levels at key locations on the
ship weather decks, the qualitymonitoring of communication emitters
(QMS), and real-time automated frequency management.
The technical approach of this task is to build upon current
photonic technology and bridgethe gap where technology deficiencies
exist to demonstrate a 2-MHz to 50-GHz EO EM fieldsensor. Photonic
and wideband antenna component performances are being extended
wherenecessary to meet the shipboard application requirements. The
task will culminate in a ship-board demonstration of a prototype EO
field sensor at which time the effort will be transitionedfor
further engineering development.
This four-year R&D task (FY 91 to FY 94) is being performed
in three phases. Phase I wasstarted and completed in FY 91; Phase
II was started in FY 92 and will be completed in FY 94;
1
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and Phase M11 is to begin in FY 93 and conclude in FY 94. Phase
I (Milestones 1 and 2) con-sisted of assessing the present
state-of-the-art of EO field sensors and determining their
suitabil-ity for shipboard use. Phase II (Milestones 3, 4, 5, and
6) of this effort deals with solvingproblems determined from Phase
I breadboard testing of the candidate EO EME monitoringsystem,
developing alternative sensor schemes to improve system performance
and extendfrequency coverage, optimizing the RF probe design for
efficient wideband coverage, anddetermining design guidelines for a
concept demonstration of the EO sensor. Phase HI of thiswork will
consist of designing, constructing, and packaging the best
available EO field sensorinto a prototype demonstration system.
System performance will be measured and the environ-mental
ruggedness of the prototype system will be assessed. Shipboard
testing and a subsequentfinal report will finish the advanced
development work assosciated with this effort.
This report describes the work performed in Phase HI (Milestones
3, 4, 5, and 6) of this Task.
1.2 BACKGROUND
Electromagnetic field sensing using EO and fiber optic
techniques has been of interest to theNavy for some time. Previous
work in this area has considered antenna-coupled lithium
niobatebulk crystal and waveguide modulators, where high
sensitivities have been attained for systembandwidths below 1 GHz.
1-3] This class of field sensor, which uses an optical
intensitymodulator for the RF electrical-to-optical conversion, is
referred to as an externally modulatedfield monitoring system. An
alternative EO field sensing approacl has focused on the
directcurrent modulation of an antenna-coupled high-speed injection
laser diode.[451 This is referredto as a directly modulated field
monitoring system. Both approaches use fiber optic links totransmit
the EME information to a remote processing site. Directly modulated
short-haul fiberoptic systems possess a simpler design and are
easier to implement, whereas externally modu-lated systems have
been shown to be more sensitive and possess larger 3-dB bandwidths.
[6.71
Phase I work of this project demonstrated that the externally
modulated EO EME monitoringsystem will ultimately outperform the
directly modulated system and better meet shipboardEMCON monitoring
system requirements. This conclusion was based on the superior
noisecharacteristics of high-power solid-state lasers- (which can
be used for externally modulatedsystems) compared to those of
injection laser diodes and on the superior high-frequencymodulation
characteristics of external waveguide modulators.
As stated above, Phase I[ of this program began in FY 92 and
involved determining solutionsto problems encountered in Phase I
and extending the performance capabilities of the selectedEM field
monitoring system. Anechoic chamber testing of an externally
modulated EO EMEsystem has been performed in the 2- to 18-GHz
frequency range. A root mean square (RMS)electric field sensitivity
of 15 RiV/m and a spurious free dynamic range (SFDR) of 102 dB in
a1-Hz resolution bandwidth have been measured with this 2- to
18-GHz field detection system.These chamber results indicate that
the externally modulated EO EME system can be useful forthe
proposed shipboard applications. Greater sensitivity is expected by
increasing the opticalpower of the laser and by improving the
optical modulator's RF efficiency. Both these modifica-tions have
been implemented. Laboratory testing of these refined EO sensor
systems has beenperformed, and an increased sensitivity and dynamic
range have been attained over the prior
*Available from Amoco Laser Co.
2
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anechoic chamber results. A broadband fiber optic link noise
figure of < 30 dB has beenmeasured, which translates into a 1-Hz
RMS electric field senstivity of approximately 1 jiV/m.
A number of other system modifications have been investigated in
an attempt to work outall the problem areas associated with this
sensor. These Phase II modifications include thedevelopment of
modified optical fiber cable assemblies to minimize polarization
drift, theincorporation of polarization-insensitive semiconductor
modulator designs and polarization-control devices to eliminate the
need for polarization-maintaining fiber, and the operation
oftemperature-insensitive fiber optic links.
Developing more efficient optical modulators and extending their
frequency range to 50 GHzis an ongoing thrust of the Phase II work,
with both in-house and industrial efforts. Theconventional design
for externally modulated EO EME monitoring systems includes
lithiumniobate optical waveguide modulators. Although lithium
niobate waveguide modulators arerelatively mature and available
with bandwidths to 18 GHz,* these broadband devices havecomplicated
traveling-wave electrode designs and possess half-wave voltages in
the 10-V range.An alternative to lithium niobate waveguide
modulators for the externally moduated EO EMEsystem are III-V
semiconductor waveguide modulators. Both interferometric and
absorptionmodulators are possible,18,91 a'hough emphasis in this
report has been on semiconductorelectroabsorption (EA) waveguide
modulators. Both quantum-confined Stark effect[1°0 (QCSE)and
Franz-Keldysh effect['1I (FKE) EA devices operating at 1.52 Itm and
1.32 gtm, respectively,are included. Both of these devices rely for
operation on material absorption coefficient changeswith an applied
electric field. Fiber optic link sensitivity. spurious free dynamic
range (SFDR),and bandwidth predictions using these EA devices based
on theoretical and experimental resultsare presented and compared
to those of lithium niobate-based fiber optic links.
A final photonic probe design for the FY 94 shipboard
demonstration has been completed.Fiber optic link implementations
include a power-by-light (PBL) system to optically power
themodulator as well as a modulator bias-control circuit to
automatically adjust for environmentalchanges. These two system
insertions improved the usefulness of the EO field sensor, and
theresults of testing these system modifications will be
presented.
Phase II antenna development efforts include the development of
a compact HF/very highfrequency (VHF) antenna as well as a wideband
spiral antenna useable to 50 GHz. Advancedantenna developmental
efforts include the development of compact HF/VHF antennas.
Oneapproach being pursued is the development of a small-aperture,
high-dielectric spiral antenna,which should provide useable gain
down into the HF band. Other low-frequency antennacandidates
include an enlarged spiral antenna, a sectoralized monopole
antenna, and an electri-cally short dipole antenna. Results using
these HF/VHF/ultrahigh frequency (UHF) antennastructures are
presented. A miniature spiral antenna design with gain to 50 GHz is
also beingdeveloped and will be discussed. These antenna R&D
efforts will continue into FY 94.
An initial test plan for the shipboard demonstration of the
photonic EM field probe has beendeveloped. The prototype sensor
will be placed at selected topside sites of an LSD-41 classsurface
ship and controlled emissions will be recorded. Detailed testing
procedures will bedescribed.
*Available from United Technologies Photonics.
3
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2.0 SHIPBOARD EO EME MONITORING
In this section, an externally modulated EO EME monitoring
system is described andlaboratory results on a number of
breadboarded systems are presented and compared withsimulations.
The photonic link as well as the wideband RF probe will also be
discussed.
2.1 DESCRIPTION OF EXTERNALLY MODULATED EO EME
MONITORINGSYSTEM
A schematic diagram of an antenna-coupled externally modulated
EO EME monitoringsystem is shown in figure 2-1. Its primary
components include a high-power, low-noise,polarized optical
source, polarization-maintaining single-mode optical fiber (PMF)
for theuplink, an antenna-coupled optical waveguide modulator,
standard single-mode fiber (SMF) forthe downlink, a high-speed
p-i-n photodiode, and a signal processor. The signal processor
willnormally consist of a wideband spectrum analyzer and a central
processing unit. Low-noisepreamplifiers are not included in the EO
EME monitoring system configuration and are notconsidered necessary
for this relatively high signal level application. The antenna and
opticalmodulator are positioned at a selected point about the ship
for EM field detection, whileamplitude and frequency information is
remotely received and processed. The intent of thisgeometry or
configuration is to minimize the perturbation of the EM field by
eliminating anyelectrical transmission lines between the probe and
receiving station. If passive modulatorbiasing is not feasible,
electrical powering or biasing of the optical modulator can be
achievedoptically via a high-power laser diode, a multimode optical
fiber, and photovoltaic cells foroptical-to-electrical power
conversion. This optical powering scheme is generically shown
infigure 2-2 and its implementation is discussed in Section
3.1.
Initial laboratory prototypes of the EO EME monitoring system
operate from 2 to 18 GHzand use a wideband spiral antenna which
possesses a 50 Q output impedance and a minimumgain of 0 dBi across
this band. A system is presently being constructed for shipboard
testing,which is operable from 2 MHz to 18 GHz. This prototype
system has been designed to operatewith a number of different
optical modulator types that are being developed as part of this
R&Deffort and that are described below. A four-fiber, all
dielectric optical cable has been designedand fabricated for this
prototype system; it allows for analog EME information as well
asmodulator biasing signals to be transferred to a remote site. The
optics of the sensor head ispackaged in a plastic box that mates to
the wideband antenna via an SMA-connector. Antennaor probe output
powers ranging from less than -60 dBm to greater than 20 dBm can be
expectedat some topside positions.[12] This implies that the
monitoring system must realize an SFDRexceeding 80 dB and a
sensitivity below -60 dBm. This system performance must eventually
beattained from 2 MHz to 50 GHz by some combination of field
detection units. This will beaccomplished by combining the
broadband RF information from the banded antennas onto asingle
fiber optic link, as is shown in figure 2-3. Hence, optical
modulators with the potentialfor extremely wide passband operation
are required for this shipboard application. The opticalwaveguide
modulator alternatives being pursued in this work will now be
discussed.
4
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PMF ( I lM).oaOTVLASER
MULTIOCTAVEANTENNA
Figure 2-1. Schematic diagram of an antenna-coupled
externallymodulated EQ EME monitoring system.
PHOTOVOLTAIC
MODULATOC
OPTICAL FIBER
Figure 2-2. Optical powering scheme using GaAs photovoltaic
cellsfor the E O EME monitoring system.
5
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2-500-MHz O.5-18-GHz 18-50-GHzANTENNA ANTENNA ANTENNA
Figure 2-3. Triplexing scheme to combine RF information from 2
MHz through
50 GHz.
2.1.1 Lithium Niobate Optical Waveguide Modulators
The most used crystal for producing broadband optical waveguide
modulators is lithiumniobate. This material has a large linear EO
coefficient and low-loss single-mode waveguidescan be easily formed
by using titanium indiffusion or proton-exchange techniques. MZ
wave-guide modulators fabricated with traveling-wave electrodes on
x-cut lithium niobate are the mostpromising of the commercially
available high-bandwidth modulator devices. A schematic ofthis
interferometric waveguide modulator is shown in figure 2-4. In
these devices, light iscoupled from a single-mode optical fiber
into a lithium niobate waveguide. The light is equallysplit at the
input Y-branch and then recombined at a similia output Y-branch.
The light in onearm of the interferometer is phase-modulated by an
RF signal and results in intensity modulationfor the combined
output optical waveguide mode. The intensity-modulated output
opticalpower is then coupled back into a single-mode optical fiber
for transmission. The expression forthe output optical power as a
function of the input optical power for an ideal MZ modulator
isgiven by
Sout(V) = Sin tm cos2 (xV/2Vx) (2.1)
whereSi = input optical powertm = modulator transmission loss
factorV = modulator half-wave voltage
6
-
sPLITTrNG RECOMBINATION
Figure 2-4. Schematic of lithium niobate MZ interferometric
optical i dItor.
For analog applications, in addition to the RF signal, a DC bias
of V,/2 is applied to achievemaximum RF sensitivity. The highest
speed commercially available lithium niobate MZmodulators possess
an 18-GHz modulation bandwidth, a V. of approximately 10 volts, and
a tmof approximately 0.5.* An x-cut lithium niobate MZ modulator
with performance speciicationsclose to this has been procured and
tested in this effort. The x-cut crystal was chosen over thez-cut
crystal because of its demonstrated superior thermal stability.
This device was used in the2- to 18-GHz laboratory system that was
tested and whose experimental results will be discussedin Sections
2.3 and 2.4. Alternative optical waveguide modulators that have
attracted consider-able recent attention are II-V
semiconductor-based interferometric and EA modulators. Theseare now
discussed.
2.1.2 rn-v Semiconductor Optical Waveguide ModulatorsGaAs-based
MZ modulators operating at 1.3 tiun are beginning to become
available. The
operation of these modulators is identical to that of the
lithium niobate modulators discussedabove. Better phase matching
between the optical and electrical signals is expected with theGaAs
traveling-wave modulator than with the lithium niobate modulator.
This should allow forhigher bandwidth devices without sacrificing
modulation efficiency. Obtaining low opticalinsertion loss is a
remaining challenge for the GaAs MZ modulators. A DC to 50-GHz
proto-type GaAs modulator possessing an RF Vx = 10 V and a 10-dB
optical insertion loss is to bedelivered to the Navy for use in
this effort. For the bandwidth, the modulation efficiency of
thisdevice exceeds any available lithium niobate modulator,
although the optical insertion loss issignificantly higher (10 dB
compared to 3 dB). When this semiconductor modulator is received,it
will be tested and evaluated for use in this shipboard application.
Another promising Ill-Vsemiconductor optical modulator is the EA
waveguide modulator. This type of modulator willnow be
discussed.
Semiconductor EA waveguide modulators fabricated in either a
p-i-n or p-n junctionstructure are also being considered for this
analog fiber optic application. Here, a reverse bias
*Available from GEC Advanced Optical Products.
7
-
across the junction modulates the electric field in the
waveguide and changes the absorptincoefficient of the material. The
semicductor EA waveguide modulator possesses an exponen-tial
optical power transmission relation given by
Sow(V) = S ntm exp[ - FAa(V) L] (2.2)
whereSmn - input optical powern = modulator ansmission loss
factor
L = waveguide lengthF = optical mode and active absorbing layer
overlap integralzia = change in absorption coefficient
In quantum well structures, Aa is due to the QCSE. In thick
(> 500 A) or bulk semiconductorlayers, Aa is due to the FKE.
QCSE modulators typically have a largeeda at low appliedvoltage,
but possess a relatively small F, while FKE modulators typically
have a small Aa, but alarge F. It can be seen from Eq. 2.2 that
these two parameters equally affect the modulator'sperformance and
therefore both devices can obtain comparable modulation
performance.Prebiasing the EA modulator to the quasilinear region
enables analog operation in the samemanner as with MZ modulators.
The modulator linearity can be assessed by measuring theharmonic
and intermodulation distortion about a given bias position, which
ultimately deter-mines the SFDR.
Two modulators, a 1.52-1mn QCSE modulator and a 1.32-pm FKE
modulator, have beenfabricated and tested for use in the EO EME
monitoring system. The liquid phase epitaxy (LPE)grown 1.32-pun FKE
modulator is schematically shown in figure 2-5 and uses an
InGaAsPactive absorbing waveguide layer. This ridge-waveguide
modulator has a device length of 300fpm, a waveguide thickness of
0.4 pm, a device capacitance of 0.2 pF, and a r x- 0.7. For
digitalapplications, an extinction ratio of > 30 dB at less than
10 V has been obtained with thismodulator.
The 1.52-pum QCSE modulator uses an InGaAs/InP quantum well
ridge-waveguide structureand was grown using metalorganic chemical
vapor deposition (MOCVD). A schematic of thismodulator is shown in
figure 2-6. This device has a length of 650 pm, an undoped
superlatticewaveguide thickness of 1 pm, and a r = 0.1. It is the
ten 70-A InGaAs quantum wells buried inthe center of the
superlattice waveguide of this modulator structure that are
responsible for the1.52-pm absorption modulation. This waveguide
modulator has also displayed a > 30-dBextinction ratio at less
than 10 V applied bias. For more details about these specific
modulators,the interested reader is referred elsewhere.[13.141
Analog photonic links and electromagneticfield monitoring systems
employing these optical modulator types will be analyzed next.
8
-
AW~n P4:WNACT
- AePLATID AS lE P 4WAVEGUWDE MOVE 4
Figure 2-5. Structure of LPE-grown 1.32-Iun InGaAsPFKE optical
modulator.
n. - InPSUBSTRAT CONTACT
Fiur 2-.SrctrLfMOV-rw 1.52m Ism p -sP/lODmultiple0 quan-- well
QCSE opj a noduator
100 S In9
-
2.2 ANALYSIS OF EO EME MONITORING SYSTEM
The externally modulated EO EME system described in Section 2.1
will now be theoreticallyanalyzed. Performance will be predicted
and important noise sources identified. Particularattention will be
paid to the performance characteristics of the optical modulator.
For digitalapplications, a modulator's usefulness is determined
primarily by its bandwidth, switchingvoltage (bias required to
obtain a 10-dB extinction ratio), and optical insertion loss. For
analoglink applications like the one at hand, the modulator's
bandwidth, linearity, RF efficiency, andoptical insertion loss are
important. These parameters will now be discussed for both
MZmodulator and EA modulator-based systems, and a performance
comparison analysis will bemade between the lithium niobate and
semiconductor modulator options.
To assess the relative usefulness of the candidate optical
waveguide modulators discussed inSection 2.1, an externally
modulated fiber optic link was breadboarded. Modulator
characteris-tics were measured for the FKE and QCSE modulators and
compared to those of a high-speed1.32-nIm lithium niobate MZ
waveguide modulator. The MZ modulator performance specifica-tions
used for this comparison are a V, = 10 V, a fiber-to-fiber
insertion loss of 3 dB, and a 3-dBelectrical bandwidth of 18 GHz,
specifications which at this time compare favorably with
othercommercially available MZ modulators. As described in the
previous Section, the releventQCSE modulator parameters for
transverse magnetic (TM) mode operation are a device lengthof L =
650 gjm, a confinement factor of F = 0.07, and a device capacitance
of 1.2 pF. Therelevent FKE modulator parameters are a device length
of 300 Iim, a confinement factor of 0.7,and a device capacitance of
0.2 pF.
The relative transmission versus applied bias or optical
transfer functions for the FKE,QCSE, and MZ modulators are shown in
figure 2-7. The important aspects of these curves forthis analog
application are (1) the bias required to obtain quasilinear
operation, (2) the slope ofthe curve at the bias point, which is a
measure of the RF efficiency, and (3) the linearity aboutthe bias
point. These aspects will now be investigated.
The externally modulated fiber optic link RF insertion loss can
be expressed asH 2' K' r2 oR r2 (2.3)
f d Out
whereKf = optical loss from source to detector (not including
modulator insertion loss)rd = detector responsivity (AIWo)Rout=
detector output resistance (0)r = modulator RF efficiency factor
(Wo/We -).
For an MZ modulator
r12 = (tin Po ar/VX)2 (Rm/2) (1-p2) (2.4)
where P0 is the laser power, tm is the modulator transmission
factor, Rm is the modulator input resis-tance, and pm is the
modulator input RF reflection coefficient. For an EA modulator, rm
is givenby
r2 = (tin Po)2 (Rm/2) (1-p 2) [/L (dAa/dV)vb]2 (2.5)
10
-
1.0
0.8
PAZ
0.4
n-
0.2OCSE
0 .0 -
0.0 2.5 5.0 7.5 10.0
APPUED BIAS (V)
Figure 2-7. Relative transmission versus applied bias curves for
the1.32--tm FKE waveguide modulator, the 1.52-!.n QCSE
waveguidemodulator, and the 1.32-pm lithium niobate MZ waveguide
modulator.
where the derivative is taken at the modulator bias voltage, Vb.
Measured values for A a as afunction of voltage for the QCSE and
FKE modulators are shown in figure 2-8. It is found that(da/dV)
evaluated at Vb - 2 V is equal to 150 (cm-V)-1 for the QCSE
modulator and 50(cm-V)"' at Vb - 6 V for the FKE modulator. If we
assume that P, = 50 mW, Rm = 50 Q, P,=0, and tm = 0.10, we find
that rFKE2 = 6.9 x 10-4 (Wo2/We)and rOCSE2 = 2.9 x 10-4 (Wo2/We).
Forthe MZ modulator, we will also assume Po = 50 mW, Rm = 50 Q Pm=
0, but a lower opticalinsertion loss yielding tm = 0.50. Using
these numbers gives rMz 2 = 1.5 x 10.3 (Wo2/We) whichis slightly
larger than that of the FKE modulator. This result is entirely due
to the presentlyencountered larger EA modulators' optical insertion
loss (10 dB versus 3 dB). Hence, eventhough the insertion loss of
the EA modulators is substantially larger than that of the
MZmodulator, the RF power efficiencies are comparable due to the
larger modulator optical powertransfer function slopes evident in
figure 2-7. If the insertion loss of the EA modulators can
beimproved to the same level as the MZ modulator, then efficiencies
of rFKE2 = 1.7 x 10-2
(Wo2/We) and rOCSE2 = 7.3 x 10-3 (Wo2/We) are obtained, which
are significant improvementsover current state-of-the-art MZ
modulators. If we further assume that Ro.t = 50 Q, rd = 0.85A/IW,
and Kf = 0.1 (including 3 dB for biasing of the modulator), then
the expected RF insertionloss is -33 dB for the MZ system, -22 dB
for the FKE modulator system, and -26 dB for theQCSE modulator
system. These results translate into a lower system noise figure
(NF) for theEA modulators if low optical insertion loss devices can
be developed. It is important to note thatthe QCSE modulator
structure used in these experiments is not optimized. This
optimization
11
-
process involves improving the material quality of the quantum
wells as well as modifying themodulator geometry to increase r.
When optimized, the performance of the QCSE modulatorshould be
comparable to that demonstrated by the FKE modulator.
2MOO
OCSE
-1500
0E0
LU
500
FKE
0~
0 2 4 6 8 10APPUED VOLTAGE (V)
Figure 2-8. Experimental values for Aa versus applied bias for
the1.32- jm FKE and 1.52-Rm QCSE waveguide modulators.
The SFDR of the MZ, FKE, and QCSE modulator systems will now be
addressed. For theexternally modulated link, with the above-used
laser, detector, and laser-to-detector lossparameters, a detector
optical power of 2.5 mW is expected for each link. Again, this
assumes4,, = 0.5 for all three systems. With these detector optical
power levels, as will be discussed inSection 2.2.2, the fiber optic
links are shot-noise-limited. This assumes a laser relative
intensitynoise (RIN) of -165 dBc/Hz, which is well below the
thermal and shot-noise contributions atthese detector optical
powers. This demonstrates the importance of minimizing the
modulatorinsertion loss as well as using a high-power, low RIN
laser in obtaining shot-noise-limitedoperation.
The receiver signal power and spurious signals due to
intermodulation products and harmon-ics for the MZ, FKE, and OCSE
modulator systems are graphed versus modulation index infigures 2-9
to 2-11, respectively. A normalized receiver bandwidth of 1 Hz,
which determinesthe system noise floor, is used for these
simulations. The result is that a 108-dB SFDR in a 1-Hzbandwidth
can be expected for the MZ system. This result agrees well with
actual two-tonedistortion measurements performed on this link which
yielded a 102-dB/Hz 2/3 SFDR with a10-mW laser power and a
109-dB/Hz2/ 3 SFDR with a 50-mW laser. Ideal biasing at
thequadrature point is assumed for this calculation. The spurious
signals are obtained by expanding
12
-
-4064
S-80 B
10- SI•NAL100i~-140 14
-1604-
-180:- NOISE FO / .MoNI 18O
L0OULATION INDEX
Figure 2-9. Optical receiver output power versus modulation
index, includingharmonics and intermodulation products for the
1.32-jum MZ modulator.
102dB,.. I
S-90HNI INTERMODVD
2: - 100- SINL-120:62
o 143HARhIONC-4. 160
NOISE FLOOR tB
-20 ....LI.ZV /"-L I .. LAJL."" .~LUi2Z0 M0 0 CV 0
WI W W W
MODULATION INDEX
Figure 2-10. Optical receiver output power versus modulation
index, including
harmonics and intermodulation products for the 1.32-jun FKE
modulator.
13
-
- 20E
I ] I I I I I I
IJ LW Li Li .i Li L
103OIIATION INDEX
Figure 2-11 . Optical receiver output power versus modulation
index, includingharmonica and intermodulation products for the
1.5210m QCSE modulator.
the modulator optical power transfer function in a Taylor series
and evaluating the nonlinearterms appropriately.[15] Using the
experimental modulator transfer curves of figure 2-7, SFDRsof 102
dB and 105 dB in 1-Hz bandwidths are obtained for the FKE and QCSE
modulatorsystems, respectively. Hence, not much difference in
dynamic range performance is expectedusing either an BA or MZ
optimally biased modulator. At the chosen bias positions, all three
ofthese modulators were limited by the third-order intermodulation
product and not the fundamen-
tal harmonics. For broadband systems such as in shipboard EME
monitoring, the harmonics aswell as the intermodulation products
are important It is important to note that the resultantdynamic
ranges for the BA modulators were not as sensitive to bias position
as the MZ modula-tor dynamic range within the 0.2 to 0.8 relative
transmission ranges. fiis indicates that theSFDR performance of the
exponential transfer function is less sensitive to bias position
than thecosine-squared transfer function. This can be important in
field-deployed situations whereenvironmental fluctuations, which
can affect the modulator bias position, are unavoidable. Inthe case
of temperature fluctuations, significant modulator bias point drift
can be expected.Hence, the FA modulators seem to be very
competitive with the state-of-the-art lithium niobateMZ modulator
in link performance.
The electrical power out of the antenna can be expressed in
terms of the externally modu-lated fiber optic link parameters
as
P = (tin P0 m/2rn)2 (2.6)where f is the modulation index. For
the MZ system, this gives
PA= (in2 V=)/[2g2 Rm (1 - p2](2.7)
14
-
and for the EA system
= m2/[2 Rm (I - pI) P L 2 adV)2)](2.8)
Using the minimum detectable modulation indices given in figures
2-9 to 2-11, the minimumdetectable antenna power in a 1-Hz receiver
bandwidth is PA (min) = -135 dBm for the MZsystem, PA(min) = -146
dBm for the FKE system, and PA(min) = -142 dBm for the QCSEsystem.
Again, these numbers assume equal modulator optical insertion
losses of 3 dB, anassumption which remains to be demonstrated for
the semiconductor devices. Nevertheless,excellent link sensitivity
is expected, which translates into fairly low electric field
sensitivities.
The system bandwidth of the EO EME monitoring system is
ultimately limited by themodulation bandwidth of the optical
modulators. For the MZ modulator, the bandwidth is 18GHz with an
assumed V. = 10 V. A high-frequency V, of closer to 30 V is
actually measuredwith this lithium niobate MZ modulator. If the
manufacturer-specified V, value is used, a0.56 V/GHz
bandwidth-efficiency figure of merit results, which when
extrapolated to beyond 40GHz, implies devices with V. - 25 V. For
the EA modulators, device capacitances in the0.2-pF range are
routinely attained with the FKE devices, which implies cutoff
frequenciesgreater than 30 GHz with Vb = 5 V. An experimental
frequency response curve of the the1.32-$tm FKE modulator is shown
in figure 2-12. At 20 GHz, the highest measurementfrequency
currently available, the response is less than 1 dBe down from its
low-frequencyresponse. This device possessed a 7-V on/off voltage,
which translates into a bandwidth-effi-ciency figure of merit of
better than 0.5-V/GHz, which already exceeds that of current
lithiumniobate MZ modulators. With optimized EA modulator
structures, it is expected that the biasvoltages required to obtain
these extremely large bandwidths can be reduced to less than 5
V,which is a significant improvement over that attainable with
comparable lithium niobate MZmodulators. A 1.5-pmo quantum well
waveguide modulator with a 3-dBe bandwidth exceeding40 GHz at 5-V
bias voltage has been reported,[ 161 which does demonstrate the
usefulness ofthese modulators for ultrawideband applications.
The most useful and revealing parameters to evaluate the
performance of the fiberoptic sensing system are the link signal
to-noise (S/N) ratio and the link NF. The S/N ratio willbe
considered first. The available output electrical signal power is
given by
Psig = (m2/2) 12 Rout (2.9)
where ID is the optical detector time averaged photocurrent, and
the other parameters werepreviously defined. Three system noise
sources are considered important: shot noise, thermalnoise, and
laser RIN noise. The available shot noise power is given by
Psho = 2 q ID Row B (2.10)
where q is the electronic charge and B is the receiver
bandwidth. The available thermal noisepower is given by
Pth = K T B (2.11)
15
-
>521 ~~log MA13KME -REF -70.0 dB W/A 6.6 Hz
S3.6 d8/ 6.6 dO-0. 9023 dO
F'rmi'nic" Modulator -L - .-- is 187 H
SCA-E -6.9623 dBs 3.0 IB/€ iv
H _
A
S6 Jan 1994 15:45START 6. 136660666 0H-zSTOP 26.666666666
0Hz
Figure 2-12. Frequency response curve for 1 .32-tpn LPE-grownFKE
modulator.
where K is Boltzman's zonstant and T is the receiver
temperature. At room temperature, Pth =
-174 dBm/Hz. The available laser noise power is given by
PL = 12 RIN Ro, B (2.12)
where the RIN value is usually specified for each laser or can
be measured. The inverse of the linkS/N ratio can then be expressed
as
1/(S/N) = 2B/r 2 [(RIN) + (2 q/0D) + (KT/RoPJID)] (2.13)
where the first term in the bracket represents the laser noise
contribution, the second term representsthe shot noise
contribution, and the third term represents the thermal noise
contribution. Dependingon the values of RIN, Rout, and ID, the
system is either shot-noise-limited, RIN-limited, or
thermal-noise-limited. A plot of m21(S/N) in a 1-Hz bandwidth
versus ID is shown in figure 2-13 for Rout= 50 Q and different
values of the laser RIN. This curve demonstrates the importance of
obtaininga laser with a very low RIN value.
The fiber optic link NF is probably the most important parameter
which can be measured, for
it ultimately determines the detection sensitivity limit. The
link NF is defined as
NP = 10 loglo [(S/N)jnpu(S/N)outPUt] (2.14)
which can be expressed in terms of the link parameters as
NF - - 10 log o(H2) + 10 log10 [Pth(l + H2) + PL + Pshaot]IPh
(2.15)
16
-
-110 . .
-120:
-130'-
-140S"• RIN =-1:30 dBc/Hz
-150
CI
-160- -145 dBc/Hz-
-170-
-180-173 dBc/Hz
-190-
10-6 i0-5 i0-4 10-3 10-2 10-1 100
DETECTOR PHOTOCURRENT (A)
Figure 2-13. Plot of m2/(SIN), where m is the modulation index,
in a 1-Hzbandwidth versus optical detector photocurrent for
different laser RIN values.
where H2 is the RF insertion loss given by Eq. 2.3. It is
assumed in this NF expression that theinput and output thermal
noise powers are identical. For the MZ and EA modulator links,
theexpected NFs are 28 dB for the FKE system, 32 dB for the QCSE
system, and 39 dB for the MZsystem. Again, a clear advantage for
the EA modulators if low optical insertion loss can b,achieved.
Experimental results from a breadboarded MZ fiber optic link will
now be discussedand compared with these simulated results.
2.3 PERFORMANCE OF FIBER OPTIC LINK
In this section, the best available 2-MHz to 18-GHz fiber optic
link will be described. Adiagram of the externally modulated fiber
optic link is shown in figure 2-14. The 1.32-pimNd:YAG laser (Amoco
Model ALC1320-56S) has a fiber pigtail output power of 45 mW and
ameasured RIN value of approximately -170 dBc/Hz. The lithium
niobate MZ modulator(GEC-Marconi Model Y-35-8808) has an optical
insertion loss of 3.6 dB and a low frequencyV, = 13.5 V. The
modulation frequency response of this modulator is shown in figure
2-15,which reveals a 3-dBe bandwidth of approximately 17 GHz. The
InGaAs p-i-n photodiode(Fermionics Model HSD-30) has a
fiber-coupled responsivity of rd = 0.7 A/W at an opticalwavelength
of 1.3 fVm. A 3-dBe modulation bandwidth exceeding 16 GHz was
measured forthis optical detector. The overall fiber link frequency
response is plotted in figure 2-16, whichshows a 10-dBe falloff in
response from 50 MHz to 18 GHz (3-dBe bandwidth of 11 GHz).The
fiber link loss budget is depicted in figure 2-17, which displays
an optical loss from laserto detector of 10 dB, a value that
includes the 3 dB for modulator biasing. The total noisepower (Pth
+ Pshot +PL) of this link as a function of optical detector
photocurrent is plotted in
17
-
45-MW SINGLE-MODE1.32-pm Nd:YAG LASER
15-GHzLITHIUM NIOBATE
U Z MODULATOR
RFRF INPUT
SMF (300 m)
18-GHzInGaAs p-i-nPHOTODIODE
Figure 2-14. Schematic diagram of best available DC to
18-GHzexternally modulated fiber optic link.
REF -29.98 dDW/A U.0. HzS3.0 dlW 0.0 do-2.916 di s
> )MARKER 2-11SC, '17 .0 5 2 G 1"1 'A 'SCALE : ;,-2.916
dD
s 3.0 NB/div v _ l '.
HI .. I
S7RT 0.SSSSS .:10 Fob 1994 15:33
STOP 19.000000000G mi
Figure 2-15. Modulation frequency response of wideband
lithiumniobate optical modulator.
18
-
>s21 log MnA nwm]l-i=•1RF-40-0 01B 0.0. HaS3. 0 411/ 9.0
do-7.2998 dB4
| _!_____�_______IM )W4ER 2-1'r-C--------- 16.065 -Hz:AISCALII I
Z 1-7.2999 dl,S I 3.0 •IB/4
IAH .-- -- - . . t- --- i ° ---- • '
SH
H-- rI - - t' -- I •IN,
*I I I l _
START 0. 656666660 0HzDIP~ 18.000000090 0Hz
-Figure 2-16. Externally modulated fiber optic link RF insertion
lossversus frequency in the 50-MHz to 18-GHz range.
(45 mW) (4.5 mW)
~ * ~ 0EiCTOR.
_ -.. dB -1 dB'tAS I I II
-7 dBFigure 2-17. Diagram of the externally modulated fiber
optic link loss budget.
19
-
-150 .-155- DATA OPERATING I /
-155 ~POINT j/-160-
-- - 7-THERMALR-175-
Z SHOT /a- /
-8/ RIN --173 dBc/Hz-a-190
I II I I IIl10-6 10)-5 10-4 10-3 10-2 10-1
DETECTOR PHOTOCURRENT (A)
Figure 2-18. Total noise power (solid line) and experimental
data (circles) versusoptical detector photocurrent for the
externally modulated fiber optic link.
figure 2-18, which shows that the operating point for this
system renders shot-noise-limited opera-tion. The RF insertion loss
of this link was measured to be 32 dB at 2 GHz. This is within 3 dB
ofthe theoretical prediction using Eq. 2.3. The link NF was
measured to be 35 dB, 38 dB, and 45 dBat frequencies of 2, 10, and
18 GHz, respectively. The insertion loss and NF values can be
improvedupon by increasing the laser power and decreasing the
optical modulator V= voltage. For compari-son, the RF insertion
loss versus frequency of a 2-MHz to 500-Mi-z fiber optic link using
a opticalmodulator possessing a V. of 4 V is shown in figure 2-19.
Here, a link RF insertion loss of 11 dBand an NF of less than 20 dB
are obtained, which demonstrates the importance of maximizing
themodulation responsivity of the optical modulator. With continued
laser and ultrawideband modula-tor R&D, the potential for fiber
links with low RF insertion loss can be expected. Two-tone
measure-ments with the 18-GHz fiber optic link show an SFDR of 108
dB/Hz2/3. The SFDR is smaller thanthe linear dynamic range for this
wideband fiber optic system which implies a linear dynamic
rangegreater than 110 dB/IIz. This link will be field-tested and
subsequent shipboard measurements willbe acquired.
2.4 ELECTROMAGNETIC FIELD MEASUREMENTS IN THE 2- to
18-GHZFREQUENCY RANGE USING AN EO EME MONITORING SYSTEM
Remote electromagnetic field measurements in the 2- to 18-GHz
frequency range using anantenna-coupled externally modulated fiber
optic link will be described in this Section. Thesemeasurements
have been made with a fiber optic link whose performance was not
quite as goodas that described in the previous Section.
Nevertheless, an electric field sensitivity of 15 WtV/mand an SFDR
of 102 dB in a 1-Hz bandwidth have been measured. The results
indicate that thissystem is a feasible candidate for remotely
measuring extremely broadband, large dynamic rangeelectromagnetic
fields, especially when recent fiber optic link improvements are
incorporated.
20
-
)Oal leg W
1. -. a i
Figure 2-19. RF frequency for a DC to
500-MHz fiber optic link using a V/ = 4 V optical modulator.
2.4.1 System Configuration
A schematic diagram of the antenna-coupled electromagnetic (EM)
field detection systemused in this early work is shown in figure
2-20. It consists of a cavity-backed spiral antenna(Transco Model
9C33500), a Ti:indiffused lithium niobate MZ waveguide modulator
(GEC-Marconi Model Y-35-8808), a 1.32-tim Nd:YAG solid-state laser
(Amoco ModelALC1320-10S), a 100-meter length of PMF single-mode
optical fiber (Alcoa-Fujikura ModelN000185), a 100-meter length of
SMF (Coming Model SMF28), and a high-speed InGaAs p-i-nphotodiode
(BT&.D Model PDC4310-30-FP). The electrical output of the
broadband antennawith known characteristics is coupled to the
optical modulator and is optically remoted via PMFfor the upiink
and SMF for the downlink, with the laser and optical detector
remotely locatedwith the associated processing electronics. In an
effort to limit the electrical power requirementsand intrusiveness
of the EM field probe, this configuration contains no low-noise
amplifier at theinput to the optical modulator. The externally
modulated fiber optic link is slightly differentthan that described
in Section 2.3. The Nd:YAG laser has an output power of 10 mW and
ameasured RIN of -173 dBc/Hz. The lithium niobate MZ modulator has
a 3-dBe modulationbandwidth of 18 GHz, a 3.6-dB optical insertion
loss, and a low-frequency half-wave voltage ofVx = 13.5 V. The
optical detector has a responsivity of rd = 0.85 A/W and a 3-dBe
modulationbandwidth of 25 GHz. The fiber optic link and antenna
performance were first measured andcalibrated separately and then
mated for anechoic chamber testing of the entire EM fielddetection
system. The results of these experiments will now be discussed.
21
-
I~TQPOJ~ PUF (100 m)SMF~t (100-.NM)
Figure 2-20. Schematic diagram of the antenna-coupled
externallymodulated fiber optic link for EM field detection.
2.4.2 Results
The fiber optic link and spiral antenna individual system
responses in the 2- to 18-GHzfrequency range are shown in figure
2-21. The close to 6-dBe roll off in response for the fiberoptic
link is due to the combined response of the integrated optical
modulator (IOM) and theoptical detector. The optical insertion loss
of this link is 9.0 dB, which includes the 3 dB loss forIOM
biasing. The RF insertion loss of this link was measured to be 45
dB at a frequency of2 GHz. A lower RF insertion loss and hence
better performance has since been obtained byemploying a higher
power laser. The effect of laser optical power on fiber optic link
perfor-mance is better illustrated in figure 2-22, which graphs the
inverse S/N ratio as a function ofdetector photocurrent. The
individual thermal, shot, and RIN noise contributions are plotted
toillustrate the thermal-noise-limited, shot-noise-limited, and
RIN-limited regions of operation,The operating point for this fiber
optic link is borderline between thermal-noise-limited
andshot-noise-limited. More optical power would increase the SIN
ratio and decrease the NF of thelink. Link NF values of 49 dB, 52
dB, and 55 dB have been measured at modulation frequenciesof 2 GHz,
10 GHz, and 18 GHz, respectively. Two-tone distortion me ents were
per-formed near 9.5 GHz to determine the SFDR of the link. The
theoretical and experimentalresults of these measurements are
summarized in figure 2-23. A 102-dB/Hz2 SFDR has beenempirically
measured by fitting the experimental data points for the
fundamental and third-orderintermodulation power levels to
theoretical curves. Excellent agreement is obtained when anIOM Vx
of 30 V is assumed instead of the 13.5-V low-frequency value. This
discrepancy is notthat alarming considering that the two-tone
measurements were performed at 9.5 GHz and theVx of 13.5 V was
essentially measured at DC.
22
-
- .40n dulo~flM *** ll',% I* -" - -ý Z-
SII,
1 11.01 • 0. a1, do
STAWVr 2. 9Wz 22 Now 181l l0S1STO 19.0000o1, o 9 4z
(a) Fiber optic link.
4--
2-6---- e0-"
-2 VS ?R: 2.:
-4"POIARITIONILHC
"BEIMWtTH (3dB): D DE
-101-
2 4 6 8 10 12 14 16 18FREQUENCY (GHz)
(b) Spiral antenna.
Figure 2-2 1. Modulation frequency response from 2 to18 GHz of
the externally modulated fiber optic linkand the spiral
antenna.
23
-
-110
a DATA-120
THERMAL NOISE LMT
-130
-140
1 "5- OPERATING
SHOT NOISE LII-160-
-170-
-180
-190 RIN LIMIT
10-6 105 10-4 10-3 10-2 10-1 100
DET'ECOR PHOTOCURRENT (A)
Figur 2-22. Plot of m21(S/N), where m is the modulation index,
fortotal noise (solid line), shot noise, thermal noise, and RIN
contributionsversus detector photocurrent for the fiber optic link
used in this work.
"-20- MODULATOR VX z 30 VRIF INSERTION LOSS a, 48.9 dB v
.;\,
-40 THIRD ORDER INTERCEPT 3 8.1 dBmSFDR a95.3 dB
-60 NOISE BANDWIDTH= 1 Hz
-go-
-100-1
Ef
-160 NOISE FLOOR
-120 -100 -80 -60 -40 -20 0 20 40
MODULATOR DRIVE POWER (d0l)
Figure 2-23. Plot of experimental (circles) and theoretical
detectedpower for fundamental (solid line) and third-order
intermodulationproducts (broken line) versus power applied to the
IOM.
24
-
The wideband RF probe used for this work is a 2-inch-diameter
cavity-backed spiral antennawhose gain characterisitics were given
in Figure 2-2 1b. It is a left-hand circularly polarizedantenna
that has a 3-dB beamwidth of 80 degrees and a voltage standing wave
ratio (VSWR) ofless than 2:1 across the 2- to 18-GHz frequency
range. The gain versus frequency characteristicswere measured
against standard gain horn antennas in the 2- to 18-GHz range for
linearlypolarized radiation, and > 0 dBi gain exists for
frequencies above 4 GHz. The circularlypolarized spiral antenna is
a good choice for multioctave frequency coverage if a small,
passivereceive antenna is required.
EM field measurements were made with the antenna-coupled fiber
optic link after separatetesting of the antenna and link was
completed. A 2- to 18-GHz far-field anechoic chamber wasused in
which to perform the experiments. Measurements were performed at
2.25, 9.52, and 16GHz using three different standard gain horn
source antennas. The spiral antenna outputelectrical power was
calculated from the expression
PA = Aef Winc
= (A2GA/4,ir) Winc (2.16)
= (PM,) (Gtr) (GA) ( 2/4xR)2
where
Aeff effective aperture of spiral antenna (m2)
inc f RF intensity incident on antenna (W/m 2)
Pin = RF power applied to transmit antenna (W)
Gtr = gain of transmit antenna (dBi)
GA = gain of spiral antenna (dBi)
X. = RF source wavelength (m)R = propagation length (m)
The "actual" electric field strength at the probe site was
calculated from the expression
EA = (2 t/ Wnc) 1/ 2 (V/m) (2.16)
where ij = 377 Q is the free space impedance. The "measured"
electric field strength wasobtained using Eq. 2.17 along with the
fiber optic link RF insertion loss data. Starting with
theelectrical power out of the optical detector, PA was deduced by
factoring in the fiber optic linkloss. From PA, the incident RF
intensity WV11 was determined, and from that the electric
fieldstrength was found. It is important to have an accurate
calibration of both the spiral antennagain and the fiber optic link
loss as a function of frequency if precise electric field
measurementsare to be made.
The EM field detection system response for boresight radiation
at the three frequenciesinvestigated is shown in figure 2-24. In
this figure, the electrical output power of the optical
25
-
detector is plotted versus incident RMS electric field strength.
A 10-Hz spectrum analyzerresolution bandwidth was used in this
measurement. Extremely linear response is obtained fromthe highest
field levels down to the minimum detectable levels for each
frequency investigated.To assess the accuracy of these field
measurements, electric field values were obtained from thedetected
optical power levels, since the RF insertion loss of the link as
well as the gain of theantenna were known. The result of this
measurement is shown in Figure 2-25 where measuredfield strengths
are plotted versus calculated electric field strengths. There is
some deviationfrom expected field levels, which is attributed to
the anechoic chamber testing procedure. Thepointing accuracy for
the three relatively high gain horn antennas was not precisely
controlled,resulting in actual electric field levels that differed
from the calculated levels. Nonetheless,RMS electric field
sensitivities of approximately 15 IV/m, 44 VV/m, and 107 •IV/m have
beenachieved in a 1-Hz resolution bandwidth for the 2.25, 9.52, and
16-GHz frequencies, respec-tively. The RMS electric field detection
ranges for the three frequencies investigated aresummarized in
table 2.1. These experimentally attained electric field detection
ranges imply thatthis system can be useful for remotely performing
broadband, large dynamic range electric fieldmeasurements.
-6O
4 2.25 GHz 4+
* 9.52 GHz 4+-0 a 16.0GHz .4.
N• 4. oO -100 +.
44 o4+
S-120 +
o 4. .I�14 + 13
o 0 ql4r 4.Oa. +1
0 +
-160 • o4.D+W • • O•
-18 ... ! . .. " .. - .. • . ..- ' .. ......
-180
10"6 10-5 10- 10-3 10-2 10-1 100 101
CALCULATED ELECTRIC FIELD (V/m)
Figure 2-24. EM field detection system output versus incidentRMS
electric field level.
26
-
101101 - THEORY +
+2.25 GHZ +j
100 9.52GHz + +
a 18.OGz U
S10"1 413
10-2•4p
a .
10 ..4÷10-3
-', + +
-5
10
10-6 10 -4 10-3 10-2 10-1 10 0 10 1
CALCULATED ELECTRIC FIELD (V/m)
Figur' 2-z5. Measured RMS electric field levels versuscalculated
RMS electric field levels.
Table 2.1. Externally modulated EO EME monitoring systemelectric
field level detection ranges at 2.25, 9.52, and 16.0 GHz.
RMS Electric Field Detection Range
Frequency (1 Hz Bandwidth) (1 kHz Bandwidth)(GHz)
2.25 15 t&V/m - 0.8 V/m 0.02 V/m - 90 V/m9.52 44 •IV/m - 6
V/m 0.04 V/m - 550 V/m
16.0 107 IAV/m- 17 V/m 0.1 V/m - 1585 V/m
27
-
3.0 IMPROVEMENTS TO EO EME MONITORING SYSTEM
The Phase I testing of the breadboarded EO EME monitoring
systems proved quite valuablein assessing the technology and
determining areas of further development. In this Section,system
improvements and modifications which have been implemented as part
of the Phase H1efforts associated with this program will be
discussed. Performance improvements in the fiberoptic link and
wideband RF probes have been attained. One improvement is the
development ofstable, remote optically powered and controlled fiber
optic links, and another is remote polariza-tion control of fiber
optic links. Link performance improvements have been achieved due
to theinsertion of superior optical components. The current status
of the ultrawideband antennadevelopment efforts will be reviewed.
The fiber optic link improvements and modifications willbe
discussed first.
3.1 IMPROVEMENTS TO FIBER OPTIC LINK
The performance of analog photonic components has been steadily
improving year by year.Since the start of this R&D effort in FY
91, significant advances have been made in the areas ofsolid-state
laser and optical modulator technology, and these advances affect
the performance ofthe EQ EME monitoring system. In this section,
fiber optic analog link performance improve-ments and link
modifications during Phase II of this effort will be discussed.
The link performance of the "best available" 18-GHz fiber optic
link has been measured.This link is similar to that described in
Section 2.3, but has slightly superior optical componentsand is the
link to be used in the FY 94 shipboard demonstration under this
project. It consists ofa 50-mW Nd:YAG laser (Amoco Laser Company
Model ALC1320-50S), a 300-meter length ofPMF (3M Model FS-HB-6621)
for the uplink, a 300-meter length of SMF (Corning ModelSMF28) for
the downlink, an 18-GHz optical modulator (UTP Model APE
MZM-1.3-18-T-01),and an 18-GHz InGaAs p-i-n photodiode (Fermionics
Corporation Model HSD30). Postdetec-tion amplification with a
low-noise amplifier (LNA) is implemented to reduce the overall NF
ofthe receiver. The spectrum analyzer used in this work (Hewlett
Packard Model 8566B) has anNF of approximately 40 dB.
The fiber optic link transfer function (electrical in/electrical
out) is shown in figure 3-1. AnRF insertion loss of 35 dB is shown
at 18 GHz along with a 3-dBe link bandwidth of 15 GHz.This system
possesses an NF of less than 35 dB at 2 GHz as well as an SFDR of
>100 dB/Hz/.This link has been calibrated, packaged, and is
currently undergoing environmental testing,which continues into
Phase III of this effort.
To address the > 18-GHz system requirements, current R&D
efforts include a 40-GHzlithium niobate MZ modulator being
developed by Boeing, a 40-GHz MI-V semiconductor EAmodulator being
developed by Fermionics, and a 50-GHz M1-V semiconductor MZ
deviceprocured from GEC-Marconi Materials Technology. The status of
these R&D efforts will nowbe discussed.
3.1.1 Ultrawideband Optical Modulators
Efforts to further develop the semiconductor EA modulators for
shipboard analog photoniclink applications are continuing. A Navy
development program with Fermionics Corporation isdirectly
supporting the shipboard EME monitoring effort. The purpose of this
Office of Naval
28
-
>S21 log MAG3REF -30.0 dD 2.0 l-Iz1 3.0 dB/ -29.397 dOV
-29.397 dBhP
C-A IF AVERAGIG FACTOR
START 2.SSSOOSO 004HzSTOP 18.OOUOOS I0,.Hz
Figure 3-1. Fiber optic link transfer function.
Research-sponsored and NRaD-administered effort is to develop a
high-speed M-V semiconduc-tor EA modulator. The objective of the
work is to commercialize a 40-GHz InGaAsP/InP FKEmodulator operable
at a wavelength of 1.32 tM for digital as well as analog optical
linkapplications. The operation of the FKE modulator was discussed
in Section 2.1.2 of this report.At present, a >20-GHz
fiber-pigtailed modulator has been delivered to the Navy.[17.18]
Thisdevice suffers from a rather large optical insertion loss (8.8
dB) and only a moderate RFefficiency (10 dB with 7 V voltage
swing). Theoretical simulations and other experimental worksuggest
that these parameters can be significantly improved. Nevertheless,
this device is beingtested for shipboard use. Current work is
focused upon improving the modulator performanceparameters and
delivering a 40-GHz semiconductor EA modulator in FY 94.
Under direct NRaD sponsorship supporting the EO EME monitoring
task, Boeing isdeveloping a wideband traveling-wave lithium niobate
MZ modulator. The goal of this effort isto deliver an MZ modulator
which is suitable for operation at 1.3 ain, is operable from DC to
40GHz, has a V. of 20 dB, and a fiber-to-fiber opticalinsertion
loss of 20-GHz bandwidth optical modulatorsand it intends to
slightly modify the structure to efficiently achieve a >40-GHz
modulationbandwidth. A 40-GHz lithium niobate optical modulator
possessing the performance specifica-tions stated above is to be
delivered to the Navy for testing in FY 94.
A wideband HI-V semiconductor MZ optical modulator has also been
procured fromGEC-Marconi Materials Technolgy. According to
specifications, this device is to operate at 1.3Ium, be operable
from DC to 50 GHz, and have an RF V. = 10 V, a >20-dB extinction
ratio, andan optical insertion loss of
-
areas except the optical insertion loss. As this technology
advances, lower fiber-to-fiber opticalinsertion losses should be
attained. The GaAs-based modulator with a < 11-dB optical
insertionloss is to be delivered to the Navy for testing in FY
94.
The suitability of these three wideband optical modulator
approaches for shipboard EMEmonitoring will be experimentafiy
determined in FY 94 and recommendations will be forthcom-ing.
3.1.2 Optically Powered and Controlled Remote Fiber Optic
Links
Many fiber optic antenna remoting applications, including
shipboard EME monitoring,require standoff electrical powering of
the antenna-coupled components. Self-contained batterypacks are one
solution. However, batteries possess a finite lifetime that can
limit their usefulnessfor many applications. In some cases, the
optical powering can be accomplished using a PBLsystem, which has
the advantage that all-dielectric fiber optic cables can be used.
This can beessential for applications like the EME monitoring
system, where EMI effects are to be mini-mized. In addition, for
multioctave externally modulated fiber optic links, the electrical
biasingof the optical modulator is critical in minimizing nonlinear
distortion. Although accuratemethods of passively biasing the
modulators are being investigated,[ 19] the environmentalstability
of the bias position has not been established. Consequently, active
modulator biasing isalso under investigation. Combining remote
optical powering and remote optical modulator biascontrol has been
demonstrated.[2°,2 1] Here, we will demonstrate this technique with
twomultioctave fiber links and show how accurately the bias point
must be controlled in order for nodynamic range penalty to
result.
The two externally modulated fiber optic links investigated in
this work operated from 30 to500 MHz (proton exchange modulator)
and from 2 to 18 GHz (titanium diffused modulator).Each link
consists of a 1.32-pgm Nd:YAG laser, PMF for the uplink, a lithium
niobate MZmodulator, SMF for the downlink, and a high-speed
photodiode. The PBL system consists of ahigh-power AlGaAs laser
diode array, a multimode optical fiber (100 pm/140 pgm), and
ahigh-efficiency GaAs photocell. The AlGaAs laser diode is
intensity-modulated at low fre-quency (= 100 Hz) to introduce a
reference modulation onto the optical modulator. The secondharmonic
of this low-frequency modulation is then remotely detected and
analyzed using alock-in amplifier. The output of the lock-in
amplifier is used to actively control the opticalpower of the
AIGaAs laser diode, which in turn controls the optical modulator
bias position. Aschematic of this fiber optic system is shown in
figure 3-2. The performance requirements andthe ability of this
optical powering and biasing control technique to minimize
modulator-inducedharmonic distortion ov,_; wide temperature ranges
will now be discussed.
The PBL unit consists of a 250-mW AlGaAs laser diode array
(Spectra Diode Labs. ModelSDL-2432-P2) emitting at 810 nm which is
directly coupled into an optical fiber with a corediameter of 100
Rm (manufactured by Spectran Corp.). At the modulator side of the
link, theoptical power is converted to electrical power using a
12-V GaAs photocell (Photonic PowerSystems Model PPC-12S-ST).
Approximately 50 mW of optical power is delivered through
theoptical link to the photocell. A graph of the output voltage as
a function of series resistance foran input power of 50 mW is shown
in figure 3-3. A series resistance of 2.7 k,, is chosen whichallows
for an output voltage of between 0 and 12 V to be obtained. A plot
of the output voltageas a function of input optical power is shown
in figure 3-4. Very little current is required(
-
104*OSCILLATORWVITH PC OFFSET
CPU
LOCK-ORAMPLIFIER
DC SECTUMT ESRHEDASU
ANALYZER
IIF
PHOTODI0ODE
0.11.0•m MM
100 a
OPTIAL RIO
pup2-N CANXE (100 soN4:YVAG LAWNR
I-I010a
RESITANCE(KL O-HS
r,2. Schematic of orti I powering system and modulator bias
control circuitry.
143
12"
10"
4-
2-
0 1'0 ... 300' O 40RESISTANCE (KILO-OHMS)
Figure 3-3. Photocell output electrical voltage as a function of
seriesresistance for 50-mW input optical power.
31
-
12
10
w
4
2
0 25 50 75 100 125 150 175 200 225 250OPTICAL POWER (mW)
Figure 3-4. Photocell output voltage versus input optical power
using a2.7-kQ series resistance.
(voltage X current) is required for this remote unit. A 0.47-piF
capacitor is placed in series withthe 2.7-kQ resistor to damp out
voltage fluctuations caused by transmitted optical
powerfluctuations through the multimode fiber. The transmitted
optical power is modulated with a100-Hz small modulation depth
signal for the modulator bias control. This optical
poweringtechnique allows for the modulator voltage to easily be
changed simply by varying the trans-mitted optical power.
The requirement for the fiber optic link is that the SFDR is not
reduced by any error inbiasing the modulator. The reciever signal
powers for the fundamental through the third-orderintermodulation
product for the MZ link excited with equal intensity RF tones at w,
and w2 aregiven by
102 sin2(8)jo2(m)j1 2(m) fundamental; wj,wo2 (3.1)
1o2 cOS2(8)Jo2 (m)j 22 (m) 2nd harmonic; 2wl,2wo2 (3.2)
Io2cos2 (8)j64(m) 2nd-order intermodulation; w1* o (3.3)
o02sin 2(6)Jj1 2(M)j22(m) 3rd-order intermodulation; (3.4)2wj1-±
o)2, 2wo2 ±- a),
where Io is the optical power incident on the detector, 8 is the
modulator bias position, m is themodulation index, and Jn(m) is the
Bessel function of the first kind of order n. The modulationindex
is given by m = (7V2)(Vd/Vx), where Vd is the instantaneous drive
voltage at wl and wo2,
32
-
and V., is the modulator half-wave voltage. For links that have
bandwidths less than one octave,the SFDR is measured using the
third-order intermodulation signals. The relative levels of
thefundamental signal to the spur at the third-order
intermodulation frequencies can be computedby taking the ratio of
Eq. 3.4 to Eq. 3.1 given above. The ratio is constant for a
specificmodulation index m and is independent of modulator bias 6.
However, if the link is designed formultioctave bandwidth
operation, the second-order intermodulaton signals must be
inspected.The second-order intermodulation signals, ow ± 0)2, are
larger than the second-order harmonics.The ratio of the fundamental
to the second-order intermodulation signals is a function of
themodulator bias point. Figure 3-5 shows the relative levels of
Eq. 3.1 to 3.4 as a function ofphase bias error for a modulation
index set to a value when the third-order products equal thenoise
floor of the system shown in figure 3-2. The maximum allowable bias
error is thencalculated based on the requirement that the SFDR not
be degraded by the modulator biasing.
0
m -20 fl
cr -40LU f- +/- f2•0 . ---2 -- -- - -- - ---- 2.............--
---
-120 2ff 2f-800 -100 2f..-.1-.-.
) -120 2f2 +-f 1"
cc -140
a.0 10 20 30 40 50 60 70 80 90
PHASE ERROR (deg)
Figure 3-5. Fundamental and spurious signal levels as a function
of phase bias errorfor the minimum detectable modulation index.
The receiver bandwidth is an important factor in finding the
sensitivity of the phase biaserror of the MZ modulators. For
instance, if the receiver bandwidth is reduced, the system
noisefloor is reduced and the spurious signals become more
prevalent. In figure 3-5, the noise floorwould move lower and the
second and third-order products would appear at a lower input
drivelevel (smaller modulation index). Therefore, the output
absolute noise floor level affects themeasurable spurious signal
levels allowed, which determines the required modulator bias
pointaccuracy.
33
-
The 30- to 500-MHz fiber optic link used for this measurement
has an RF insertion loss of11 dB, an NF of 20 dB, and an SFDR of
109 dB/Hz23. The 2- to 18-GHz link has an RFinsertion loss of 27
dB, an NF of 35 dB, and a 108-dB/HzY SFDR for optimum
modulatorbiasing. The effect of a 1-degree phase bias error on the
SFDR for the 30- to 500-MHz link isshown in figure 3-6. The plot
assumes a 30-kHz receiver bandwidth, which is appropriate
forchannelized receiver or ultrawideband applications. It is
already apparent that the SFDR of thislink is being severely
limited by the harmonic levels. A similiar result is found for the
2- to18-GHz link. To emphasize this point further, a plot of the
phase error tolerance (the second-order intermodulation signal
level equals the third-order intermodulation level) versus
receiverbandwidth for the lower frequency link is shown in figure
3-7. This curve suggests that formultioctave channelized receiver
applications, a modulator phase bias error of 0.5 degree or lessis
required. This level of modulator phase bias accuracy is extremely
difficult to attain usingpassive modulator biasing techniques.
0 RF INSERTION LOSS = 11.6 dBE-20 V•t=4VE -20 -SFDR =80 dB1
-40 BANDWIDTH =30 kHzrr -40
' -60a -80 f2/o /-
W -100O -120 NOISE FLOOR Io -12,0 ,
"W /I-14 /U 140/ 2fj +/- f2 ,
-160 2f2 fl
-120 -100 -80 -60 -40 -20 0 20POWER APPLIED TO MODULATOR
(dBm)
Figure 3-6. Effect of 1-degree modulator phase bias error on the
30-500-MHz fiber optic link.
A plot of the modulator phase bias drift versus temperature for
the modulators used in the30- to 500-MI-Iz and 2- to 18-GHz links
is shown in figure 3-8. Both the proton-exchanged andthe
ion-diffused modulators show unacceptable bias point drifts over
the 200 C to 1000 Ctemperature range. Even the most stable
modulator designs will have difficulty maintaining thebias point to
the degree required. Hence, active control of the modulator bias
point is imperativefor high-performance multioctave link
applications. Computer-controlled bias point stabilizationresults
between 200 C and 100°C are shown in figure 3-9 for both links
investigated. A secondharmonic suppression as high as 20 dB was
obtained for both links, which translates intoincreased SFDR. Using
the computer control, the bias points for both links were
maintained to
34
-
MODULATION LEVEL (dBm)-35 -30 -25 -20 -15 -10 -5 0
2.5
2-J20
1.5 a12 R
cc
0 .5 Cl)
0~a10 Hz 10 kHz 10 MHz 10 GHz
RECEIVER BANDWIDTH (Hz)
Figure 3-7. Modulator phase bias tolerance as a function of
receiverbandwidth for the 500-MHz fiber optic link.
60.0
40.0
"R 20.00 ION DIFFUSED,
0. 18 GHz BANDWIDTH
0 -20.0LU n PROTON/ -40.0 O< EXCHANGED,n -60.0 500 MHz
BANDWIDTH
-80.0
-100.0 I :oi l !If) UO U) LO U) U*) U) LI)N C .J 113 n (D r t-
co
TEMPERATURE (-C)
Figure 3-8. Modulator phase bias drift as a function of
temperature forproton-exchanged and titanium-indiffused lithium
niobate optical modulators.
35
-
-30.00
-35.00
9 -40.00
Z -4500OPEN LOOP0_ -45.00 /u)
o3-50.00
a.. - COMPUTERCONTROLLED
-60.00
-65.00 ,
-70.00
N I) U) 0 UO 0 tI 0 Ln 0 LnNY C Y V) IV "W n kn cc(0
NTEMPERATURE ('C)
(a) Ion-diffused modulator.
-30.00
-35.00
.-40.00Z0 - OPEN LOOPen -45.00Enw( -50.00D 0 COMPUTERU) .55.00
CONTROLLED
-60.00
-65.00 J ', 2 , ' , ' , : ' ' '" 'n 8 ) 0 L LO O o Lo Otno
TEMPERATURE ('C)
(b) Proton-exchanged modulator.
Figure 3-9. Computer-controlled modulator phase bias
stabilization results as a functionof temperature for both low- and
high-frequency fiber optic links.
36
-
within I degree, which was limited by the time consant of the
feedback circuitry. Better biaspoint control should be possible
using this technique along with optimized feedback algorithms.This
should allow multioctave fiber optic links employing MZ optical
modulators to approachthird-order intermodulation-limited SFDRs.
This system improvement should enable EO EMEmonitoring systems to
operate over the wide temperature ranges expected in the
shipboardenvironment.
3.1.3 Remote Optical Polarization Control of EM Field Sensor
The performance of externally modulated remote fiber optic links
similiar to that shown infigure 2-1 is subject to polarization
fading if the proper polarization is not presented to themodulator.
The use of PMF is a common method for overcoming this problem, and
due to itsrelative maturity, it is the method to be used in the FY
94 shipboard demonstration for thisproject. However, PMF is
difficult to use, is prone to damage, and can add significant cost
to thesystem. Alternatively, two techniques have been investigated
which eliminate the need for PMFin remote link applications. One
technique uses a quasidepolarized optical source by combiningtwo
orthogonally polarized light beams in a single-mode optical
fiber.[22] In this case, themodulator ideally accepts the same
amount of optical power irrespective of fiber polarizationeffects.
With this technique, it is critical that the power of the two
lasers is identical, that thepolarizations are truly orthogonal,
and that the laser beat frequency is pushed beyond thefrequency
range of interest. These are not trivial experimental tasks.
Another technique relieson active polarization control with
standard SMF to compensate for polarization drifts in thefiber
between the source and the controller and in the fiber leading to
the sensor. With thistechnique, much lower cost fiber can be used,
which would permit simple field repairs and theuse of optic.i
sources that are pigtailed with standard fiber rather than PMF.
This polarization-control technique has been investigated for its
application to the shipboard EME monitoringsystem.
Several polarization-controlling systems have been developed
based on either lithiumniobate, rotating waveplates, fiber
squeezers, or fiber cranks.[231 However, these systems areeither
expensive, have high insertion loss, or mechanically fatigue the
fiber, which can some-times result in breakage. Liquid crystals can
be used for polarization control and do not sufferfrom these
limitations.[24-] While response times of liquid crystals were once
considered to betoo slow for some applications,[12 methods are
being investigated to improve the response timeuntil it is
practical for virtually all communication and sensor applications.
This Sectiondescribes work demonstrating the use of a
liquid-crystal polarization controller in a remote fiberoptic link
configuration similiar to that used in the EO EME monitoring
system.
The experimental setup is shown in figure 3-10. The source is a
10-mW Nd:YAG solid-statelaser emitting at 1.32 gm, followed by a
quarter-wave plate and a half-wave plate. The wave-plates are
mounted on stages which can be rotated to produce any desired
polarization andeffectively simulate the effect of birefringence in
standard SMF between the source and thepolarization controller. The
liquid-crystal polarization controller consists