Project 1

Wide Input Wide Output Dc-Dc Converter

Abstract

WIWO(wide input wide output) presents a new wide-input–wide-output dc–dc

converter, which is an integration of buck and boost converters via a tapped inductor,

Coherent transition between step-down and step-up modes is achieved by a proper

control scheme that is by applying proper control to the two active switches, the

converter exhibits both buck and boost features]. This paper presents theoretical

concepts and experimental results.

EEE DEPT. HKBKCE 2012 Page 1


Chapter 1.Introduction

The BUCK, boost, buck–boost, and Cu´ k converters are the four basic dc–dc

non-isolating converters that have found wide applications in industry. The buck

converter can step down the dc voltage, whereas the boost converter is capable to

perform a step-up function. In applications where both step-up and step-down

conversion ratios are required, the buck–boost and Cu´k converters can be used.

Simplicity and robustness are among the advantages of the buck–boost converter.

However, the pulsating input and output currents cause high conduction losses, and

thus, impair the efficiency of buck–boost. Furthermore, the buck–boost converter

uses the inductor to store the energy from the input source, and then, release the

stored energy to the output. For this reason, the magnetic components of buck– boost

are subjected to a significant stress. These disadvantages limit the applications of the

buck–boost converter mainly to low power level. The isolated version of buck–boost,

referred to aas the flyback converter, can achieve greater step-up or step-down

conversion ratio utilizing a transformer, possibly, with multiple outputs. As compared

with the buck–boost converter, the Cu´k converter has higher efficiency and smaller

ripples in input and output currents.

A significant improvement of the Cu´k converter performance can be achieved

by applying the zero ripple concept. The Cu´k converter can be found in many high-

performance power applications. In theory buck and boost converters can generate

almost any voltage, in practice, the output voltage range is limited by component

stresses that increase at the extreme duty cycle. Consequently, buck converter losses

mount at low duty cycle, whereas boost converter efficiency deteriorates when the

duty cycle tends to unity. Accordingly, voltage conversion range of the buck

converter below 0.1–0.15 becomes impractical whereas that of the boost converters’

is limited to below 8–10. Additional problems associated with narrow duty cycle are

caused by MOSFET drivers rise and fall times as well as pulse width-modulated

(PWM) controllers that have maximum pulse width limitations. These problems

become even more severe at higher voltages and higher frequencies.

Introducing a transformer helps attaining large step-up or step-down voltage

conversion ratio. Transformers turn ratio should be chosen as to provide the



desired voltage gain while keeping the duty cycle within a reasonable range for

higher efficiency. The transformer, however, brings in a whole new set of

problems associated with the magnetizing and leakage inductances, which cause

voltage spikes and ringing, increased core and cooper losses as well as increased

volume and cost.

In a quest for converters with wide conversion range, quite a few authors

proposed using converters with nonlinear characteristics. Single-transistor

converter topologies, with quadratic conversion ratios, were proposed in [1] and

demonstrated large step-down conversion ratio. This method has successfully

achieved wide conversion range in the step down direction. A different approach

to obtain wide conversion range utilizing coupled inductors was proposed in [2].

With only minor modification of the tapped-inductor buck, [2] shows low

component count and solves the gate-drive problem by exchanging the position of

the second winding and the top switch. The problem of a high turn-OFF voltage

spike on the top switch was solved by applying a lossless clamp circuit. Due to the

coupled inductor action, the converter demonstrated high step-down dc–dc

conversion ratio, whereas the converter’s efficiency was improved by the

extended duty cycle. A tapped-inductor buck with soft switching was introduced

in [3]

.

3.Tapped-Inductor Buck

Derivations of the tapped-inductor buck were also suggested in [4] and [5].

An- other modification of the tapped-buck converter was realized in [6] for power

factor correction (PFC) application. With the addition of a line-frequency

commutated switch and a diode, both flyback and buck characteristics were

achieved and large step-down was demonstrated.



Tapped-Inductor Boost Converters

Some applications, especially battery-operated equipment, require high

voltage boosting. To attain very large voltage step- up, cascaded boost converters that

implement the output voltage increasing in geometric progression were introduced in

[7]. These converters effectively enhance the voltage transfer ratio; however, their

circuits are quite complex. In comparison, tapped-inductor boost converters proposed

in [8] and [9] attain a comparable voltage step-up preserving relative circuit

simplicity.In [10], the boost converter output terminal and flyback converter output

terminal are connected in series to increase the output voltage gain with the coupled

inductor. The boost converter also functions as an active clamp circuit to recycle the

snubber energy.



CHAPTER 2.LITERATURE SURVEY

Brief Survey of Converters

There are two types of converters: Non-Isolated Converters and Isolated

Converters.

2.1 Non-isolated converters

The non-isolating type of converter is generally used where the voltage needs

to be stepped up or down by a relatively small ratio (say less than 4:1), and there is no

problem with the output and input having no dielectric isolation. Examples are

24V/12V voltage reducers, 5V/3V reducers and 1.5V/5V step-up converters. There

are five main types of converter in this non-isolating group.

i) Buck converters

ii) Boost converters

iii) Buck-boost converters

iv) Cuk converters

v) Charge-Pump converter

The buck converter is used for voltage step-down/reduction, while the boost

converter is used for voltage step-up. The buck-boost and Cuk converters can be used

for either step-down or step-up, but are essentially voltage polarity reversers or

‘inverters’ as well. (The Cuk converter is named after its originator, Slobodan Cuk of

Cal Tech university in California.) The charge-pump converter is used for either

voltage step-up or voltage inversion, but only in relatively low power applications.

2.1.1 Buck converter:

The basic circuit configuration used in the buck converter is shown in Fig.1.

As you can see there are only four main components: switching power MOSFET Q1,

flywheel diode D1, inductor L and output filter capacitor C1. A control circuit (often a

single IC) monitors the output voltage, and maintains it at the desired level by

switching Q1 on and off at a fixed rate (the converter’s operating frequency), but with

a varying duty cycle.



Fig.2.1.1: The basic circuit for a Buck type of DC-DC converter

When Q1 is turned on, current begins flowing from the input source through

Q1 and L, and then into C1 and the load. The magnetic field in L therefore builds up,

storing energy in the inductor - with the voltage drop across L opposing or ‘bucking’

part of the input voltage. Then when Q1 is turned off, the inductor opposes any drop

in current by suddenly reversing its EMF, and now supplies current to the load itself

via D1.

The DC output voltage which appears across the load is a fraction of the input

voltage, and this fraction turns out to be equal to the duty cycle. So we can write:

Vout/Vin = D,

or Vout = Vin x D

where D is the duty cycle, and equal to Ton/T, where T is the inverse of the

operating frequency.

So by varying the switching duty cycle, the buck Converter’s output voltage

can be varied as a fraction of the input voltage. A duty cycle of 50% gives a step

down ratio of 2:1, for example, as needed for a 24/12V step-down converter. The

current ratio between output and input will be the reciprocal of the voltage ratio;

ignoring losses for a moment, and assuming our converter is perfectly efficient. So

Iout/In = Vin/Vout



So when we are stepping down the voltage by 2:1, the input current is only

half the value of the output current. Or it would be, if it were not for the converter’s

losses. Because real-world converters aren’t perfect the input current is typically at

least 10% higher than this.

2.1.2 Boost converter:

The basic boost converter is no more complicated than the buck converter, but

has the components arranged differently (Fig.2.1.2) in order to step up the voltage.

Again the operation consists of using Q1 as a high speed switch, with output voltage

control by varying the switching duty cycle. When Q1 is switched on, current flows

from the input source through L and Q1, and energy is stored in the inductor’s

magnetic field. There is no current through D1, and the load current is supplied by the

charge in C1. Then when Q1 is turned off, L opposes any drop in current by

immediately reversing its EMF - so that the inductor voltage adds to (i.e., ‘boosts’)

the source voltage, and current due to this boosted voltage now flows from the source

through L, D1 and the load, recharging C1 as well.

a

b



Fig.2.1.2: A non-ideal boost converter: (a) schematic, (b) inductor voltage and

capacitor current waveforms.

The output voltage is therefore higher than the input voltage, and it turns out that the

voltage step-up ratio is equal to:

Vout/Vin = 1/(1-D)

where 1-D is actually the proportion of the switching cycle that Q1 is off, rather than

on. So the step-up ratio is also equal to:

Vout/Vin = T/Toff

Again, if we assume that the converter is 100% efficient the ratio of output

current to input current is just the reciprocal of the voltage ratio:

Iin/Iout = Vout/Vin

So if we step up the voltage by a factor of 2, the input current will be twice the

output current. Of course in a real converter with losses, it will be higher

2.1.3 Buck-boost converter

The main components in a buck-boost converter are again much the same as

in the buck and boost types, but they are configured in a different way (Fig.2.1.3).

Fig.2.1.3: The Buck-Boost converter.

This allows the voltage to be stepped either up or down, depending on the

duty cycle. Here when MOSFET Q1 is turned on, inductor L is again connected

directly across the source voltage and current flows through it, storing energy in the

magnetic field. No current can flow through D1 to the load, because this time the

diode is connected so that it is reverse biased. Capacitor C1 must supply the load

current in this ‘Ton’ phase. But when Q1 is turned off, L is disconnected from the

source. Needless to say L again opposes any tendency for the current to drop, and

instantly reverses it’s EMF. This generates a voltage which forward biases D1, and



current flows into the load and to recharge C1. With this configuration the ratio

between the output and input voltages turns out to be:

Vout/Vin = -D/(1-D)

which again equates to

Vout/Vin = -Ton/Toff

So the buck-boost converter steps the voltage down when the duty cycle is

less than 50% (i.e., Ton < Toff), and steps it up when the duty cycle is greater than

50% (Ton > Toff). But the output voltage is always reversed in polarity with respect

to the input . so the buck-boost converter is also a voltage inverter.

When the duty cycle is exactly 50%, for example, Vout is essentially the same

as Vin, except with the opposite polarity. So even when it’s not being used to step the

voltage up or down, the buck-boost converter may be used to generate a negative

voltage rail in equipment operating from a single battery. As before, the ratio between

output and input currents is simply the reciprocal of the voltage ratio, if we ignore

losses.

2.1.4 CUK CONVERTER:

The basic circuit of a Cuk converter is shown in Fig.2.1.4, it has an additional

inductor and capacitor. The circuit configuration is in some ways like a combination

of the buck and boost converters, although like the buck-boost circuit it delivers an

inverted output. Virtually all of the output current must pass through C1, and as ripple

current so C1 is usually a large electrolytic with a high ripple current rating and low

ESR (equivalent series resistance), to minimize losses.

L1 C1 L2

Fig.2.1.4: The Cuk converter

When Q1 is turned on, current flows from the input source through L1 and Q1,

storing energy in L1’s magnetic field. Then when Q1 is turned off, the voltage across

L1 reverses to maintain current flow. As in the boost converter current then flows

from the input source, through L1 and D1, charging up C1 to a voltage somewhat



higher than Vin and transferring to it some of the energy that was stored in L 1. Then

when Q1 is turned on again, C1 discharges through via L2 into the load, with L2 and C2

acting as a smoothing filter. Meanwhile energy is being stored again in L1, ready for

the next cycle. As with the buck-boost converter, the ratio between the output voltage

and the input voltage again turns out to be:

Vout/Vin = -D/(1-D)

= -Ton/Toff

where the minus sign again indicates voltage inversion. So like the buck-boost

converter, the Cuk converter can step the voltage either up or down, depending on the

switching duty cycle. The main difference between the two is that because of the

series inductors at both input and output, the Cuk converter has much lower current

ripple in both circuits. In fact by careful adjustment of the inductor values, the ripple

in either input or output can be nulled completely.

2.1.5 Charge-pump converter

All of the converters we’ve looked at so far have depended for their

operation on storing energy in the magnetic field of an inductor. However there’s

another type of converter which operates by storing energy as electric charge in

a capacitor, instead. Converters of this type are usually called charge-pump

converters, and they’re a development from traditional voltage doubling and

‘voltage multiplying’ rectifier circuits.

The basic circuit for a voltage doubling charge-pump converter is shown

in Fig.2.1.5, and as you can see, it mainly uses four MOSFET switches and a

capacitor C1 — usually called the ‘charge bucket’ capacitor.

Operation is fairly simple. First Q1 and Q4 are turned on, connecting C1

across the input source and allowing it to charge to Vin. Then these switches

are turned off, and Q2 and Q3 are turned on instead. C1 is now connected in

series with the input voltage source, across output reservoir capacitor C2. As a

result some of the charge in C1 is transferred to C2, which charges to twice

the input voltage. This cycle is repeated at a fairly high frequency, with C2

providing the load current during the part of the cycle when Q2 and Q3 are

turned off.

As you can see all of the energy supplied to the load in this type of

converter flows through C1, and as ripple current. So again this capacitor needs



to have a relatively high value, have low ESR (to minimise losses) and be able to

cope with a heavy ripple current.

A slightly different circuit configuration from that shown in Fig.2.1.5 can

be used to deliver an inverted voltage of the same value as Vin, instead of a

doubled voltage. This type of converter finds use in generating a negative supply

rail for electronic circuits running from a single battery.

On the whole, though, the fact that charge-pump converters rely for their

operation on charge stored in a capacitor tends to limit them to relatively low

current applications. However for this type of operation they’re often cheaper

and more compact than inductor-type converters.

Fig 2.1.5: A basic Charge-Pump converter which doubles the input voltage.

2.2 ISOLATED CONVERTERS:



All of the converters above have virtually no electrical isolation between the

input and output circuits; in fact they share a common connection. This is fine for

many applications, but it can make these converters quite unsuitable for other

applications where the output needs to be completely isolated from the input. Here is

where a different type of inverter tends to be used - the isolating type. There are two

main types of isolating inverter in common use: the ‘flyback’ type and the ‘forward’

type. Like most of the non-isolating converters, both types depend for their operation

on energy stored in the magnetic field of an inductor; or in this case, a transformer.

2.2.1 Flyback converter:

The basic circuit for a flyback type converter is shown in Fig.2.2.1. In many

ways it operates like the buck-boost converter of Fig.2.1.3, but using a transformer to

store the energy instead of a single inductor.

Fig.2.2.1: The Flyback converter

When MOSFET Q1 is switched on, current flows from the source through

primary winding L1 and energy is stored in the transformer’s magnetic field. Then

when Q1 is turned off, the transformer tries to maintain the current flow through L1 by

suddenly reversing the voltage across it, generating a ‘flyback’ pulse of back-EMF.

Q1 is chosen to have a very high breakdown voltage, though, so current simply can’t

be maintained in the primary circuit. But because of transformer action an even

higher flyback pulse is induced in secondary winding L2.



And here diode D1 is able to conduct during the pulse, delivering current to

the load and recharging filter capacitor C1 (which provides load current between

pulses). So as you can see, the flyback converter again has two distinct phases in its

switching cycle. During the first phase Q1 conducts and energy is stored in the

transformer core via the primary winding L1. Then in the second phase when Q1 is

turned off, the stored energy is transferred into the load and C1 via secondary winding

L2. The ratio between output and input voltage of a flyback converter is not simply a

matter of the turns ratio between L2 and L1, because the back-EMF voltage in both

windings is determined by the amount of energy stored in the magnetic field, and

hence depends on the winding inductance, the length of time that Q1 is turned on, etc.

However the ratio between L2 and L1 certainly plays an important role, and most

flyback converters have a fairly high turns ratio to allow a high voltage step-up ratio.

Because of the way the flyback converter works, the magnetic flux in its

transformer core never reverses in polarity. As a result the core needs to be fairly

large for a given power level, to avoid magnetic saturation. Because of this flyback

converters tend to be used for relatively low power applications, like generating high

voltages for insulation testers, Geiger counter tubes, cathode ray tubes and similar

devices drawing relatively low current. Although it’s not shown in Fig.2.2.1, a third

small winding can be added to the flyback transformer to allow sensing of the flyback

pulse amplitude (which is reasonably close to the output voltage Vout). This voltage

can be then fed back to the MOSFET switching control circuit, to allow it to

automatically adjust the switching to regulate the output voltage.

2.2.2 Forward converter

In contrast with the flyback converter, where there are two distinct phases for

energy storage and delivery to the output, the forward converter uses the transformer

in a more traditional manner, to transfer the energy directly between input and output

in the one step. The most common type of forward converter is the push-pull type,

and the basic circuit for this type is shown in Fig.2.2.2.1.



Fig.2.2.2.1: The basic circuit for a Forward converter

Forward converter is another popular switched mode power supply (SMPS)

circuit that is used for producing isolated and controlled dc voltage from the

unregulated dc input supply. As in the case of fly-back converter the input dc supply

is often derived after rectifying (and little filtering) of the utility ac voltage. The

forward converter, when compared with the fly-back circuit, is generally more energy

efficient and is used for applications requiring little higher power output (in the range

of 100 watts to 200 watts). However the circuit topology, especially the output

filtering circuit is not as simple as in the fly-back converter.

Fig. shows the basic topology of the forward converter. It consists of a fast

switching device ‘S’ along with its control circuitry, a transformer with its primary

winding connected in series with switch ‘S’ to the input supply and a rectification and

filtering circuit for the transformer secondary winding. The load is connected across

the rectified output of the transformer-secondary.

The transformer used in the forward converter is desired to be an ideal

transformer with no leakage fluxes, zero magnetizing current and no losses. The basic

operation of the circuit is explained here assuming ideal circuit elements and later the

non-ideal characteristics of the devices are taken care of by suitable modification in

the circuit design. In fact, due to the presence of finite magnetizing current in a



practical transformer, a tertiary winding needs to be introduced in the transformer and

the circuit topology changes slightly.

2.3 Resonant Converters

Resonant converters use a resonant circuit for switching the transistors when

they are at the zero current or zero voltage point; this reduces the stress on the

switching transistors and the radio interference. We distinguish between ZVS- and

ZCS-resonant converters (ZVS: Zero Voltage Switching, ZCS: Zero Current

Switching). To control the output voltage, resonant converters are driven with

constant pulse duration at a variable frequency. The pulse duration is required to be

equal to half of the resonant period time for switching at the zero-crossing points of

current or voltage. There are many different types of resonant converters. For

example the resonant circuit can be placed at the primary or secondary side of the

transformer. Another alternative is that a serial r parallel resonant circuit can be used,

depending on whether it is required to turn off the transistor, when the current is zero

or the voltage is zero.

Future renewable energy systems will need to interface several energy sources

such as fuel cells, photovoltaic (PV) array with the load along with battery backup. A

three-port converter finds applications in such systems since it has advantages of

reduced conversion stages, high-frequency ac-link, multiwinding transformer in a

single core and centralized control. This has been described [1]. Some of the

applications are in fuel-cell systems, automobiles, and stand-alone self-sufficient

residential buildings.

A three-port bidirectional converter has been proposed in [2] for a fuel-cell

and battery system to improve its transient response and also ensure constant power

output from fuel-cell source. The circuit uses phase-shift control of three active

bridges connected through a three-winding transformer and a network of inductors.



Fig. 2.3.1. System overview: a power electronic converter regulates the energy flow

between the fuel cell generator, an energy storage device, and the load.

To extend the soft-switching operation range in case of port voltage

variations, duty-ratio control is added in [3]. Another method to solve port voltage

variations is to use a front-end boost converter, as suggested in [3] for ultra-capacitor

applications. This topology comprises a high-frequency three-winding transformer

and three half-bridges, one of which is a boost half-bridge interfacing a power port

with a wide operating voltage. The three half-bridges are coupled by the transformer,

thereby providing galvanic isolation for all the power ports. The converter is

controlled by phase shift, which achieves the primary power flow control, in

combination with pulse width modulation (PWM).

Fig. 2.3.2. Three-port energy management system



Because of the particular structure of the boost half-bridge, voltage variations

at the port can be compensated for by operating the boost half-bridge, together with

the other two half-bridges, at an appropriate duty cycle to keep a constant voltage

across the half-bridge. The resulting waveforms applied to the transformer windings

are asymmetrical due to the automatic volt-seconds balancing of the half-bridges.

With the PWM control it is possible to reduce the rms loss and to extend the zero-

voltage switching operating range to the entire phase shift region.

To increase the power-handling capacity of the converter, three-phase version

of the converter is proposed in [5]. A high-power converter to interface batteries and

ultracapacitors to a high voltage dc bus has been demonstrated in [6] using half

bridges, a battery and an ultracapacitor. The converter consists of three half-bridges

and a high-frequency multi-winding transformer as shown below.

Fig:2.3.3. multiple-input ZVS bi-directional dc–dc converter.



Bi-directional power flow between input and output is achieved by adjusting

the phase-shift angles of the voltages across the two sides of the transformer. Soft-

switching is implemented naturally by snubber capacitors and transformer leakage

inductance.

Since the power flow between ports is inversely proportional to the impedance

offered by the leakage inductance and the external inductance, impedance has to be

low at high power levels. To get realizable inductance values equal to or more than

the leakage inductance of the transformer, the switching frequency has to be reduced.

Hence, the selection of switching frequency is not independent of the value of

inductance. A series-resonant converter has more freedom in choosing realizable

inductance values and the switching frequency, independent of each other. Such a

converter can operate at higher switching frequencies for medium and high-power

converters.

A three-port series resonant converter operating at constant switching

frequency and retaining all the advantages of a three-port structure is proposed in [1].

Other circuit topologies [7]–[12] are suggested in the literature for a three-port

converter such as the current-fed topologies [11] that have more number of magnetic

components and flyback converter topologies [12] that are not bidirectional. A

constant-frequency phase-controlled parallel-resonant converter was proposed in [13],

which uses phase shift between input bridges to control the ac-link bus voltage, and

also between input and output bridge to control the output dc voltage. Such high-

frequency ac-link systems using resonant converters have been extensively explored

for space applications and telecommunications applications. The series-resonant

three-port converter proposed in this paper uses a similar phase shift control but

between two different sources. The phase shifts can be both positive and negative,

and are extended to all bridges, including the load-side bridge along with

bidirectional power flow. A resonant converter topology is suggested in [16] but

phase-shift control is not utilized for control of power flow; instead, the converters

are operated separately based on the power flow direction.



2.4 ENERGY RECOVERY SNUBBER

Snubbers are an essential part of power electronics. Snubbers are small

networks of parts in the switching circuit whose function is to control the effects of

circuit reactances.

Snubbers enhance the performance of the switching circuits and result in

higher reliability, higher efficiency, higher switching frequency, smaller size, lower

weight and lower EMI. The basic intent of a snubber is to absorb energy from the

reactive elements in the circuit. The benefits of this may include circuit damping,

controlling the rate of change of voltage or current or clamping voltage over shoot.

In performing these functions a snubber limits the amount of stress which the

switch must endure and this increases the reliability of the switch. When a snubber is

properly designed and implemented the switch will have lower average power

dissipation much lower peak power dissipation, lower peak operating voltage and

lower peak operating current.

Since the WIWO operates a coupled inductor, the energy stored in the

leakage inductances becomes a problem to deal with. Besides increased switching

losses, discharge of the leakage inductance energy causes oscillations and increased

voltage spikes across the switches. The resulting voltage stress becomes intolerable

at higher voltages and higher power. If not snubbed, overvoltage breakdown of the

MOSFET devices may occur.

The proposed lossless snubber is comprised of a snubber capacitor CS and a

pair of fast diodes DS1 and DS2 . The snubber is fitted to WIWO, as shown in Fig

5.7.5. The snubber is effective both in buck mode and in boost mode.

Detailed description of the snubber operation is out of scope of this paper; in

brief, however, the operation is as follows. With WIWO in the buck mode, at the

instant when the S2 switch is turned off, the snubber diode DS1 conducts L1 leakage

current and allows the stored energy to be discharged into the snubber capacitance

CS and to the output of the circuit. This takes one- half resonant cycle dictated by

the leakage inductance and the snubber capacitance. CS will remain charged until

the S2 switch is turned ON again at the onset of the subsequent switching cycle. With

S2 turned ON, the energy stored in the snubber capacitor is discharged into L2

winding via DS2 and recycled.

In the boost mode, the snubber operation is similar. However, here, S1



interrupts the current and is subject to the voltage spike while S2 switch is constantly

ON with zero voltage VD S2 across.

Fig 2.4 Energy recovering snubber for WIWO power stage.


DC Input

DC-DC Converter

MOSFET Driver/ Pulse

Driver

PIC / PWM Controller

Supply

Load


Chapter3. BLOCK DIAGRAM

3.1 Motivation in the search for new switching topology

Fig 3.1.0 Block diagram

Fig 3.1.1 Buck-derived converters with tapped inductors



Fig 3.1.2 Voltage conversion ratio of buck-derived converters with tapped

inductors. (a) 0 < n < ∞. (b) 0 < n < ∞. (c) n > 1. (d) n > 1.

Fig 3.1.3 Boost-derived converters with tapped

inductors.



Fig 3.1.4 Voltage conversion ratio of boost-derived converters with tapped

inductors. (a) 0 < n < ∞. (b) 0 < n < ∞. (c) 0 < n < 1. (d) 0 < n < 1.

The basic buck and boost converters can be transformed into a number of new

topologies by bringing in the tapped inductor. The proposed tapped-inductor buck-

derived converters are shown in Fig 3.1.1, with their corresponding voltage

conversion ratios plotted in Fig 3.1.2. The proposed tapped-inductor boost- derived

topologies and their corresponding voltage conversion ratios are given in Figs. 3.1.3

and 3.1.4. Here, D is the duty ratio of switch S, M is the voltage conversion ratio,

and n is the turn ratio of the tapped inductors, which is defined as n = n2 : n1 . As

the turn ratio n tends to infinity, the conversion ratio of the buck-derived converters

approach the characteristic of a basic buck topology. On the other hand, as the turn

ratio n goes to zero, the conversion ratio of the boost-derived converters approach the

characteristic of a basic boost topology. Inspection of the conversion ratio plots, as

given in Fig. 3.1.1(a), reveals that the proposed buck-derived converter achieves wider

voltage step- down than a basic buck converter. Also, by examining Fig 3.1.3(a), it



becomes evident that the suggested boost-derived converter attains a wider voltage

step-up than a basic boost converter.

The converter topologies shown in Figs. 3.1.1(a) and 3.1.3(a) are strikingly

similar. The idea proposed here is that these two topologies may be combined to

form a new two-switch topology, with an extended conversion range. Same

conclusion can be reached comparing the converters given in Figs. 3.1.1(c) and 3.1.3(c).

The proposed WIWO range converter topology is described in the next section.

3.2. WIWO DC–DC CONVERTER TOPOLOGY

Fig. 3.2. WIWO dc–dc converter topology.

3.2.1 Proposed WIWO DC–DC Converter Topology

The proposed WIWO dc–dc converter is illustrated in Fig. 3.2. The converter is

comprised of two active switches S1 and S2, tapped inductors L1 and L2 with turns

ratio n = n2 : n1 , diode D, and capacitive output filter C.

Specifically, note that the tapped inductor in Figs. 3.1.1 and 3.1.3 is

reconfigured into a pair of coupled inductors in Fig. 3.2.1. Being equivalent

electrically, this reconfiguration is beneficial from a practical point of view. In Fig.

3.2, S1 and S2 are connected to a common junction or midpoint. The midpoint is

periodically switched by S1 to ground, which allows recharging the bootstrap power

supply and reliable operation of the flying driver of the top switch S2.

Consequently, a standard half-bridge driver chip can be used with the low-side driver

operating the bottom switch S1 and the bootstrap high-side driver activating the top



switch S2.

WIWO can operate either in the step-down or the buck mode or in the step-up

or the boost mode. To operate the WIWO in the buck mode, the switch S1 is

assigned a high-frequency switching signal with a predetermined duty cycle D,

whereas S2 is switched complementarily to S1. The diode D is kept ON by the

inductor L2 current, which is assumed to be continuous. To operate WIWO in the

boost mode, the controller keeps S2 switch continuously ON and issues the

required duty cycle signal for the S1 switch. Thus, the diode D is forced to switch

on and off complementarily to S1.In both modes, the capacitor C filters the

pulsating current and provides a smoothed output voltage for the load R.

3.3 Control Scheme

Fig. 3.3. Proposed WIWO dc–dc converter and PWM control circuitry.

For the proper operation of WIWO, a modified PWM control circuitry is

required. The implementation is not unique. One possible realization of the

modulator is shown in Fig. 3.3. Here, a window comparator is employed to derive

the required switching signals for S1 and S2 by comparing the sawtooth ramp with

amplitude of Vm to the two control voltages VC and V . The control voltage VC



for the upper comparator is delivered by an external source, whereas the lower

comparator input signal is derived by the PWM circuitry, downshifting the control

voltage VC by Vm: V = VC−Vm. The relationship between the control voltage VC

and the sawtooth ramp amplitude Vm can be expressed by means of a variable m

as VC=mVm . WIWO operates in the buck mode when 0 <VC <Vm, i.e., when

0 ≤ m < 1. Here, the upper comparator generates the required duty cycle for the

S2 switch, whereas the lower comparator is in “1” state and commands the NAND

gate to provide the complimentary duty cycle for the S1 switch. Therefore, WIWO

operates similarly to a synchronous buck converter. On the other hand, for

Vm<VC<2Vm ,or 1≤m<2, the upper comparator is in “1” state and keeps S2

continuously ON, whereas the lower comparator and the NAND gate provide the

required duty cycle for the S1 switch. Thus, the converter enters the boost mode.

3.4 Operating Principle of the WIWO Converter

In the following, the steady-state operation of the proposed WIWO converter

is described. The analysis is performed assuming that the circuit is comprised of

ideal components. The coupling coefficient of the tapped inductor is assumed to be

unity. Under continuous inductor current (CCM) condition, the proposed WIWO

converter exhibits four topological states, as shown in Fig3.4.1. Here, the large

output filter capacitor is replaced by an ideal voltage source. The waveforms and

timing of WIWO for both buck and boost modes are illustrated in Fig3.4.2.



Fig3.4 .1 Four topological states of the WIWO converter. (a) Buck-mode

charging state. (b) Buck-mode discharging state. (c) Boost-mode charging state.

(d) Boost-mode discharging state.



Fig 3.4.2. Waveforms of the WIWO dc–dc converter. (a) Buck mode. (b) Boost

mode.

Buck Mode: State 1 (t0 ≤ t < t1 ) is the buck-mode charging state [see

Figs. 3 .4.1(a) and 3 .4 .2 (a)]. Here, the switch S2 is turned on and S1 is turned off.

The diode D conducts and the coupled inductors L1 and L2 are charged. The energy

is also transferred from dc source to load. State 2 (t1 ≤ t ≤ t2 ) is the buck-mode

discharging state [see Figs3.4.1(b) and 3.4.2(a)]. Here, the switch S2 is turned off also

cutting off the current in the L1 winding, whereas S1 is turned on and the diode D

conducts L2 current to the load.

Boost Mode: State 3 (t0 ≤ t < t1

) is the boost-mode charging state [see

Figs. 3.4.1(c) and 3.4.2(b)]. Here, the switches S1 and S2 are turned on charging the

L1 inductor. The diode D is cut off by the negative voltage induced inL2 winding.

The output voltage is supported by the capacitor C. State 4 (t1 ≤ t ≤ t2 ) is the

boost-mode discharging state [see Figs. 3.4.1(d) and 3.4.2(b)]. Here, the switch S2 is



still ON whereas S1 is turned off. Both windings L1 and L2 conduct through the

diode D and discharge the stored energy to the output.

3.5 Steady-State Analysis

The steady-state models of the proposed WIWO converter are shown in Fig

3.4.1. These models preserve the tapped-inductor symbol. More suitable for analysis

purposes, however, are the models of Fig 3.5. Here, the role of the magnetizing

inductance Lm is clearly shown. The detailed analysis was carried out in [11]

using state-space averaging technique. WIWO voltage conversion ratio, output

voltage ripple, voltage stresses, etc., were obtained. The characteristics of WIWO

are summarized in Table I(appendix) for a general case of n and separately for the

special case of n = 1.

Fig. 3.5. Switched circuit models. (a) State 1. (b) State 2. (c) State 3. (d) State 4.

WIWO voltage transfer characteristics M (n, m) are plotted in Fig3.6.1.

Clearly, for n=1, the voltage transfer ratio is smooth at the vicinity of the buck to

boost switchover point m=1, whereas for other values of n, the curves exhibit a

slope change. This statement can be verified analytically by calculating the

derivatives of M (m) at m = 1. Using the expressions for voltage conversion ratio

given in Table I, the result is ((n + 1)/n) for the buck mode and (n + 1) for the

boost mode. Obviously, the slope of WIWO dc–dc characteristic becomes



continuous for n = 1. Table I also presents the line-to-output and control-to-output

transfer functions. The small-signal transfer functions of the WIWO converter

were derived by linearizing the state-space equations around the operating point

[11]. The line-to-output and control-to-output transfer functions reveal strong

dependence on the operating point and a right-half-plane (RHP) zero. This is

also the case in other tapped-inductor t opo log i e s [13], [14]. These

characteristics make the WIWO compensation network design somewhat difficult.

3.6. RIPPLE AND EFFECTS OF RIPPLE

The most common meaning of ripple in electrical science is the small

unwanted residual periodic variation of the direct current (dc) output of a power

supply which has been derived from an alternating current (ac) source. This ripple is

due to incomplete suppression of the alternating waveform within the power supply.

As well as this time-varying phenomenon, there is a frequency domain ripple

that arises in some classes of filter and other signal processing networks. In this case

the periodic variation is a variation in the insertion loss of the network against

increasing frequency. The variation may not be strictly linearly periodic. In this

meaning also, ripple is usually to be considered an unwanted effect, its existence

being a compromise between the amount of ripple and other design parameters.

3.6.1 Time-domain ripple

Fig 3.6.1.1: Full-wave rectifier with a smoothing capacitor

Ripple factor (γ) may be defined as the ratio of the root mean square (rms)

value of the ripple voltage to the absolute value of the dc component of the output

voltage, usually expressed as a percentage. However, ripple voltage is also commonly

expressed as the peak-to-peak value. This is largely because peak-to-peak is both


http://en.wikipedia.org/wiki/Peak-to-peak

http://en.wikipedia.org/wiki/Absolute_value

http://en.wikipedia.org/wiki/Voltage

http://en.wikipedia.org/wiki/Root_mean_square

http://en.wikipedia.org/wiki/Frequency

http://en.wikipedia.org/wiki/Insertion_loss

http://en.wikipedia.org/wiki/Signal_processing

http://en.wikipedia.org/wiki/Filter_(signal_processing)

http://en.wikipedia.org/wiki/Frequency_domain

http://en.wikipedia.org/wiki/Waveform

http://en.wikipedia.org/wiki/Alternating_current

http://en.wikipedia.org/wiki/Direct_current

http://en.wikipedia.org/wiki/Periodic_function

http://en.wikipedia.org/wiki/Electricity

http://en.wikipedia.org/wiki/File:Rectifier_with_reservoir.svg


easier to measure on an oscilloscope and is simpler to calculate theoretically. Filter

circuits intended for the reduction of ripple are usually called smoothing circuits.

The simplest scenario in ac to dc conversion is a rectifier without any

smoothing circuitry at all. The ripple voltage is very large in this situation; the peak-

to-peak ripple voltage is equal to the peak ac voltage. A more common arrangement

is to allow the rectifier to work into a large smoothing capacitor which acts as a

reservoir. After a peak in output voltage the capacitor (C) supplies the current to the

load (R) and continues to do so until the capacitor voltage has fallen to the value of

the now rising next half-cycle of rectified voltage. At that point the rectifiers turn on

again and deliver current to the reservoir until peak voltage is again reached. If the

time constant, CR, is large in comparison to the period of the ac waveform, then a

reasonably accurate approximation can be made by assuming that the capacitor

voltage falls linearly. A further useful assumption can be made if the ripple is small

compared to the dc voltage. In this case the phase angle through which the rectifiers

conduct will be small and it can be assumed that the capacitor is discharging all the

way from one peak to the next with little loss of accuracy.

Fig 3.6.1.2: smoothed Ripple voltage from a full-wave rectifier

With the above assumptions the peak-to-peak ripple voltage can be calculated

as:

For a full-wave rectifier:


http://en.wikipedia.org/wiki/Phase_angle

http://en.wikipedia.org/wiki/Time_constant

http://en.wikipedia.org/wiki/Capacitor

http://en.wikipedia.org/wiki/Rectifier

http://en.wikipedia.org/wiki/Rectifier#Rectifier_output_smoothing

http://en.wikipedia.org/wiki/Oscilloscope

http://en.wikipedia.org/wiki/File:Smoothed_ripple.svg


For a half-wave rectification:

where

Vpp is the peak-to-peak ripple voltage and I is the current in the circuit

F is the frequency of the ac power and C is the capacitance

For the rms value of the ripple voltage, the calculation is more involved as the

shape of the ripple waveform has a bearing on the result. Assuming a sawtooth

waveform is a similar assumption to the ones above and yields the result

where

γ is the ripple factor and R is the resistance of the load

Another approach to reducing ripple is to use a series choke. A choke has a

filtering action and consequently produces a smoother waveform with less high-order

harmonics. Against this, the dc output is close to the average input voltage as opposed

to the higher voltage with the reservoir capacitor which is close to the peak input

voltage. With suitable approximations, the ripple factor is given by

Where ω is the angular frequency 2πf and L is the inductance of the choke

More complex arrangements are possible; the filter can be an LC ladder rather

than a simple choke or the filter and the reservoir capacitor can both be used to gain

the benefits of both. The most commonly seen of these is a low-pass Π-filter

consisting of a reservoir capacitor followed by a series choke followed by a further

shunt capacitor. However, use of chokes is deprecated in contemporary designs for


http://en.wikipedia.org/wiki/Topology_(electronics)#Simple_filter_topologies

http://en.wikipedia.org/wiki/Low-pass

http://en.wikipedia.org/wiki/Inductance

http://en.wikipedia.org/wiki/Reservoir_capacitor

http://en.wikipedia.org/wiki/Harmonics

http://en.wikipedia.org/wiki/Choke_(electronics)

http://en.wikipedia.org/wiki/Sawtooth_wave

http://en.wikipedia.org/wiki/Sawtooth_wave


economic reasons. A more common solution where good ripple rejection is required

is to use a reservoir capacitor to reduce the ripple to something manageable and then

pass through a voltage regulator circuit. The regulator circuit, as well as regulating

the output, will incidentally filter out nearly all of the ripple as long as the minimum

level of the ripple waveform does not go below the voltage being regulated to.

The majority of power supplies are now switched mode. The filtering

requirements for such power supplies are much easier to meet owing to the frequency

of the ripple waveform being very high. In traditional power supply designs the ripple

frequency is either equal to (half-wave), or twice (full-wave) the ac line frequency.

With switched mode power supplies the ripple frequency is not related to the line

frequency, but is instead related to the frequency of the chopper circuit.

3.6.2 Effects of ripple

Ripple is undesirable in many electronic applications for a variety of

reasons,the ripple frequency and its harmonics are within the audio band and will

therefore be audible on equipment such as radio receivers, equipment for playing

recordings and professional studio equipment.

The ripple frequency is within television video bandwidth. Analogue TV

receivers will exhibit a pattern of moving wavy lines if too much ripple is present.

The presence of ripple can reduce the resolution of electronic test and

measurement instruments. On an oscilloscope it will manifest itself as a visible

pattern on screen.

Within digital circuits, it reduces the threshold, as does any form of supply rail

noise, at which logic circuits give incorrect outputs and data is corrupted. High

amplitude ripple currents shorten the life of electrolytic capacitors.

3.7 PULSE WIDTH MODULATION( PWM)

Pulse Width Modulation, or PWM, is a technique for getting analog results with digital means. Digital control is used to create a square wave, a signal switched between on and off. This on-off pattern can simulate voltages in between full on (5 Volts) and off (0 Volts) by changing the portion of the time the signal spends on versus the time that the signal spends off. The duration of "on time" is called the


http://en.wikipedia.org/wiki/Electrolytic_capacitors

http://en.wikipedia.org/wiki/Chopper_(electronics)

http://en.wikipedia.org/wiki/Switched-mode_power_supply

http://en.wikipedia.org/wiki/Voltage_regulator


pulse width. To get varying analog values, you change, or modulate, that pulse width. If you repeat this on-off pattern fast enough with an LED for example, the result is as if the signal is a steady voltage between 0 and 5v controlling the brightness of the LED.

In the graphic below, the green lines represent a regular time period. This duration or period is the inverse of the PWM frequency. In other words, with PWM frequency at about 500Hz, the green lines would measure 2 milliseconds each. A call to analogWrite is on a scale of 0 - 255, such that analogWrite(255) requests a 100% duty cycle (always on), and analogWrite(127) is a 50% duty cycle (on half the time) for example.

A 100-W prototype WIWO converter was designed for input voltage range of

12–48 Vdc and a constant output voltage of 28 Vdc . The turn ratio of the tapped

inductor was set to n =1 with a total inductance of 400 µH. The switching

frequency of 200 kHz was chosen. The tapped inductors were wound on C058548A2

toroidal powder core, chosen for its low leakage, with 50 turns of AWG20 wire for

both windings. The design yielded 400 µH inductance with only 560 nH leakage

inductance. Two FDD2572 MOSFETs were paralleled to comprise the top switch

and two IRFR3518 were used for the low switch providing low Rds−ON and low



gate capacitance. Schottky diode 20CTQ150 was selected due to superior reverse

recovery characteristics.

Fig.3.6.1. Voltage transfer characteristics M (n, m) of the WIWO dc–dc

converter.

Experimental waveforms of WIWO converting 48 V input to 28 V output

(buck mode) are shown in Fig.3.6.2. In the buck mode, S2 is the leading switch,

gated by the duty cycle command shown as the bottom trace in Fig. 3.6.2, whereas

the bottom switch S1 is switched complementarily, similarly to a synchronous

converter. Switch voltages (see Fig.3.3 for definition) are shown as top two

waveforms in Fig.3.6.2. The middle traces show the winding currents. These were

measured by ac probe, so only the ripple components could be observed. As could be

seen, as the S2 switch conducts, both windings carry the same current. At the S2 is

turned off, the input current ceases whereas the output current is doubled in

amplitude, consistent with WIWO models in Fig.3.5(a) and (b). The ramp portion

of the current is hardly noticeable due to the relatively high frequency and

sufficiently large inductance value. The leakage inductance of L1 developed a turn-

OFF voltage spike across S1 that is smoothed by the snubber circuitry. The snubber is

used to clamp the voltage spike, as described later.



Fig.3.6.2. Experimental waveforms of the WIWO converter in the buck mode.

Top trace: drain voltage V1 of S1 switch (50 V/division, 2 µs/division); second

top trace: drain voltage V2 of S2 switch (50 V/division, 2 µs/division); middle

trace: input current Ii (0.2 A/division, 2 µs/division); second bottom trace: output

current Io (0.2 A/division, 2 µs/division); bottom trace: S2 switch gating voltage

(20 V/ division, 2 µs/division).

The experimental waveforms of WIWO in the boost mode with 12 V input

and 28 V output, under full-load condition, are shown in Fig 3.6.3. To supply the

power requirements of the load at lower input voltage range, WIWO calls for greater

input current, and therefore, turn-OFF voltage spike on S1 is observed. In the boost

mode, the S1 switch is the leading switch that is issued the duty cycle command,

shown as the bottom trace in Fig.3.6.3. Since in the buck mode the S2 switch is

constantly ON, the drain voltage of S2 and the drain voltage of S1 are almost

identical. The winding currents were measured by a high-frequency ac probe, and

therefore, only ac current components are shown as two middle traces in Fig.3.6.3.

As S1 switch conducts, the input winding carries the input current and is charging,

whereas the output current is cut off. As the S1 switch



Fig.3.6.3 Experimental waveforms of WIWO in the boost mode (see Fig.3.3 for

designation of variables). Top trace: drain voltage of S1 switch (20 V/division,2

µs/division); second top trace: drain voltage of S2 switch (20 V/division,2

µs/division); middle trace: input current Ii (0.5 A/division, 2 µs/division); second

bottom trace: output current Io (0.5 A/division, 2 µs/division); bottom trace: S1

switch gating voltage (20 V/division, 2 µs/division).



Fig.3.6.4 Comparison of K with Kcr i t for n =

1.

is cut off, both windings carry the same current and are discharging into the output

capacitor and feeding the load. For this reason, the currents ripple components appear

in antiphase, as predicted by WIWO models in Fig. 3.5(c) and (d). Also could be

seen is the snubber circuit resonant discharge as the snubber recycles the stored energy.



Fig 3.6.5.Comparison of the experimental and theoretical voltage conversion ratio

under different loading conditions. (a) K = 2. (b) K = 0.2. (c) K = 0.02.

With decreased load, the converter enters the discontinuous conduction mode



(DCM). To measure the tendency of the converter to operate in DCM, the parameter

K = (2Lm /RTs ) is defined as suggested in [15]. The critical value of K for n = 1 is

compared with K = 2, 0.2, 0.02 in Fig.3.6.4. The experimental voltage conversion

ratio M as function of m for different values of K plotted on top of the theoretical

curve is given in Fig.3.6.5 (a)–(c). Due to the parasitic resistances in the circuit, the

experimental voltage conversion ratio M is slightly lower than theoretical prediction.

For very same reason, the experimental M cannot become infinite and drops as m

approaches 2. A narrow buck to boost-mode transition can be observed on the WIWO

characteristic in the vicinity of m = 1. The conversion ratio in DCM is higher than that

in CCM, as shown in Fig. 3.6.5(b) and (c).

The efficiency of the experimental WIWO dc–dc converter for different dc

input voltages versus the load current is plotted in Fig.3.7.1. The output voltage was

kept at the nominal value of 28 Vdc. No attempt was made to optimize the

preliminary design, still the converter demonstrated high efficiency.



CHAPTER 4. HARDWARE ANDSOFTWARE IMPLEMENTATION

4.1 Introduction to Matlab

4.1.1 SimPowerSystems

SimPowerSystems and SimMechanics of the Physical Modeling product family

work together with Simulink to model electrical, mechanical and control systems.

4.1.2 Role of Simulation in Design

Electrical power systems are combinations of electrical and electromechanical

devices like motors and generators. Engineers working in this discipline are constantly

improving the performance of the systems. Requirements for drastically improved

efficiency have forced power system designers to use power electronic devices and

sophisticate control system concepts that tax traditional analysis tools and techniques.

Further complicating the analyst’s role is the fact that the system is often so nonlinear

that the only way to understand it is through simulation.

Land based power generation from hydroelectric, steam or other devices are not

the only use of power systems. A common attribute of these systems is their use of

power electronics and control systems to achieve their performance objectives.

SimPowerSystems is a modern tool that allows scientists and engineers to

rapidly and easily build models that simulate power systems. SimPowerSystems uses

the Simulink environment, allowing to build a model using simple click and drag

procedures. Not only can draw the circuit topology rapidly, but analysis of the circuit

can include its interactions with mechanical, thermal, control, and other disciplines.

This is possible because all the electrical parts of the simulation interact with the

extensive Simulink modeling library. Since Simulink uses MATLAB as its

computational engine, designers can also use MATLAB toolboxes and Simulink block

sets. SimPowerSystems and SimMechanics share a special Physical Modeling block

and connection line interface.



4.1.3 SimPowerSystems Libraries

SimPowerSystems can be made to work rapidly. The libraries contain models of

typical power equipment such as transformers, lines, machines and power electronics.

These models are proven ones coming from textbooks and their validity is based on the

experience of the Power Systems testing and Simulation Laboratory of Hydro Quebec, a

large North American utility located in Canada and also on the experience of Ecole de

Technologies superieure and Universite Laval.

The capabilities of SimPowerSystems for modeling typical electrical systems are

illustrated in demonstration files. And for users who want to refresh their knowledge of

power system theory, there are also self-learning case studies.

4.2 Design and Simulating of a Simple Circuit

SimPower Systems allows building and simulating of electrical circuits

containing linear and nonlinear elements.

In this section it is possible to

1 Explore the powerlib library of SimPowerSystems

2 Learn how to build a simple circuit from the powerlib library

3 Interconnect Simulink blocks with your circuit

This section contains discussion of the following topics:

1 Building the Electrical Circuit with powerlib Library

2 Interfacing the Electrical Circuit with Simulink

3 Measuring Voltages and Currents

4 Basic Principles of Connecting Capacitors and Inductors

5 Using the Powerlib Block to Simulate SimPowerSystems Models



4.3 Designing a Electrical Circuit with Powerlib Library

The graphical user interface makes use of the Simulink functionality to

interconnect various electrical components. The electrical components are grouped in a

special library called powerlib.

SimPowerSystems library is opened by entering the following command at the

MATLAB prompt.

Powerlib

This command displays a Simulink window showing icons of different block libraries.

Fig 4.1 Powerlib Library

It is possible to open these libraries to produce the windows containing the

blocks to be copied into given circuit. Each component is represented by a special icon

having one or several inputs and outputs corresponding to the different terminals of the

component.



4.4 Interfacing Electrical Circuit with Simulink

The Voltage Measurement block acts as an interface between the

SimPowerSystems blocks and the Simulink blocks. The Voltage Measurement block

converts the measured voltages into Simulink signals.The Current Measurement block

from the Measurements library of powerlib can also be used to convert any measured

current into a Simulink signal.

It is also possible to interface from Simulink blocks to the electrical system. For

example, it is possible to use the Controlled Voltage Source block to inject a voltage in

an electrical circuit, as shown in the following figure.

Fig 4.2 Interfacing electrical circuit with simulink



4.5 Measuring Voltages and Currents

To measure a current using a Current Measurement block, the positive direction

of current is indicated on the block icon (positive current flowing from positive terminal

to negative terminal). Similarly, when to measure a voltage using a Voltage

Measurement block, the measured voltage is the voltage of the +ve terminal with

respect to the -ve terminal. However, when voltages and currents of blocks from the

Elements library are measured using the Multi-meter block, the voltage and current

polarities are not immediately obvious because blocks might have been rotated and

there are no signs indicating polarities on the block icons. Unlike Simulink signal lines

and input and output ports, the Physical Modeling connection lines and terminal ports of

SimPowerSystems lack intrinsic directionality. The voltage and current polarities are

determined, not by line direction, but instead by block orientation.

4.5.1 Power Supply

Single phase AC supply is given to bridge rectifier. It converts AC into DC. The

present chapter introduces the operation of power supply circuits built using filters,

rectifiers, and then voltage regulators. Starting with an AC voltage, a steady DC voltage

is obtained by rectifying the AC voltage, then filtering to a DC level, and finally,

regulating to obtain a desired fixed DC voltage.

4.5.2 Converter

Converter is a device which convert AC to DC since high voltage dc supply is

required at the input of the inverter.

4.5.3 Filtering Unit

Filter circuits which is usually a capacitor acting as a surge arrester always

follows the rectifier unit. This capacitor is also called as a decoupling capacitor or a

bypassing capacitor. It passes only low frequency signals and bypasses high frequency

signals.



4.5.4 Driver Circuit

The function of driver circuit is to amplify the voltage generated in the

microcontroller. In the above circuit the microcontroller generates 2v to 5v. This

voltage is not sufficient for driving the MOSFET. Therefore this voltage is amplified

using the driver circuit which will be discussed in detail in the next chapter.

4.5.5 Microcontroller

Microcontroller (AT89C51) is one of the most commonly used microcontrollers

especially in automotive, industrial appliances and consumer applications. However, as

the functionality of the components such as timers, A/D converters, I/O Ports are

explained in detail in Chapter 7.0.1 and the reader can flashback to this section to view

the schematics and the specifications.

4.6 POWER SUPPLY UNIT

All electronic circuits works only in low DC voltage, so a power supply unit

is required to provide the appropriate voltage supply for their proper functioning.

This unit consists of transformer, rectifier, filter and regulator. AC voltage of typically

230V RMS is connected to a transformer which step down the voltage to the desired

AC voltage. A diode rectifier that provides the full wave rectified voltage that is

initially filtered by a simple capacitor filter to produce a DC voltage. This resulting

DC voltage usually has some ripple or AC voltage variation. A regulator circuit can

use this DC input to provide DC voltage that not only has much less ripple voltage but

also remains at the same DC value, even when the input DC voltage varies somewhat or

the load connected to the output DC voltage changes.

Fig 4.3 General Block of Power Supply Unit

4.6.1 Transformer



A transformer is a static device in which electric power in one circuit is

transferred into electric power of same frequency in another circuit .It can raise or

lower the voltage in the circuit, but with a corresponding decrease or increase in

current. It works with the principle of mutual induction. In this project a step-down

transformer is used to provide necessary supply of 12 V for the electronic circuits.

4.6.2 Rectifier

A DC level obtained from a sinusoidal input can be improved 100% using a

process called full wave rectification. Here in this project for full wave rectification a

bridge rectifier is used.

In the bridge rectifier the diodes may be of variable types like 1N4001, 1N4003,

1N4004, 1N4005, IN4007 etc can be used. But in this project 1N4007 is used because it

can withstand up to 1000V.

4.6.3 Filters

In order to obtain a dc voltage of 0 Hz, a low pass capacitive filter circuit is used

where a capacitor is connected at the rectifier output and a DC voltage without ripples is

obtained across it. The filtered waveform is essentially a DC voltage with negligible ripples

and it is ultimately fed to the load.

4.6.4 Regulators

The filtered output voltage from the capacitor is finally regulated. The voltage

regulator is a device, which maintains the output voltage constant irrespective of the change

in supply variations, load variations and temperature changes. Here a fixed voltage

regulator namely LM7805 is used. The IC LM7805 is a +5V regulator which is used for

microcontroller.



Features and Description of Regulators

1. Output Current up to 1A

2. Output Voltages of 5, 6, 8, 9, 10, 12, 15, 18, 24V

3. Thermal Overload Protection

4. Short Circuit Protection

5. Output Transistor Safe Operating Area Protection

The KA78XX/KA78XXA series of three-terminal positive regulator are

available in the TO-220/D-PAK package and with several fixed output voltages,

making them useful in a wide range of applications. Each type employs internal

current limiting, thermal shutdown and safe operating area protection, making it

essentially indestructable. If adequate heat sinking is provided, they can deliver over

1A output current. Although designed primarily as fixed voltage regulators, these

devices can be used with external components to obtain adjustable voltages and

currents.

Electrical Characteristics of KA7805A

Load and line regulation are specified at constant junction temperature. Change

in Vo due to heating effects must be taken into account separately. Pulse testing with

low duty is used.

Electrical Characteristics of KA7805A

Load and line regulation are specified at constant junction temperature. Change

in Vo due to heating effects must be taken into account separately. Pulse testing with

low duty is used.

This circuit can give +5V output at about 150 mA current, but it can be

increased to 1 A when good cooling is added to 7805 regulator chip. The circuit has

over overload and thermal protection.



Figure 4.4: Circuit diagram of the power supply.

The capacitors must have enough high voltage rating to safely handle the input

voltage feed to circuit. The circuit is very easy to build for example into a piece of

overboard.

Figure 4.5: Pin out of the 7805 regulator IC.

1. Unregulated voltage in

2. Ground

3. Regulated voltage out

Component list

1. 7805 regulator IC

2. 100 uF electrolytic capacitor, at least 25V voltage rating

3. 10 uF electrolytic capacitor, at least 6V voltage rating

4. 100 nF ceramic or polyester capacitor


http://www.epanorama.net/circuits/psu_5v.gif

http://www.epanorama.net/circuits/7805.gif


Figure 4.6: circuit diagram of power supply



4.7 DRIVER CIRCUIT

The main purpose of driver circuit is to enhance the switching voltage for

the MOSFET or any switching device and also to isolate the power circuit from the

microcontroller circuit. Since the power circuit current must not enter into the

microcontroller circuit, MCT2E which is the opto coupler will be connected to the

buffer CD4050 which send pulse signals of 5V from microcontroller to the driver

circuit. MCT2E is the device which isolates the power circuit with the

microcontroller circuit. After it gets the signal from the microcontroller it will get

enhanced using the 2N2222 transistor to higher level of voltage.After this voltage

gets regulated by the use of darlington pair. The darlington is made of 2N2222 (NPN)

and SK100 (PNP) transistor.330 OHM

MCT2E

1 K22 K

100 OHM

100 OHM

100 OHM

1 K

1000 mF/25 A

G

GROUND

330 OHM

MCT2E

1 K22 K

100 OHM

100 OHM

100 OHM

1 K

1000 mF/25 A

G

GROUND

330 OHM

MCT2E

1 K22 K

100 OHM

100 OHM

100 OHM

1 K

1000 mF/25 A

G

GROUND

330 OHM

MCT2E

1 K22 K

100 OHM

100 OHM

100 OHM

1 K

1000 mF/25 A

G

GROUND

330 OHM

MCT2E

22 K

100 OHM

1 K

100 OHM

1 K

100 OHM

100 OHM

G

GROUND

100 OHM

GROUND

1000 mF/25 A

G

330 OHM

100 OHM

1000 mF/25 A

1 K

22 K

MCT2E

1 K

Fig 4.7: Circuit diagram of driver circuit unit

Components used:

1. MOSFET

IRFP460

2. Diode

1N4007

3. Capacitors

1000µF/50V

1000µF/25V



4. Opto coupler

MCT2E

5. Transistors

2N2222

SK100

6. Resistors

1K

100Ω

7. Transformers

230V/12V

4.8 Opto coupler

4.8.1Description

Opto coupler or opto isolator is a combination of light source and light detector

in the same package. They are used to couple signal from one point to the other

optically, by providing a complete electrical isolation between them. This kind of

isolation is provided between a low power control circuit and high power output

circuit, to protect the control circuit.

Fig 4.8 Opto coupler schematic diagram



Depending on the type of light sources and detector used it is possible to get a

variety of optocouplers, they are as follows:

1. LED LDR optocoupler.

2. LED photodiode optocoupler.

3. LED phototransistor optocoupler.

4.8.2 Characteristics

1. Current transfer ratio (CTR).

2. Isolation voltage.

3. Response time.

4. Common mode rejection.

4.8.3 LED Phototransistor Opto coupler

The LED phototransistor opto coupler is an infrared LED acts as the high

source and the phototransistor acts as a photo detector. This is the most popularly used

optocoupler, because it does not need any additional amplification. When the pulse at

the input goes high, the LED turns ON.

The light emitted by the LED is focused on the CB junction of the

phototransistor. In response to this light, photo current starts flowing which acts as

base current for the phototransistor. The collector current of phototransistor starts

flowing. As soon as the input pulse reduces to zero the LED turns OFF and the

collector current of phototransistor reduces to zero.

4.8.4 Applications

1. AC to DC converters used for DC motor speed control.

2. High power choppers.

3. High power inverters.



4.9 MOS Transistors

1. Two primary types:

a. MOSFET (Metal-Oxide-Semiconductor FET).

b. JFET( Junction FET)

2. MOS transistors can be:

a. N-Channel

i. Enhancement mode

ii. Depletion mode

b. P-Channel

i. Enhancement mode

ii. Depletion mode

Fig 4.9 Diagram of MOSFET

MOSFET

Fig 4.9 Symbolic representation of MOSFET



5.0.Pulse Generator

5.0.1MICROCONTROLLER (AT89C51)

Introduction to 8051 Microcomputer

Fig 5.0.1 shows a functional block of the internal operation of an 8051

microcomputer. The internal components of the chip are shown within the broken line

box.

Fig 5.0.1 8051 functional block diagram.



Fig 5.0.2 shows the external code memory and data memory connected to the 8051

chip.

Note – part of the external code memory can be located within the chip but

ignore this feature for now. Also, variants of the chip will allow a lot more memory

devices and I/O devices to be accommodate within the chip but such enhanced

features will not be considered right now.

Fig 5.0.2 8051 chip with external memory

Fig 5.0.3 Simplified diagram of a Pentium processor

A modern PC is powered by a Pentium processor (or equivalent), which is really

a very powerful microprocessor. Where the 8051 microcontroller represents the low end



of the market in terms of processing power, the Pentium processor is one of the most

complex processors in the world. Fig 5.0.3 shows a simplified block diagram of the

Pentium processor and a simple comparison between the 8051 and the Pentium is given

in the table 5.0.4 below.

5.0.4 Simple comparison of Pentium Vs 8051

FEATURE 8051 PENTIUM COMMENT

Clock Speed

12MHz typical

but 60MHz ICs

available

1,000 MHz

(1GHz.)

8051 internally divides clock by 12 so for

12MHz. clock effective clock rate is just

1MHz.

Address bus 16 bits 32 bits

8051 can address 216 or 64Kbytes of memory.

Pentium can address 232 or 4 Giga Bytes of

memory.

Data bus 8 bits 64 bits

Pentium’s wide bus allows very fast data

transfers.

ALU width 8 bits 32 bits

But Pentium has multiple 32 bit ALUs along

with floating-point units.

Applications

Domestic

appliances,

Peripherals,

automotive etc.

Personal

Computers

And other high

performance areas.

Power

consumption

Small fraction of

a watt

Tens of watts

Pentium runs hot as power consumption

increases with frequency.

Cost of chip

About 2 Euros.

In value

About 200 Euros –

Depending on

spec.

Table 7.0.4 Comparison of Pentium Vs 8051

The basic 8051 chip includes a number of peripheral I/O devices including two

Timer/Counters, 8-bit I/O ports and a UART (Universal Asynchronous Receiver

Transmitter). The inclusion of such devices on the 8051 chip is shown in Fig 8.4.

These I/O devices will be described later.



Fig 5.0.5 8051 showing the on-chip I/O devices

5.0.2Memory and Register Organization



The 8051 has a separate memory space for code (programs) and data. In an

actual implementation the external memory may, in fact, be contained within the

microcomputer chip. However, the definitions of internal and external memory to be

consistent with 8051 instructions which operate on memory is used. Note the

separation of the code and data memory in the 8051 architecture is a little unusual.

5.0.2.1 External Code Memory

The executable program code is stored in this code memory. The code memory

size is limited to 64KBytes (in a standard 8051). The code memory is read-only in

normal operation and is programmed under special conditions e.g. it is a PROM or a

Flash RAM type of memory.

5.0.2.2 External RAM Data Memory

This is read-write memory and is available for storage of data up to 64Kbytes

(in a standard 8051).

Internal Memory

The 8051’s on-chip memory consists of 256 memory bytes organized as

follows:

First 128 bytes: 00h to 1Fh Register Banks

20h to 2Fh Bit Addressable RAM

30 to 7Fh General Purpose RAM

Next 128 bytes: 80h to FFh Special Function Registers

The first 128 bytes of internal memory is organized as shown in Fig 8.2 and is

referred to as Internal RAM or IRAM.



7.0.3 Pin Description

P 89C 51R C 2

2930

40

20

3 1

19

18

9

3938373635343332

12345678

2122232425262728

1011121314151617

PSENA LE

VC C

GND

E A

X1

X2

R ST

P 0 . 0 /AD 0P 0 . 1 /AD 1P 0 . 2 /AD 2P 0 . 3 /AD 3P 0 . 4 /AD 4P 0 . 5 /AD 5P 0 . 6 /AD 6P 0 . 7 /AD 7

P 1 . 0 / T2P 1 . 1 / T2E XP 1 . 2 /EC I

P 1 . 3 /C EX0P 1 . 4 /C EX1P 1 . 5 /C EX2P 1 . 6 /C EX3P 1 . 7 /C EX4

P 2 . 0 /A 8P 2 . 1 /A 9P 2 . 2 /A 10P 2 . 3 /A 11P 2 . 4 /A 12P 2 . 5 /A 13P 2 . 6 /A 14P 2 . 7 /A 15

P 3 . 0 /R XDP 3 . 1 / TXDP 3 . 2 / IN T0P 3 . 3 / IN T1P 3 . 4 / T0P 3 . 5 / T1P 3 . 6 /W RP 3 . 7 /R D

Fig 5.0.3 Pin description of 89C51



VCC

Supply voltage.

GND

Ground.

Port 0

Port 0 is an 8-bit open-drain bi-directional I/O port. As an output port, each pin

can sink eight TTL (Transistor Transistor Logic) inputs. When 1s are written to port 0

pins, the pins can be used as high impedance inputs. Port 0 may also be configured to

be the multiplexed low order address/data bus during accesses to external program and

data memory. In this mode P0 has internal pull-ups. Port 0 also receives the code bytes

during Flash programming, and outputs the code bytes during program verification.

External pull-ups are required during program verification.

Port 1

Port 1 is an 8-bit bi-directional I/O port with internal pull-ups. The Port 1 output

buffers can sink/source four TTL (Transistor Transistor Logic) inputs. When 1s are

written to Port 1 pins they are pulled high by the internal pull-ups and can be used as

inputs. As inputs, Port 1 pins that are externally being pulled low will source current

because of the internal pull-ups. Port 1 also receives the low-order address bytes during

Flash programming and verification.

Port 2


buffers can sink/source four TTL(Transistor Transistor Logic) inputs. When 1s are

written to Port 2 pins they are pulled high by the internal pull-ups and can be used as

inputs. As inputs Port 2 pins that are externally being pulled low will source current

because of the internal pull-ups. Port 2 emits the high-order address byte during fetches



from external program memory and during accesses to external data memory that use

16-bit addresses.

Port 3


buffers can sink/source four TTL inputs. When 1s are written to Port 3 pins they are

pulled high by the internal pull-ups and can be used as inputs. As inputs, Port 3 pins that

are externally being pulled low will source current because of the pull-ups.

RST

Reset input. A high on this pin for two machine cycles while the oscillator is

running resets the device.

ALE/PROG

Address Latch Enable output pulse for latching the low byte of the address

during accesses to external memory. This pin is also the program pulse input (PROG)

during Flash programming.

In normal operation ALE is emitted at a constant rate of 1/6 the oscillator

frequency, and may be used for external timing or clocking purposes. Note, however,

that one ALE pulse is skipped during each access to external Data Memory. If desired,

ALE operation can be disabled by setting bit 0 of SFR location 8EH. With the bit set,

ALE is active only during a MOVX or MOVC instruction. Otherwise, the pin is weakly

pulled high. Setting the ALE-disable bit has no effect if the microcontroller is in

external execution mode.

PSEN

Program Store Enable is the read strobe to external program memory. When the

AT89C51 is executing code from external program memory, PSEN is activated twice

each machine cycle, except that two PSEN activations are skipped during each access to

external data memory.

EA/VPP

External Access Enable. EA must be strapped to GND in order to enable the

device to fetch code from external program memory locations starting at 0000H up to

FFFFH. Note, however, that if lock bit 1 is programmed, EA will be internally latched

on reset. EA should be strapped to VCC for internal program executions. This pin also



receives the 12V programming enable voltage (VPP) during Flash programming, for

parts that require 12V VPP.

XTAL1

Input to the inverting oscillator amplifier and input to the internal clock

operating circuit.

XTAL2

Output from the inverting oscillator amplifier.

5.0.4 Features of AT89C51

1. Compatible with MCS-51™ Products

2. 4K Bytes of In-System Reprogrammable Flash Memory

3. Endurance: 1,000 Write/Erase Cycles

4. Fully Static Operation: 0 Hz to 24 MHz

5. Three-level Program Memory Lock

6. 128 x 8-bit Internal RAM

7. 32 Programmable I/O Lines

8. Two 16-bit Timer/Counters

9. Six Interrupt Sources

10. Programmable Serial Channel

11. Low-power Idle and Power-down Modes

The AT89C51 is a low-power, high-performance CMOS 8-bit microcomputer with 4K

bytes of Flash programmable and erasable read only memory (PEROM). The device is

manufactured using Atmel’s high-density nonvolatile memory technology and is

compatible with the industry-standard MCS-51 instruction set and pin out. The on-chip

Flash allows the program memory to be reprogrammed in-system or by a conventional

nonvolatile memory programmer. By combining a versatile 8-bit CPU with Flash on a

monolithic chip, the Atmel AT89C51 is a powerful microcomputer which provides a

highly-flexible and cost-effective solution to many embedded control applications.



5.1 APPLICATIONS

A continuously conducting diode D has a considerable forward voltage drop.

This is not desirable for low-output- voltage applications. The voltage drop can be

reduced using a synchronous rectifier with low Rds -ON instead of the diode, as

shown in Fig 7.0.1.

Fig 5.1.1. Experimental WIWO converter efficiency.

Fig5.1.2. WIWO dc–dc converter with the synchronous rectifier.



Fig 5.1.3. Bidirectional WIWO dc–dc converter.

Interchanging the position of the inductor L2 and switch S3, as shown in

Fig 5.1.3, the WIWO topology becomes symmetrical. This also allows driving the top

switch S3 with another flying driver. An additional advantage of the circuit in Fig

5.1.3 is the ability to sustain a bidirectional power flow. The direction of the power

flow can be controlled applying a single-pole double-throw switch, which may be

controlled manually or automatically, as illustrated in Fig5.1.3. This WIWO can be

used in a battery charging and discharging application. With the switch in position 1,

the power flows from the left port to the right port, whereas with the switch in

position 2, the power flows in a reverse direction from the right port to the left port.



Fig 5.1.4. WIWO PFC ac–dc converter.

The WIWO dc–dc converter can also be used for PFC application (see

Fig5.1.4). Here, a sinusoidal line voltage is fed into the rectifier input. The WIWO dc–

dc converter can accept the rectified voltage and directly produce the required low dc

output. With the line voltage greater than the output, the converter works in the buck

mode. As the line drops below the output voltage, WIWO enters the boost mode.

Fig. 5.1.5 Hardware Implementation

5.2 SOFTWARE CODINGS



5.2..1 Introduction to Keil Microvision2

Keil μVision2 features include

1. Project Setup for the Make and Build Process

2. Editor facilities for Modifying and Correcting Source Code

3. Program Debugging and Additional Test Utilities

4. The Device Database makes it easy to start writing programs for a particular

CPU. Just select the microcontroller to be used and μVision2 sets the necessary

options automatically.

5. New devices may be added to the database as the need arises. μVision2 provides

a Books tab in the Project window where extensive on-line manuals for the tool

chain and selected CPU are found. Double-click on a book title to open the on-

line manual.

6. Most dialogs have a help button which provides detailed information about the

dialog controls. To get help on menu items, select the item and press F1.

7. μVision2 lets us set the options for all files in a target, a group, or even a single

source file.

8. The options dialog opens via the local menu in the Project window.

9. In the Target page of this dialog, the CPU and memory parameters of the target

system may be specified.

10. μVision2 uses this information to configure basic tool options including the

linker/locater settings and the simulator driver.

11. The output page defines the output files generated by the assembler, compiler,

and linker.

Build project

1. Start compiling and assembling target application with the build target button on

the toolbar.



2. The tool chain stores include and source file dependencies. This information is

used during the make process to build only those files that have changed.

3. Optionally, incremental retranslations are performed when global optimizing is

enabled.

4. The Build page of the output window lists tool information during the code

generation. Double-click on error messages to correct syntax errors in the

program. Errors are correctly located even after insert or delete source lines.

Break points

1. μVision2 allows to set program breakpoints while writing source text.

Simply the buttons on the editor toolbar are used to mark the breakpoints

on source lines.

2. After making the program, the Debugger with the debug toolbar button.

3. Breakpoints that are set while editing are activated in our debugging

session. μVision2 marks the status of the source lines in the attributes

column of the editor window. This provides a quick overview of the

current breakpoint settings.

Utilities

μVision2 contains many powerful functions that helps to complete projects on

time. For example, the Find in files dialog performs a text search in all specified files.

The search results are displayed in the Find in Files page of the Output window. This

feature is used to locate all uses of a function or variable.

Code Execution

1. The buttons on the toolbar are used to step through application program.

2. The run button executes code until a breakpoint is reached.

3. When Trace Recording is enabled, the Show traced records button lists the last

1024 instructions that were executed. Trace recording allows analyzing the

program flow prior to a breakpoint.

Simulator



Peripheral Simulation

μVision2 simulates the on-chip peripherals of numerous microcontrollers. When

a CPU is selected from the device database to configure the project, μVision2

automatically configures the peripheral simulator.

With its logical and timing simulation, it is possible to test an application before

the target hardware is even available. The simulator makes it easy to test hardware

defects and critical situations which are difficult to debug with real hardware.

CHAPTER 5 . CONCLUSION

This paper has presented a new WIWO dc–dc converter, which is an integration

of buck and boost converters with coupled inductors. The paper described WIWO

principles of operation and offers a comprehensive summary of WIWO analytical

characteristics. Simulation and experimental results were also reported. A modified

PWM modulator scheme required to make the converter work coherently was also

suggested. A prototype WIWO dc–dc converter was built and tested. The converter

demonstrated in practice the WIWO dc–dc conversion ratio.

The new converter topology has several advantages. The WIWO retains the

features of both the buck and the boost converters; however, it achieves wider step-up

and wider step-down dc–dc conversion range. The WIWO converter can operate with

an input source with broadly varying voltage or, alternatively, feed loads with variable

operating voltage such as dc motors. The converter has a simple structure and

moderate component count. The advantageous buck feature allows turning off the

output voltage on demand. WIWO is also inherently capable of limiting the inrush

current and can protect the output in the case of a short circuit. Due to the nonlinear

characteristics, WIWO can avoid operating at extreme duty cycle. As a result, WIWO

efficiency remains high even throughout large input voltage swing. The transition



between the operating modes is inherently smooth, and causes no transient disturbance

in the average current. Among the disadvantages of WIWO is the coupled inductor

whose leakage causes oscillation and high voltage spike across the switches. Clamp

circuits are needed to clamp voltage spikes upon switches, so as to recycle the leakage

energy. Another disadvantage of WIWOs is that small-signal transfer functions include

an RHP zero, and therefore, WIWO is somewhat difficult to stabilize using a single

voltage loop. To resolve the dynamic problem, current loop should be employed,

which is a good practice in any case. An additional disadvantage is that WIWO does

not provide isolation. This, however, may not be much of a problem in systems with

multiple stages.

Modifications of the WIWO to synchronous WIWO dc–dc converter, bidirectional

WIWO dc–dc converter, and WIWO dc–dc converter for PFC are possible. Numerous

advantages indicate WIWO as a viable candidate for many industrial applications.

BIBLOGRAPHY

[1] D. Maksimovic and S. Cuk, “Switching converter with wide dc conversion

range,”

IEEE Trans. Power Electron., vol. 6, no. 1, pp. 151–157, Jan. 1991.

[2] K. Yao, M. Ye, M. Xu, and F. C. Lee, “Tapped-inductor buck converter for high-

step-down dc–dc conversion,” IEEE Trans. Power Electron., vol. 20, no. 4, pp.

775–780, Jul. 2005.

[3] J.-H. Park and B.-H. Cho, “Nonisolation soft-switching buck converter with

tapped-inductor for wide-input extreme step-down applications,” IEEE Trans.

Circuits Syst. I, Reg. Papers, vol. 54, no. 8, pp. 1809–1818, Aug. 2007.

[4] K. Yao, Y. Ren, J. Wei, M. Xu, and F. Lee, “A family of buck type dc–dc

converters with autotransformers,” in Proc. Appl. Power Electron. Conf. Expo.

(APEC 2003), pp. 114–120.

[5] K. Nishijima, K. Abe, D. Ishida, T. Nakano, T. Nabeshima, T. Sato, and K.

Harada, “A novel tapped-inductor buck converter for divided power distribution

system,” in Proc. IEEE PESC Conf. (PESC 2006), Jun., 18–22, pp. 1–6.

[6] G. Spiazzi and S. Buso, “Power factor preregulator based on modified tapped-



inductor buck converter,” in Proc. IEEE PESC Conf., 1998, vol. 2, pp. 873–879.

[7] F. L. Luo and H. Ye, “Positive output cascade boost converters,” Proc.Inst.

Electr. Eng. Electr. Power Appl., vol. 151, no. 5, pp. 590–606, Sep.2004.

[8] Q. Zhao and F. C. Lee, “High efficiency, high step-up dc–dc converters,”IEEE

Trans. Power Electron., vol. 18, no. 1, pp. 65–73, Jan. 2003.

[9] N. Vazquez, L. Estrada, C. Hernandez, and E. Rodriguez, “The tapped- inductor

boost converter,” in Proc. IEEE Int. Symp. Ind. Electron., Jun.4–7, 2007, pp.

538–543.

[10] K. C. Tseng and T. J. Liang, “Novel high efficiency step-up converter,” Proc.

Inst. Electr. Eng. Electr. Power Appl., vol. 151, no. 2, pp. 182–190, Mar. 2004.

[11] H. Cheng, “Wide input wide output (WIWO) dc–dc converter,” Master’s thesis,

Univ. California, Irvine, Dec. 2007.

[12] H. Cheng and K. Smedley, “Wide input wide output (WIWO) dc–dc converter,”

in Proc. IEEE Appl. Power Electron. Conf. Expo., 2008, pp. 1562–1568.
