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© Semiconductor Components Industries, LLC, 2017December, 2017 −
Rev. 1
1 Publication Order Number:NCP4306/D
NCP4306
Secondary SideSynchronous RectificationDriver for High
EfficiencySMPS Topologies
The NCP4306 is high performance driver tailored to control
asynchronous rectification MOSFET in switch mode power
supplies.Thanks to its high performance drivers and versatility, it
can be used invarious topologies such as DCM or CCM flyback, quasi
resonantflyback, forward and half bridge resonant LLC.
The combination of externally or fixed adjustable
minimumoff-time and on-time blanking periods helps to fight the
ringinginduced by the PCB layout and other parasitic elements. A
reliable andnoise less operation of the SR system is insured due to
the SelfSynchronization feature. The NCP4306 also utilizes
Kelvinconnection of the driver to the MOSFET to achieve high
efficiencyoperation at full load and utilizes a light load
detection architecture toachieve high efficiency at light load.
The precise turn−off threshold, extremely low turn−off delay
timeand high sink current capability of the driver allow the
maximumsynchronous rectification MOSFET conduction time and
enablesmaximum SMPS efficiency. The high accuracy driver and 5 V
gateclamp enables the use of GaN MOSFETs.
Features
• Self−Contained Control of Synchronous Rectifier in CCM, DCM
andQR for Flyback or LLC Applications
• Precise True Secondary Zero Current Detection• Typically 15 ns
Turn off Delay from Current Sense Input to Driver• Rugged Current
Sense Pin (up to 200 V)• Ultrafast Turn−off Trigger Interface /
Disable Input (10.5 ns)• Adjustable or Fixed Minimum ON−Time•
Adjustable or Fixed Minimum OFF-Time with Ringing Detection•
Improved Robust Self Synchronization Capability• 7 A / 2 A Peak
Current Sink / Source Drive Capability• Operating Voltage Range up
to VCC = 35 V• Automatic Light−load Disable Mode• GaN Transistor
Driving Capability• Low Startup and Disable Current Consumption•
Maximum Operation Frequency up to 1 MHz• TSOP6 and SOIC-8 Packages•
This is a Pb−Free Device
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SOIC−8 NBCASE 751−07
MARKING DIAGRAMS
Typical Applications
• Notebook Adapters• High Power Density AC / DC Power
Supplies (Cell Phone Chargers)• LCD TVs• All SMPS with High
Efficiency
Requirements
TSOP−6 CASE 318G−02
1
8
XXXXX = Specific Device CodeA = Assembly LocationL = Wafer LotY
= YearW = Work Week� = Pb−Free Package
XXXXXALYWX
�1
8
IC(Pb−Free)
XXXAYW��
1
(Note: Microdot may be in either location)
1
SOIC−8 NB TSOP−6
IC(Pb−Free)
See detailed ordering and shipping information on page 2 ofthis
data sheet.
ORDERING INFORMATION
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ORDERING INFORMATION TABLE
Table 1. AVAILABLE DEVICES
Device Package Package Marking Packing Shipping †
NCP4306AAAZZZADR2G SOIC8 6AAAZZZA SOIC−8(Pb−Free)
2500 / Tape and Reel
NCP4306AADZZZADR2G 6AADZZZA
NCP4306AAHZZZADR2G 6AAHZZZA
NCP4306DADZZDASNT1G TSOP6 6AC TSOP−6(Pb−Free)
3000 / Tape and Reel
NCP4306DAHZZAASNT1G 6AD
†For information on tape and reel specifications, including part
orientation and tape sizes, please refer to our Tape and Reel
PackagingSpecification Brochure, BRD8011 / D.
See the ON Semiconductor Device Nomenclature document (TND310 /
D) for a full description of the naming conventionused for image
sensors. For reference documentation, including information on
evaluation kits, please visit our web site atwww.onsemi.com.
Figure 1. Typical Application Example – LLC Converter with
optional LLD and Trigger Utilization
+VBULK
LLCSTAGE
CONTROL
M1
M2 N1
C1
OK1
N3
N2
Tr1
R1 C3
+VOUT
RTN
D1
R2
C4
M4
M3
C2
NCP4306
VCCMIN_TOFF
MIN_TONLLD
DRV
GNDCS
TRIG
VCCMIN_TOFF
MIN_TONLLD
DRVGND
CSTRIG
NCP4306R
MIN
_TO
FF
RM
IN_T
ON
RLL
D
RM
IN_T
OF
F
RM
IN_T
ON
RLL
D
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Figure 2. Typical Application Example – DCM, CCM or QR Flyback
Converter with optional LLD and disabled TRIG
VBULK
FLYBACKCONTROLCIRCUITRY
M1
C1
OK1
R1
D3
C2
TR1
R3
M2
GNDC4
D5NCP4306
RM
IN_T
OF
F
RLL
D
R2
DRV
CSFB
VCC
C3D4
C5
MIN_TOFF
MIN_TON
LLD
+VOUTVBULK
FLYBACKCONTROLCIRCUITRY
M1
C1
OK1
R1
D3
C2
TR1
R3
M2
GNDC4
D5NCP4306
RM
IN_T
ON
RLL
D
R2
DRV
CSFB
VCC
C3D4
C5
VCC DRV
GND
CS
TRIG
+VOUT
Figure 3. Typical Application Example – DCM, CCM or QR Flyback
Converter with NCP4306 in TSOP6 (v Cxxxxxx)
VBULK
C1 R1
D3
C2
TR1
FLYBACKCONTROLCIRCUITRY
M1DRV
CSFB
VCC
C3
OK1
M2D4
R3
C4GND
C5
D5NCP4306
RM
IN_T
OF
F
RLL
D
+VOUT
VCC
MIN_TOFF
LLD
GND
CS
DRV
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Figure 4. Typical Application Example – Primary Side Flyback
Converter and NCP4306 in TSOP6
VBULK
C1R1
D3
C2
TR1
C6
M2
R5 C8
GND
+VOUT
C7
R3
R4
C5
C4R3
ZCD
C3
D4
VCC
DRV M1
R2
CSCOMP
VCC
MIN_TOFFGND
CS
DRV
MIN_TON
NCP4306
RM
IN_T
OF
F
RM
IN_T
ON
PRIMARYSIDE
FLYBACKCONTROLLER
PIN FUNCTION DESCRIPTION
Table 2. PIN FUNCTION DESCRIPTION
TSOP6Bxxxxxx
TSOP6Cxxxxxx
TSOP6Dxxxxxx
TSOP6Exxxxxx
TSOP6Fxxxxxx
TSOP6Gxxxxxx
SOIC8Axxxxxx
Pin Name Description
6 6 6 6 6 6 1 VCC Supply voltage pin
− 5 5 5 − 2 MIN_TOFF Adjust the minimum off timeperiod by
connecting resistor toground
5 − 4 − 5 − 3 MIN_TON Adjust the minimum on timeperiod by
connecting resistor toground
4 4 − − − 4 4 LLD This input modulates the driverclamp level and
/ or turns the driv-er off during light load conditions
− − − 4 4 5 5 TRIG / DIS Ultrafast turn−off input that can
beused to turn off the SR MOSFETin CCM applications in order
toimprove efficiency. Activatesdisable mode if pulled−up formore
than 100 μs
3 3 3 3 3 3 6 CS Current sense pin detects if thecurrent flows
through the SRMOSFET and / or its body diode
2 2 2 2 2 2 7 GND Ground connection for the SRMOSFET driver and
VCCdecoupling capacitor. Groundconnection for minimum tON andtOFF
adjust resistors, LLD andtrigger inputsGND pin should be wired
directlyto the SR MOSFET sourceterminal / soldering point
usingKelvin connection
1 1 1 1 1 1 8 DRV Driver output for the SR MOSFET
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Figure 5. Internal Circuit Architecture – NCP4306
Minimum ON timegenerator
MIN_TON
CSdetection
CS
MIN_TOFF
TRIG/DIS
DRV
VCC
GND
VCC managmentUVLO
DRIVER
VDD
LLDDisable detection
ELAPSED
EN
Minimum OFFtime generator
RESET
ELAPSED
Control logic
EN
DISABLE
Disable detection
DISABLE
DISABLETRIG
dV/dt
Exception timegenerator
ENELAPSED
10 μA VTRIG
DRVOUT
EXT_ADJ
INT_ADJ
EXT_ADJ
INT_ADJ
INT_ADJ
CS_ONCS_OFFCS_RESET
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ABSOLUTE MAXIMUM RATINGS
Table 3. ABSOLUTE MAXIMUM RATINGS
Rating Symbol Value Unit
Supply Voltage VCC −0.3 to 37.0 V
TRIG / DIS, MIN_TON, MIN_TOFF, LLD Input Voltage (Note 3) VTRIG
/ DIS, VMIN_TON, VMIN_TOFF, VLLD −0.3 to VCC V
Driver Output Voltage VDRV −0.3 to 17.0 V
Current Sense Input Voltage VCS −4 to 200 V
Current Sense Dynamic Input Voltage (tPW = 200 ns) VCS_DYN −10
to 200 V
MIN_TON, MIN_TOFF, LLD, TRIG Input Current IMIN_TON, IMIN_TOFF,
ILLD, ITRIG −10 to 10 mA
DRV Pin Current (tPW = 10 μs) IDRV_DYN −3 to 12 A
VCC Pin Current (tPW = 10 μs) IVCC_DYN 3 A
Junction to Air Thermal Resistance, 1 oz 1 in2 Copper
Area,SOIC8
RθJ−A_SOIC8 200 °C / W
Junction to Air Thermal Resistance, 1 oz 1 in2 Copper
AreaTSOP6
RθJ−A_TSOP6 250 °C / W
Maximum Junction Temperature TJMAX 150 °C
Storage Temperature TSTG −60 to 150 °C
ESD Capability, Human Body Model (except pin CS) (Note 1) ESDHBM
2000 V
ESD Capability, Human Body Model Pin CS ESDHBM 600 V
ESD Capability, Machine Model (Note 1) ESDMM 200 V
ESD Capability, Charged Device Model (Note 1) ESDCDM Class C1
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Stresses exceeding those listed in the Maximum Ratings table may
damage the device. If any of these limits are exceeded, device
functionalityshould not be assumed, damage may occur and
reliability may be affected.1. This device series contains ESD
protection and exceeds the following tests:
Except pin CS: Human Body Model 2000 V per JEDEC Standard
JESD22−A114E.All pins: Machine Model Method 200 V per JEDEC
Standard JESD22−A115−ACharged Machine Model per JEDEC Standard
JESD22−C101F
2. This device meets latchup tests defined by JEDEC Standard
JESD78D.3. If voltage higher than 22 V is connected to pin, pin
input current increases. Internal ESD clamp contains 24 V Zener
diode with 3 kΩ in series.
It is recommended to add serial resistance in case of higher
input voltage to limit input pin current.
Table 4. RECOMMENDED OPERATING CONDITION
Parameter Symbol Min Max Unit
Maximum Operating Voltage VCC 35 V
Operating Junction Temperature TJ −40 125 °C
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ELECTRICAL CHARACTERISTICS
Table 5. ELECTRICAL CHARACTERISTICS −40 ºC ≤ TJ ≤ 125 ºC; VCC =
12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kΩ or internally set
values; VLLD = 3.0 V or LLD internallydisabled; VTRIG / DIS = 0 V;
VCS = 4 V, unless otherwise noted. Typical values are at TJ = +25
ºC
Parameter Test Conditions Symbol Min Typ Max Unit
SUPPLY SECTION
VCC UVLO VCC rising VCCON 3.7 4.0 4.2 V
VCC falling VCCOFF 3.2 3.5 3.7
VCC UVLO Hysteresis VCCHYS 0.5 V
Start−up Delay VCC rising from 0 to VCCON + 1 V @ tr= 10 μs
tSTART_DEL 50 80 μs
Current Consumption,tMIN_TON = tMIN_TOFF = 1 μs, tLLD = 130
μs
CDRV = 0 nF,fCS = 100 kHz
xAxxxxx ICC 1.8 2.5 mA
xBxxxxx 1.7 2.4
CDRV = 1 nF,fCS = 100 kHz
xAxxxxx 2.8 4.0
xBxxxxx 2.1 3.4
CDRV = 10 nF,fCS = 100 kHz
xAxxxxx 12 15
xBxxxxx 6.7 9.0
Current Consumption ICC 1.4 2.2 mA
Current Consumption below UVLO VCC = VCCOFF – 0.1 V ICC_UVLO 35
60 μACurrent Consumption in Disable Mode t > tLLD , VLLD = 0.55
V ICC_DIS 60 100 μA
VTRIG / DIS = 5 V; VLLD = 0.55 V 60 100
t > tLLD, LLD set internally 37 80
VTRIG / DIS = 5 V, LLD set internally 37 80
DRIVER OUTPUT
Output Voltage Rise−Time CDRV = 10 nF, 10 % to 90 % VDRVMAX,VCS
= 4 to −1 V
tr 60 100 ns
Output Voltage Fall−Time CDRV = 10 nF, 90 % to 10 % VDRVMAX,VCS
= −1 to 4 V
tf 25 45 ns
Driver Source Resistance RDRV_SOURCE 2 Ω
Driver Sink Resistance RDRV_SINK 0.5 Ω
Output Peak Source Current IDRV_SOURCE 2 A
Output Peak Sink Current IDRV_SINK 7 A
Maximum Driver Pulse Length tDRV_ON_MAX 4 ms
Maximum Driver Output Voltage VCC = 35 V, CDRV > 1 nF, (ver.
xAxxxxx)
VDRVMAX 9 10 11 V
VCC = 35 V, CDRV > 1 nF, (ver. xBxxxxx)
4.5 5.0 5.5
Minimum Driver Output Voltage VCC = VCCOFF + 200 mV, (ver.
xAxxxxxx)
VDRVMIN 3.4 3.7 3.9 V
VCC = VCCOFF + 200 mV, (ver. xBxxxxxx)
3.4 3.7 3.9
CS INPUT
Total Propagation Delay From CS toDRV Output On
VCS goes down from 4 to −1 V,tf_CS
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Table 5. ELECTRICAL CHARACTERISTICS (continued)−40 ºC ≤ TJ ≤ 125
ºC; VCC = 12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kΩ or
internally set values; VLLD = 3.0 V or LLD internallydisabled;
VTRIG / DIS = 0 V; VCS = 4 V, unless otherwise noted. Typical
values are at TJ = +25 ºC
Parameter UnitMaxTypMinSymbolTest Conditions
CS INPUT
dV / dt Detector Low Threshold VCS_DVDT_L 0.5 V
dV / dt Detector Threshold (Note 4) ver. xxDxxxx tdV / dt 13 25
37 ns
TRIGGER DISABLE INPUT
Minimum Trigger Pulse Duration VTRIG / DIS = 5 V; Shorter pulses
maynot be proceeded
tTRIG_PW_MIN 10 ns
Trigger Threshold Voltage VTRIG_TH 1.6 2.0 2.2 V
Trigger to DRV Propagation Delay VTRIG / DIS goes from 0 to 5 V,
tr_TRIG / DIS
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Table 5. ELECTRICAL CHARACTERISTICS (continued)−40 ºC ≤ TJ ≤ 125
ºC; VCC = 12 V; CDRV = 0 nF; RMIN_TON = RMIN_TOFF = 10 kΩ or
internally set values; VLLD = 3.0 V or LLD internallydisabled;
VTRIG / DIS = 0 V; VCS = 4 V, unless otherwise noted. Typical
values are at TJ = +25 ºC
Parameter UnitMaxTypMinSymbolTest Conditions
LLD ADJUST
LLD Reduced Time Disable mode activated tLLD_RED 0.5 ×tLLD
μs
LLD Blanking Time tLLD_BLK 0.25 ×tLLD
μs
Disable Recovery Time tMIN_TOFF = 130 ns tLLD_DIS_REC 1.5 3.0
μs
EXCEPTION TIMER
Exception Time (ver. xxHxxxx) tEXC 4 ×tMIN_TON
μs
Exception Timer Ratio Accuracy RatioEXC −15 +15 %
4. Test signal:
VCS [V]
4.0
1.5
−1.0t [ns]
t dV/dt
VCS_DVDT_H
VCS_DVDT_L
Figure 6. Test Signal
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TYPICAL CHARACTERISTICS
3,3
3,4
3,5
3,6
3,7
3,8
3,9
4,0
4,1
4,2
−40 −20 0 20 40 60 80 100 120
VCC on
VCC off
Figure 7. VCCON and VCCOFF Levels
TJ[°C]
VC
C[V
]
0,0
0,2
0,4
0,6
0,8
1,0
1,2
1,4
1,6
0 5 10 15 20 25 30 35
TJ = 125 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ = −20 °CTJ
= −40 °C
0,0
0,5
1,0
1,5
2,0
2,5
3,0
3,5
0 5 10 15 20 25 30 35
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ
= −20 °CTJ = −40 °C
Figure 8. Current Consumption VCS = 4 V
VCC[V]
I CC
[mA
]
Figure 9. Current Consumption, fCS = 100 kHz, CDRV = 1 nF, Ver.
xAxxxxx
Figure 10. Current Consumption, fCS = 100 kHz, Ver. xAxxxxx
VCC[V]
I CC
[mA
]
0
2
4
6
8
10
12
14
−40 −20 0 20 40 60 80 100 120
CDRV = 0 nF
CDRV = 1 nF
CDRV = 10 nFI CC
[mA
]
TJ[°C]
0,0
0,5
1,0
1,5
2,0
2,5
0 5 10 15 20 25 30 35
Figure 11. Current Consumption, fCS = 100 kHz, Ver. xBxxxxx
0
1
2
3
4
5
6
7
8
9
10
−40 −20 0 20 40 60 80 100 120
Figure 12. Current Consumption, fCS = 100 kHz, CDRV = 1 nF, Ver.
xBxxxx
I CC
[mA
]
TJ[°C]
I CC
[mA
]
VCC[V]
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0
°C
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ
= −20 °CTJ = −40 °C
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0
°C
CDRV = 0 nF
CDRV = 1 nF
CDRV = 10 nF
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0
10
20
30
40
50
60
−40 −20 0 20 40 60 80 100 12020
30
40
50
60
70
80
90
100
−40 −20 0 20 40 60 80 100 120
Figure 13. Current Consumption below UVLO, VCC = VCCOFF − 0.1
V
TJ[°C]
I CC
_UV
LO[μ
A]
I CC
_DIS
[μA
]
TJ[°C]
Figure 14. Current Consumption in Disable ModeVCS = 4 V, t >
tLLD
20
30
40
50
60
70
80
90
100
−40 −20 0 20 40 60 80 100 1200
20
40
60
80
100
120
0 5 10 15 20 25 30 35
Figure 15. Current Consumption in Disable Mode,VTRIG/DIS = 5
V
Figure 16. Current Consumption in Disable Mode,VCS = 4 V, t >
tLLD
20
30
40
50
60
70
80
90
100
0 5 10 15 2025
30 35−1,0
−0,9
−0,8
−0,7
−0,6
−0,5
−0,4
−0,3
−0,2
−0,1
0,0
−1 −0,8 −0,6 −0,4 −0,2 0 0,2 0,4 0,6 0,8 1
Figure 17. Current Consumption in Disable Mode,VTRIG/DIS = 5
V
Figure 18. CS Input Current
I CC
_DIS
[μA
]
I CC
_DIS
[μA
]
I CC
_DIS
[μA
]
I CS[m
A]
TJ[°C] VCC[V]
VCS[V]VCC[V]
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °C
TJ = 25 °CTJ = 0 °CTJ = −20 °CTJ = −40 °C
TJ = 125 °CTJ = 105 °C
TJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ = −20 °CTJ = −40
°C
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ
= −20 °CTJ = −40 °C
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0
°C
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Figure 19. Supply Current vs. CS Voltage
−120
−110
−100
−90
−80
−70
−60
−50
−40
−40 −20 0 20 40 60 80 100
1200,00,20,40,60,81,01,21,41,61,82,02,2
−1,0 −0,8 −0,6 −0,4 −0,2 0,0 0,2 0,4 0,6 0,8 1,0
Figure 20. CS Turn−on Threshold
I CC
[mA
]
VCS[V]
VT
H_C
S_O
N[m
V]
TJ[°C]
−2,0
−1,5
−1,0
−0,5
0,0
0,5
1,0
−40 −20 0 20 40 60 80 100 1200,40
0,45
0,50
0,55
0,60
−40 −20 0 20 40 60 80 100 120
0
100
200
300
400
500
−40 −20 0 20 40 60 80 100 12010
15
20
25
30
35
40
45
50
55
60
−40 −20 0 20 40 60 80 100 120
Figure 21. CS turn−off Threshold Figure 22. CS Reset
Threshold
Figure 23. CS Input Leakage VCS = 200 V Figure 24. Propagation
Delay from CS to DRV Output On
TJ[°C]
TJ[°C]
TJ[°C]
TJ[°C]
VT
H_C
S_O
FF[m
V]
VT
H_C
S_R
ES
ET[V
]
I CS
_LE
AK
AG
E[n
A]
t PD
_ON
[ns]
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ
= −20 °CTJ = −40 °C
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Figure 25. Propagation Delay from CS to DRV Output Off
4
6
8
10
12
14
16
18
20
22
24
−40 −20 0 20 40 60 80 100 120
Figure 26. Trigger Pin Threshold
1,0
1,2
1,4
1,6
1,8
2,0
2,2
0 5 10 15 20 25 30 35
TJ[°C]
t PD
_OF
F[n
s]
VCC[V]
VT
RIG
_TH
[V]
1,7
1,8
1,9
2,0
2,1
2,2
2,3
−40 −20 0 20 40 60 80 100 1207
8
9
10
11
12
13
14
15
−40 −20 0 20 40 60 80 100 120
Figure 27. Trigger Pin Threshold Figure 28. Trigger Pin Pull
Down Current
3
5
7
9
11
13
15
17
−40 −20 0 20 40 60 80 100 120
TJ[°C] TJ[°C]
TJ[°C]
Figure 29. Propagation Delay from TRIG to DRV Output Off
Figure 30. Delay to Disable Mode, VTRIG/DIS = 5 V
VT
RIG
_TH
[V]
I TR
IG/D
IS[μ
A]
t PD
_TR
IG[n
s]
80
85
90
95
100
105
110
115
120
−40 −20 0 20 40 60 80 100 120
t DIS
_TIM
[μs]
TJ[°C]
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ
= −20 °CTJ = −40 °C
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0
°C
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Figure 31. Minimum on Time RMIN_TON = 10 k�
0,90
0,92
0,94
0,96
0,98
1,00
1,02
1,04
1,06
1,08
1,10
−40 −20 0 20 40 60 80 100 1204,5
4,6
4,7
4,8
4,9
5,0
5,1
5,2
5,3
5,4
5,5
−40 −20 0 20 40 60 80 100 120
Figure 32. Minimum on Time RMIN_TON = 50 k�
0,90
0,92
0,94
0,96
0,98
1,00
1,02
1,04
1,06
1,08
1,10
−40 −20 0 20 40 60 80 100 1204,4
4,5
4,6
4,7
4,8
4,9
5,0
5,1
5,2
5,3
5,4
−40 −20 0 20 40 60 80 100 120
Figure 33. Minimum on Time RMIN_TOFF = 10 k� Figure 34. Minimum
on Time RMIN_TOFF = 50 k�
−25,0
−20,0
−15,0
−10,0
−5,0
0,0
0 5 10 15 20 25 30 35
Figure 35. LLD Current, VLLD = 3.0 V Figure 36. LLD current,
VLLD = 2.5 V
TJ[°C]
t MIN
_TO
N[μ
s]t M
IN_T
OF
F[μ
s]I L
LD[μ
A]
TJ[°C]
t MIN
_TO
N[μ
s]
TJ[°C]TJ[°C]
VCC[V]
t MIN
_TO
FF[μ
s]
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ
= −20 °CTJ = −40 °C
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0
°C
−22,0
−21,5
−21,0
−20,5
−20,0
−19,5
−19,0
−18,5
−18,0
0 5 10 15 20 25 30 35
I LLD
[μA
]
VCC[V]
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °C
TJ = 25 °CTJ = 0 °CTJ = −20 °CTJ = −40 °C
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Figure 37. LLD Current Figure 38. LLD Time, VLLD = 1.82 V (or
Internal Option)
440
460
480
500
520
540
560
580
600
620
640
−40 −20 0 20 40 60 80 100 1200 0.5 1.0 1.5 2.0 2.5 3.0 3.5
4.0
−15.0
−16.0
−17.0−18.0−19.0
−20.0
−21.0−22.0
−23.0
−24.0
−25.0
VLLD[V]
I LLD
[μA
]
TJ[°C]
t LLD
[μs]
9,0
9,2
9,4
9,6
9,8
10,0
10,2
10,4
−40 −20 0 20 40 60 80 100 120
VCC = 12 V, CDRV = 0 nFVCC = 12 V, CDRV = 1 nFVCC = 12 V, CDRV =
10 nFVCC = 35 V, CDRV = 0 nFVCC = 35 V, CDRV = 1 nFVCC = 35 V, CDRV
= 10 nF
4,3
4,5
4,7
4,9
5,1
5,3
5,5
−40 −20 0 20 40 60 80 100 120
VD
RV[V
]
TJ[°C] TJ[°C]
VD
RV[V
]
13
18
23
28
33
38
−40 −20 0 20 40 60 80 100 120
Figure 39. Driver Output Voltage, Ver. xAxxxxx Figure 40. Driver
Output Voltage, Ver. xBxxxxx
3,4
3,6
3,8
4,0
4,2
4,4
4,6
−40 −20 0 20 40 60 80 100 120
Figure 41. dV/dt Detector Time Threshold, Ver. xxDxxxx
Figure 42. Exception Timer Ratio to tMIN_TON, Ver. xxHxxxx
TJ[°C] TJ[°C]
t dV
/dt[n
s]
Rat
ioE
XC
[−]
TJ = 125 °CTJ = 105 °CTJ = 85 °CTJ = 55 °CTJ = 25 °CTJ = 0 °CTJ
= −20 °CTJ = −40 °C
VCC = 12 V, CDRV = 0 nFVCC = 12 V, CDRV = 1 nFVCC = 12 V, CDRV =
10 nFVCC = 35 V, CDRV = 0 nFVCC = 35 V, CDRV = 1 nFVCC = 35 V, CDRV
= 10 nF
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GENERAL DESCRIPTION
The NCP4306 is designed to operate either as a standaloneIC or
as a companion IC to a primary side controller to helpachieve
efficient synchronous rectification in switch modepower supplies.
This controller features a high current gatedriver along with
high−speed logic circuitry to provideappropriately timed drive
signals to a synchronousrectification MOSFET. With its novel
architecture, theNCP4306 has enough versatility to keep the
synchronousrectification system efficient under any operating
mode.
The NCP4306 works from an available voltage with rangefrom 4.0 /
3.5 V to 35 V (typical). The wide VCC rangeallows direct connection
to the SMPS output voltage of mostadapters such as notebooks, cell
phone chargers and LCDTV adapters.
Precise turn−off threshold of the current sense
comparatortogether with an accurate offset current source allows
theuser to adjust for any required turn−off current threshold ofthe
SR MOSFET switch using a single resistor. Comparedto other SR
controllers that provide turn−off thresholds inthe range of −10 mV
to −5 mV, the NCP4306 offers aturn−off threshold of 0 mV. When
using a low RDS_ON SR(1 mΩ) MOSFET our competition, with a −10 mV
turn off,will turn off with 10 A still flowing through the SR
FET,while our 0 mV turn off turns off the FET at 0 A;significantly
reducing the turn−off current threshold andimproving efficiency.
Many of the competitor partsmaintain a drain source voltage across
the MOSFET causingthe SR MOSFET to operate in the linear region to
reduceturn−off time. Thanks to the 6 A sink current of theNCP4306
significantly reduces turn off time allowing for aminimal drain
source voltage to be utilized and efficiencymaximized.
To overcome false triggering issues after turn−on andturn−off
events, the NCP4306 provides adjustable minimum
on−time and off−time blanking periods. Blanking times canbe set
internally during production or adjustedindependently of IC VCC
using external resistors connectedto GND (internal or external
option depends on IC variant).If needed, externally set blanking
periods can be modulatedusing additional components.
An extremely fast turn−off comparator, implemented onthe current
sense pin, allows for NCP4306 implementationin CCM applications
without any additional components orexternal triggering.
An ultrafast trigger input offers the possibility to
furtherincrease efficiency of synchronous rectification
systemsoperated in CCM mode (for example, CCM flyback orforward).
The time delay from trigger input to driver turn offevent is
tPD_TRIG. Additionally, the trigger input can be usedto disable the
IC and activate a low consumption standbymode. This feature can be
used to decrease standbyconsumption of an SMPS. If the trigger
input is not wantedthan the trigger pin can be tied to GND.
An output driver features capability to keep SR
transistorturned−off even when there is no supply voltage for
theNCP4306. SR transistor drain voltage goes up and downduring SMPS
operation and this is transferred through draingate capacitance to
gate and may open transistor. TheNCP4306 keeps DRV pin pulled low
even without anysupply voltage and thanks to this the risk of
turned−on SRtransistor before enough VCC is applied to the NCP4306
iseliminated.
Finally, the NCP4306 features a Light Load Detectionfunction
that can be set internally or externally at LLD pinby resistor
connected to ground. This function detects lightload or no load
conditions and during them betweenconduction phases it decreases
current consumption. Thishelps to improve SMPS efficiency. If LLD
function is notneeded pin can be left open.
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SUPPLY SECTION
Supply voltage should be connected to VCC pin.Minimum voltage
for proper operation is 4.0 / 3.5 Vtypically and maximum level is
35 V. Decoupling capacitorbetween VCC and GND pin is needed for
proper operationand its recommended value is 1 μF. If IC is
supplied fromSMPS output voltage, few ohm resistor is
recommendedbetween SMPS output voltage and VCC pin. Resistor taskis
to divide decoupling cap from output to avoid closing HFcurrents
through NCP4306 decoupling cap, because thesecurrents may causes
drops at GND connection that affectsSR transistor sensing and
incorrect SR transistor turn−off.
SR transistor is usually used in low side configuration(placed
in return path), but it may be also used in high sideconfiguration
(placed in positive line). It is not possible touse SMPS VOUT for
SR supply in high side configuration soit is needed to provide
supply differently. One possibility isto use auxiliary winding as
shown in Figure 43. Voltage fromauxiliary winding is rectified,
filtered and use as supplyvoltage.
Figure 43. High Side Configuration Supplied from Auxiliary
Winding
C1
GND
+VOUT
D5OK1
R3
M2
M1
R1CSFB
VCC
D1
DRVD2
C3
C1 R2 C2
D3
C5C4R6
D4TR1
R7R8R9
NCP4306
FLYBACKCONTROLCIRCUITRY
If auxiliary winding is not acceptable, transformerforward
voltage can be used as supply source (Figure 44).Forward voltage is
regulated by simple voltage regulator to
fit NCP4306 VCC restriction. Penalty for this solution
isslightly lower efficiency.
Figure 44. High Side Configuration Supplied from Transformer
Forward Voltage
VBULK
+VOUT
C1 R2 C2
FB CS
DRV D2
C3D1
D3
M1
R1
R3
OK1 D6 GND
C6
M2
D4 D5
R6 R7
C4 C5
R8R9
FLYBACKCONTROLL
NCP4306
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Auxiliary winding or forward voltage can be used assupply source
also for low side configuration if VOUT is nothigh enough (Figure
45). Do not focus just on SR controller
UVLO, but also on SR transistor characteristics. Sometransistors
may be not turned−on enough even at 5 V so inthese case SR
controller supply voltage should be increased.
Figure 45. Low Side Configuration Supplied from Transformer
Forward Voltage for Low VOUT SMPS
C6
R2R5
C5
C4R4
ZCD
R3C1
R1C2
D3
D4
C3
VCC
DRV M1
CSCOMP
VBULK
D4 R6
R7
M2 D5R8
C7
C8
+VOUT < 5V
GNDC9
NCP4306
PRIMARYSIDE
FLYBACKCONTROLLER
Current Sense InputFigure 46 shows the internal connection of
the CS
circuitry on the current sense input. When the voltage on
thesecondary winding of the SMPS reverses, the body diode ofM1
starts to conduct current and the voltage of M1’s draindrops
approximately to −1 V. Once the voltage on the CS pinis lower than
VTH_CS_ON threshold, M1 is turned−on.
Because of parasitic impedances, significant ringing canoccur in
the application. To overcome false sudden turn−offdue to mentioned
ringing, the minimum conduction time ofthe SR MOSFET is activated.
Minimum conduction timecan be adjusted using the RMIN_TON resistor
or can bechosen from internal fixed values.
Figure 46. Current Sensing Circuitry Functionality
SR MOSFET
+ VOUT
M1
To Internal logicCS_RESET
High dV / dt
CS_OFF
CS_ON
VTH_CS_RESET
dV / dtDetector
VTH_CS_ON
+
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The SR MOSFET is turned−off as soon as the voltage onthe CS pin
is higher than VTH_CS_OFF (typically −0.5 mV).For the same ringing
reason, a minimum off−time timer isasserted once the VCS goes above
VTH_CS_RESET. Theminimum off−time can be externally adjusted
usingRMIN_TOFF resistor or can be chosen from internally
fixedvalues (depends on version). The minimum off−timegenerator can
be re−triggered by MIN_TOFF resetcomparator if some spurious
ringing occurs on the CS inputafter SR MOSFET turn−off event. This
feature significantlysimplifies SR system implementation in flyback
converters.
In an LLC converter the SR MOSFET M1 channelconducts while
secondary side current is decreasing (refer toFigure 47). Therefore
the turn−off current depends onMOSFET RDSON. The −0.5 mV threshold
provides anoptimum switching period usage while keeping enough
timemargin for the gate turn−off. To ensure proper switching,
themin_tOFF timer is reset, when the VDS of the MOSFET ringsand
falls down past the VTH_CS_RESET. The minimumoff−time needs to
expire before another drive pulse can beinitiated. Minimum off−time
timer is started again whenVDS rises above VTH_CS_RESET.
Figure 47. CS Input Comparators Thresholds and Blanking Periods
Timing in LLC
VDS = VCS
ISEC
VTH_CS_RESET
VTH_CS_OFF
VDRV
Min ON−time
Min OFF−time
VTH_CS_ON
Turn on delay Turn off delay
tMIN_TON
tMIN_TOFF
Min tOFF timer wasstopped here because of
VCS
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Figure 48. CS Input Comparators Thresholds and Blanking Periods
Timing in Flyback
VTH_CS_RESET
VTH_CS_OFF
VDRV
Min ON−time
Min OFF−time
VTH_CS_ON
Turn on delay Turn off delay
Min tOFF timer wasstopped here because of
VCS
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Figure 49. SR System Connection Including MOSFET and Layout
Parasitic Inductances in a) LLC and b) FlybackApplication
VL_LAYOUT VL_DRAIN VRDS_ON VL_SOURCE VL_LAYOUT
ISECVDS
LLAYOUT
To VCC
LSOURCERDS_ONLDRAIN
MOSFET equivalent circuit
NCP4306
100 nF 1 μFDecouplingCapacitors
VL_LAYOUT VL_DRAIN VRDS_ON VL_SOURCE VL_LAYOUT
ISECVDS
LLAYOUTLSOURCERDS_ONLDRAIN
MOSFET equivalent circuit
NCP4306
DecouplingCapacitors
LLAYOUT
To VCC
VL_LAYOUTVL_DRAIN VRDS_ON VL_SOURCE VL_LAYOUT
ISEC
VDS
LLAYOUTLSOURCERDS_ONLDRAIN
MOSFET equivalent circuit
NCP4306
DecouplingCapacitors
LLAYOUT
1 μF100 nF
a)
b)
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Figure 50. Waveforms from SR System Implemented in a) LLC and b)
Flyback Application and Using MOSFET inTO220 Package With Long
Leads – SR MOSFET channel Conduction Time is Reduced
a) b)
Note that the efficiency impact caused by the error voltagedue
to the parasitic inductance increases with lowerMOSFETs RDS_ON and
/ or higher operating frequency.
It is thus beneficial to minimize SR MOSFET packageleads length
in order to maximize application efficiency. Theoptimum solution
for applications with high secondary
current Δi / Δt and high operating frequency is to uselead−less
SR MOSFET i.e. SR MOSFET in SMT package.The parasitic inductance of
a SMT package is negligiblecausing insignificant CS turn−off
threshold shift and thusminimum impact to efficiency (refer to
Figure 51).
Figure 51. Waveforms from SR System Implemented in a) LLC b)
Flyback Application and Using MOSFET in SMTPackage with Minimized
Parasitic Inductance – SR MOSFET Channel Conduction Time is
Optimized
a) b)
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It can be deduced from the above paragraphs on theinduced error
voltage and parameter tables that turn−offthreshold precision is
quite critical. If we consider a SRMOSFET with RDS_ON of 1 mΩ, the
1 mV error voltage onthe CS pin results in a 1 A turn−off current
thresholddifference; thus the PCB layout is very critical
whenimplementing the SR system. Note that the CS turn−offcomparator
is referred to the GND pin. Any parasiticimpedance (resistive or
inductive – even on the magnitudeof mΩ and nH values) can cause a
high error voltage that isthen evaluated by the CS comparator.
Ideally the CSturn–off comparator should detect voltage that is
caused bysecondary current directly on the SR MOSFET channel
resistance. In reality there will be small parasitic impedanceon
the CS path due to the bonding wires, leads and soldering.To assure
the best efficiency results, a Kelvin connection ofthe SR
controller to the power circuitry should beimplemented. The GND pin
should be connected to the SRMOSFET source soldering point and
current sense pinshould be connected to the SR MOSFET drain
solderingpoint − refer to Figure 49. Using a Kelvin connection
willavoid any impact of PCB layout parasitic elements on the
SRcontroller functionality; SR MOSFET parasitic elementswill still
play a role in attaining an error voltage. Figure 52and Figure 53
show examples of SR system layouts usingMOSFETs in D2PAK and SO8FL
packages.
Figure 52. Recommended Layout When Using SRMOSFET in D2PAK
Package
Figure 53. Recommended Layout When Using SRMOSFET in SMT Package
SO8 FL
Trigger / Disable inputThe NCP4306 features an ultrafast trigger
input that
exhibits a maximum of tPD_TRIG delay from its activation tothe
start of SR MOSFET turn−off of process. This input canbe used in
applications operated in deep ContinuesConduction Mode (CCM) to
further increase efficiency and/ or to activate disable mode of the
SR driver in which theconsumption of the NCP4306 is reduced to
maximum ofICC_DIS.
NCP4306 is capable to turn−off the SR MOSFET reliablyin CCM
applications just based on CS pin information only,without using
the trigger input. However, natural delay ofthe ZCD comparator and
DRV turn−off delay increaseoverlap between primary and secondary
MOSFETsswitching (also known as cross conduction). If one wants
toachieve absolutely maximum efficiency with deep CCMapplications,
then the trigger signal coming from theprimary side should be
applied to the trigger pin. It is goodto set trigger pulse in way
there is just short overlap betweenprimary and secondary switches.
Short overlap is usually
advantageous than leaving end of conduction phase on bodydiode.
Reason is body diode has usually longer recoverytime and resulting
overlap time (simultaneously conductionprimary and secondary side
switches) is longer. There areseveral possibilities for
transferring the trigger signal fromthe primary to the secondary
side – refer to Figure 68 andFigure 69.
The trigger signal is blanked for tTRIG_BLANK after theDRV
turn−on process has begun. The blanking technique isused to
increase trigger input noise immunity against theparasitic ringing
that is present during the turn on processdue to the SMPS layout.
The trigger input is supersedes theCS input except trigger blanking
period. TRIG / DIS signalturns the SR MOSFET off or prohibits its
turn−on when theTRIG / DIS pin is pulled above VTRIG_TH.
The SR controller enters disable mode when the triggerpin is
pulled−up for more than tDIS_TIM. In disable mode theIC consumption
is significantly reduced. To recover fromdisable mode and enter
normal operation, the TRIG / DISpin has to be pulled low at least
for tDIS_END.
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VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
t1 t2 t3 t4 t5 t6 t7 t8 t9t
Figure 54. Trigger Input Functionality Waveforms Using the
Trigger to Turn−off and Block the DRV Signal
Figure 54 shows basic TRIG / DIS input functionality. Att1 the
TRIG / DIS pin is pulled low to enter into normaloperation. At t2
the CS pin is dropped below theVTH_CS_ON, signaling to the NCP4306
to start to turn the SRMOSFET on. At t3 the NCP4306 begins to drive
theMOSFET. At t4, the SR MOSFET is conducting and theTRIG / DIS pin
is pulled high. This high signal on the TRIG
/ DIS pin almost immediately turns off the drive to the
SRMOSFET, turning off the MOSFET. The DRV is notturned−on in other
case (t6) because the trigger pin is highin the time when CS pin
signal crosses turn−on threshold.This figure clearly shows that the
DRV can be asserted onlyon falling edge of the CS pin signal in
case the trigger inputis at low level (t2).
Figure 55. Trigger Input Functionality Waveforms – Trigger
Blanking
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
t1 t2 t3t
TRIG / DIS blank
Min ON−time
tTRIGBLANK
In Figure 55 above, at time t1 the CS pin falls below
theVTH_CS_ON while the Trigger is low setting in motion theDRV
signal that appears at t2. At time t2 the DRV signal andTrigger
blanking clock begin. TRIG / DIS signal goes high
shortly after time t2. Due to the Trigger blanking
clock(tTRIG_BLANK) the Trigger’s high signal does not affect theDRV
signal until the tTRIG_BLANK timer has expired. Attime t3 the TRIG
/ DIS signal is reevaluated and the DRV
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signal is turned off. The TRIG / DIS input is blanked
fortTRIG_BLANK after DRV set signal to avoid undesirablebehavior
during SR MOSFET turn−on event. The blankingtime in combination
with high threshold voltage(VTRIG_TH) prevent triggering on ringing
and spikes that arepresent on the TRIG / DIS input pin during the
SR MOSFET
turn−on process. Controller’s response to the narrow pulseon the
TRIG / DIS pin is depicted in Figure 55 – this shorttrigger pulse
enables to turn the DRV on for tTRIG_BLANK.Note that this case is
valid only if device not entered disablemode before.
Figure 56. Trigger Input Functionality Waveforms – Trigger
Blanking Acts Like a Filter
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
TRIG / DIS blank
MIN ON−TIME
VTRIG_BLANK
t1 t2 t3 t4 t5 t6t0 t
Figure 56 above shows almost the same situation as inFigure 55
with one main exception; the TRIG / DIS signalwas not high after
trigger blanking timer expired so the DRVsignal remains high. The
advantage of the trigger blankingtime during DRV turn−on is evident
from Figure 56 since it
acts like a filter on the TRIG / DIS pin. Rising edge of theDRV
signal may cause spikes on the trigger input. If it wasn’tfor the
TRIG / DIS blanking these spikes, in combinationwith ultra−fast
performance of the trigger logic, could turnthe SR MOSFET off in an
inappropriate time.
Figure 57. Trigger Input Functionality Waveforms – Trigger over
Ride, CS Turn Off and Min On−time
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min ON−time
t1 t2t3 t4 t5 t6t0 tt7 t8
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Figure 57 depicts all possible driver turn−off events indetails
when correct VCC is applied. Controller driver isdisabled based on
TRIG / DIS input signal in time t2; theTRIG / DIS input overrides
the minimum on−time period.
Driver is turned−off according to the CS (VDS) signal (t5marker)
and when minimum on−time period elapsedalready. TRIG / DIS signal
needs to be low during this event.
If the CS (VDS) voltage reaches VTH_CS_OFF thresholdbefore
minimum on−time period ends (t7) and the TRIG /DIS pin is low the
DRV is turned−off on the falling edge ofthe minimum on−time period
(t8 time marker in Figure 57).This demonstrates the fact that the
Trigger over rides theminimum on−time. Minimum on−time has higher
prioritythan the CS signal.
In Figure 58 the TRIG / DIS input is low the whole timeand the
DRV pulses are purely a function of the CS signaland the minimum
on−time. The first DRV pulse terminatedbased on the CS signal and
another two DRV pulses areprolonged till the minimum on−time period
end despite theCS signal crosses the VTH_CS_OFF threshold
earlier.
If a minimum on−time is too long the situation that occursafter
time marker t6 Figure 58 can occur, is not correct andshould be
avoided. The minimum tON period should beselected shorter to
overcome situation that the SR MOSFETis turned−on for too long
time. The secondary current thenchanges direction and energy flows
back to the transformerthat result in reduced application
efficiency and also inexcessive ringing on the primary and
secondary MOSFETs.
Figure 58. Minimum On−Time Priority
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min ON−time
t1 t2 t3 t4 t5 t6t0 tt7 t8 t9
Figure 59 shows IC behavior in case the trigger signalfeatures
two pulses during one cycle of the VDS (CS) signal.The TRIG / DIS
goes low enables the DRV just before timet1 and DRV turns−on
because the VDS voltage drops underVTH_CS_ON threshold voltage. The
TRIG / DIS signaldisables driver at time t2. The TRIG / DIS drops
down toLOW level in time t3, but IC waits for complete
minimumoff−time. Minimum off−time execution is blocked until CS
pin voltage goes above VTH_CS_RESET threshold. Next cyclestarts
in time t6. The TRIG / DIS goes low and enables theDRV before VDS
drops below VTH_CS_ON threshold voltagethus the DRV turns−on in
time t6. The TRIG / DIS signalrises up to HIGH level at time t7,
consequently DRVturns−off and IC waits for high CS voltage to start
minimumoff−time execution.
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Figure 59. TRIG / DIS Input Functionality Waveforms – Two Pulses
at One Cycle
VDS = VCS
VTH_CS_RESET
VTH_CS_OFF
VTH_CS_ON
VTRIG/DIS
VDRV
Min OFF−time
Min ON−time
t1 t2 t3 t4 t5 t6t0t
t7 t8 t9 t10
Figure 60. Trigger Input Functionality Waveforms – Disable Mode
Activation
Powerconsumption
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min ON−time
t1 t2 t3 t4t0t
tDIS_TIM
In Figure 60 above, at t2 the CS pin rises to VTH_CS_OFFand the
SR MOSFET is turned−off. At t3 the TRIG / DISsignal is held high
for more than tDIS_TIM. NCP4306 entersdisable mode after tDIS_TIM.
Driver output is disabled indisable mode. The DRV stays low
(disabled) during
transition to disable mode. Figure 61 shows disable
modetransition 2nd case – i.e. when trigger rising edge comesduring
the trigger blank period. Figure 62 shows enteringinto disable mode
and back to normal sequences.
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Figure 61. Trigger Input Functionality Waveforms – Disable Mode
Clock Initiation
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min ON−time
t1 t2 t3t0 t
Powerconsumption
tTRIGBLANK
tDIS_TIM
Figure 62. Trigger Input Functionality Waveforms – Disable and
Normal Modes
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min OFF−time
t1 t2 t3t0t
Powerconsumption
tDIS_TIM
tDIS_REC
Disable Mode
t4
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Figure 63 and Figure 64 shows exit from disable mode indetail.
NCP4306 requires time up to tDIS_REC to recover allinternal
circuitry to normal operation mode whenrecovering from disable
mode. The driver is then enabledafter complete tMIN_TOFF period
when CS (VDS) voltage isover VTH_CS_RESET threshold. Driver
turns−on in the next
cycle on CS (VDS) falling edge signal only (t5 − Figure 63).The
DRV stays low during recovery time period. TRIG / DISinput has to
be low at least for tDIS_END time to end disablemode and start with
recovery. Trigger can go back high aftertDIS_END without recovery
interruption.
Figure 63. Trigger Input Functionality Waveforms – Exit from
Disable Mode before the Falling Edge of the CS Signal
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min ON−time
t1 t2 t3t0
t
Powerconsumption
t4 t5 t6 t7 t8
Dis
able
Mod
e
Normal Modetime
Rec Waits forcompletetMIN_TOFF
Note: Rec Time = Recovery Time
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Figure 64. Trigger Input Functionality Waveforms
Disable Mode Normal Modetime
Rec Waits forcompletetMIN_TOFF
tDIS_END
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min OFF−time
t1 t2t0
Powerconsumption
t3 t5t4t
Note: Rec Time = Recovery Time
Figure 65. Trigger Input Functionality Waveforms
Disable ModeNormal ModeRecovery
tMIN_TOFF
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min OFF−time
t1 t2t0t
Powerconsumption
t3 t5t4t
t DIS
_RE
C
t6 t7 t8
Waits forcompletetMIN_TOFF
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Figure 65 shows detail IC behavior after disable mode isended.
The trigger pin voltage goes low at t1 and aftertDIS_REC IC leaves
disable mode (t2). Time interval betweent2 and t3 is too short for
complete minimum off−time so
normal mode doesn’t start. VDS voltage goes high again attime t4
and this event starts new minimum off−time timerexecution. Next VDS
falling edge below VTH_CS_ON levelactivates driver.
Figure 66. Trigger Input Functionality Waveforms
VDS = VCS
VTH_CS_ON
VTRIG / DIS
VDRV
VTH_CS_OFF
VTH_CS_RESET
Min OFF−time
t1 t2t0
Powerconsumption
t3 t5t4t
t6 t7 t8
Dis
able
Mod
e
Normal ModeRecovery
Waits forcompletetMIN_TOFF
tMIN_TOFF
tDIS_REC
Different situation of leaving from disable mode is shownat
Figure 66. Minimum off−time execution starts at time t2,but before
time elapses VDS voltage falls to negativevoltage. This interrupts
minimum off−time execution and
the IC waits to another time when VDS voltage is positiveand
then is again started the minimum off−time timer. TheIC returns
into normal mode after whole minimum off−timeelapses.
Figure 67. NCP4306 Operation after Start−Up Event
VDS = VCS
VTH_CS_ON
VCCON
VCC
VTH_CS_OFF
VTH_CS_RESET
Min ON−timetMIN_TON
VDRV
Min OFF−time
tMIN_TOFFtMIN_TOFFNot CompletetMIN_TOFF −
IC is not activated
CompletetMIN_TOFF − activates IC
tMIN_TOFF − is stopped due toVDS drops belowVTH_CS_RESET
t1
t
t14t3t2 t4
t5 t7t6 t8
t9 t10 t11 t13t12
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Start−up event waveforms are shown at Figure 67. Astart−up event
is very similar to an exit from disable modeevent. The IC waits for
a complete minimum off−time event(CS pin voltage is higher than
VTH_CS_RESET) until drivepulses can continue. Figure 67 shows how
the minimumoff−time timer is reset when CS voltage is
oscillatingthrough VTH_CS_RESET level. The NCP4306 startsoperation
at time t1 (time t1 can be seen as a wake−up eventfrom the disable
mode through TRIG / DIS or LLD pin).Internal logic waits for one
complete minimum off−timeperiod to expire before the NCP4306 can
activate the driverafter a start−up or wake−up event. The minimum
off−timetimer starts to run at time t1, because VCS is higher
than
VTH_CS_RESET. The timer is then reset, before its setminimum
off−time period expires, at time t2 thanks to CSvoltage lower than
VTH_CS_RESET threshold. Theaforementioned reset situation can be
seen again at time t3,t4, t5 and t6. A complete minimum off−time
period elapsesbetween times t7 and t8 allowing the IC to activate a
driveroutput after time t8.
Optional primary triggering techniques for CCM
flybackapplication are shown in Figure 68 and Figure 69. NCP4306can
operate properly without triggering in CCM, but use oftriggering
can reduce the commutation losses and the SRMOSFET drain voltage
spike, which results in improvedefficiency in CCM.
Figure 68. Optional Primary Triggering in Deep CCM Application
Using Auxiliary Winding
VBULK
VCC
C1 R3
D2
C3
TR1
D3
C2D1
DRV
CSFB
R2
R1
M1
M2
R4C6
C4
D4
GND
R5R6R7
OK1 C5
D5 R7
NCP4306
+ VOUT
FLYBACKCONTROLCIRCUITRY
The application shown in Figure 68 is simplest and themost cost
effective solution for primary SR triggering. Thismethod uses
auxiliary winding made of triple insulated wireplaced close to the
primary winding section. This auxiliarywinding provides information
about primary turn−on eventto the SR controller before the
secondary winding reverses.
This is possible thanks to the leakage between primary
andsecondary windings that creates natural delay in energytransfer.
This technique provides approximately 0.5%efficiency improvement
when the application is operated indeep CCM and a transformer that
has a leakage of 1% ofprimary inductance is used.
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Figure 69. Optional Primary Triggering in Deep CCM Application
Using Trigger Transformer
VBULK
VCC
C1R2
D2
C4
TR1
D3
C3D1
DRV
CSFB
R3
R1
M1
M2
R4
C7
C6
D4
GND
R5R6R7
OK1
C5
D5 R8
NCP4306
+ VOUT
FLYBACKCONTROLCIRCUITRY
TR2
Application from Figure 69 uses an ultra−small
triggertransformer to transfer primary turn−on information
directlyfrom the primary controller driver pin to the SR
controllertrigger input. Because the trigger input is rising
edgesensitive, it is not necessary to transmit the entire
primarydriver pulse to the secondary. The coupling capacitor C5
isused to allow the trigger transformer’s core to reset and alsoto
prepare a needle pulse (a pulse with width shorter than100 ns) to
be transmitted to the NCP4306 TRIG / DIS input.The advantage of
needle trigger pulse usage is that therequired volt−second product
of the pulse transformer isvery low and that allows the designer to
use very small andcheap magnetic. The trigger transformer can even
beprepared on a small toroidal ferrite core with outer diameterof 4
mm and four turns for primary and secondary windings
to assure LPRIMARY = LSECONDARY > 10 μH. Proper
safetyinsulation between primary and secondary sides can beeasily
assured by using triple insulated wire for one or,better, both
windings.
This primary triggering technique providesapproximately 0.5%
efficiency improvement when theapplication is operated in deep CCM
and transformer withleakage of 1% of primary inductance is
used.
It is also possible to use capacitive coupling (useadditional
capacitor with safety insulation) between theprimary and secondary
to transmit the trigger signal. We donot recommend this technique
as the parasitic capacitivecurrents between primary and secondary
may affect thetrigger signal and thus overall system
functionality.
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Minimum tON and tOFF adjustmentThe NCP4306 offers fixed or an
adjustable minimum
on−time and off−time blanking periods (depends on ICversion)
that ease the implementation of a synchronousrectification system
in any SMPS topology. These timersavoid false triggering on the CS
input after the MOSFET isturned on or off.
Fixed versions are defined internally and can’t bemodified later
or changed during operation.
The adjustment of minimum tON and tOFF periods aredone based on
an internal timing capacitance and externalresistors connected to
the GND pin – refer to Figure 70 fora better understanding.
Figure 70. Internal Connection of the MIN_TON Generator (the
MIN_TOFF Works in the Same Way)
VDD
VREF
tMIN_TON
I R_M
IN_T
ON
CtIR_MIN_TON
RMIN_TON
MIN_TON
GND
To Internal Logic
NCP4306
DischargeSwitch
Current through the MIN_TON adjust resistor can becalculated
as:
IR_MIN_TON �Vref
RMIN_TON (eq. 1)
If the internal current mirror creates the same currentthrough
RMIN_TON as used the internal timing capacitor (Ct)charging, then
the minimum on−time duration can becalculated using this
equation.
tMIN_ON � Vt �Vref
IR_MIN_TON� Ct �
VrefVref
RMIN_TON
� Ct � RMIN_TON (eq. 2)
The internal capacitor size would be too large ifIR_MIN_TON was
used. The internal current mirror uses aproportional current, given
by the internal current mirrorratio. Note that the internal timing
comparator delay affects
the accuracy of equations 7 and 8 when MIN_TON orMIN_TOFF times
are selected near to their minimumpossible values. Please refer to
Figure 71 and Figure 72 formeasured minimum on and off time
charts.
Figure 71. MIN_TON Adjust Characteristic Figure 72. MIN_TOFF
Adjust Characteristic
0,0
0,5
1,0
1,5
2,0
2,5
3,0
3,5
4,0
4,5
5,0
0 5 10 15 20 25 30 35 40 45 50 0,0
0,5
1,0
1,5
2,0
2,5
3,0
3,5
4,0
4,5
5,0
0 5 10 15 20 25 30 35 40 45 50
t MIN
_TO
N [μ
s]
RMIN_TON [kΩ]
t MIN
_TO
FF [μ
s]
RMIN_TOFF [kΩ]
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The absolute minimum tON duration is internally clampedto 55 ns
and minimum tOFF duration to 70 ns in order toprevent any potential
issues with the minimum tON and / ortOFF input being shorted to
GND.
The NCP4306 features dedicated anti−ringing protectionsystem
that is implemented with a minimum tOFF blankgenerator. The minimum
off−time one−shoot generator isrestarted in the case when the CS
pin voltage crossesVTH_CS_RESET threshold and MIN_TOFF period is
active.
The total off−time blanking period is prolonged due to
theringing in the application (refer to Figure 47).
Some applications may require adaptive minimum on andoff time
blanking periods. It is possible to modulate blankingperiods by
using an external NPN transistor – refer to Figure73. The
modulation signal can be derived based on the loadcurrent, feedback
regulator voltage or other applicationparameter.
Figure 73. Possible Connection for MIN_TON and MIN_TOFF
Modulation
VDD
VREFtMIN_TON
I R_M
IN_T
ON
RM
IN_T
ON
_2
MIN_TON modulation InputGND
To Internal Logic
NCP4306
DischargeSwitch
RM
IN_T
ON
IR_MIN_TONMIN_TON
ModulationCurrent
dV / dt Detection – Flyback featureThe NCP4306 includes optional
feature for flyback type
converters, which operates with shorter primary on−timethan
ringing period after demagnetization phase duringmedium / high
loads. These applications are for exampleUSB−PD or Quick Charge
adapters. Difficulty with thissituation is that minimum off−time
doesn’t elapse before
primary side switch is turned on and off again so SRcontroller
doesn’t turn on SR mosfet. Whole secondary sidecurrent flows
through body diode that makes power loss.Figure 74 shows situation
without dV / dt detection. Herecan be seen that without detection
next conduction cyclemay be not taken through activated SR
transistor. Reason isnot elapsed minimum off−time blanking
interval.
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Figure 74. Situation without dV / dt Detection Feature
VTH_CS_RESETVTH_CS_OFF
VDRV
Min OFF−time
VTH_CS_ON
Min ON−time
Primary on−time is very short (shorter than ringing period) for
low VOUT
tMIN_TOFF has to be set tolonger time length of ringing
period tMIN_TOFF
Turn−off delayTurn−on delay
tMIN_TON
tMIN_TOFFt MIN
_TO
FF
Driver is not turned−on becausetMIN_TOFF doesn’t elapse
tMIN_TOFF timer is stopped herebecause of VCS<
VTH_CS_RESET
VDS = VCS
ISEC
Figure 75 shows how system with activated dV / dtdetection
behaves. Min_toff blanking interval is also resetduring voltage
drops at CS pin, but if high negative dV / dtoccurs at CS pin,
min_toff interval is shorted and SRcontroller is ready to detect CS
voltage lower than
VCS_TH_ON and turn SR transistor on. Negative dV / dt at CSpin
after primary switch is turned off is high in compare toslope that
comes during ringing after demagnetization.Thanks to this we can
safely detect end of primary on−timefrom ringing.
Figure 75. Situation with Enabled dV / dt Detection
VTH_CS_RESET
VTH_CS_OFF
Min OFF−time
VTH_CS_ON
Min ON−time
Turn−on delay Turn−off delay
Negative dV / dtdetector at CS pin
VDRV
tMIN_TOFF
tMIN_TOFFtMIN_TOFF
CS voltage drops belowVTH_CS_ON but at timewhen min tOFF
doesn’t
elapse and just low dV / dtis detected so DRV is not
activated
tMIN_TOFF timer is stopped herebecause of VCS<
VTH_CS_RESET
tMIN_TOFF timer doesn’telapse but high dV / dt is
detected so DRV is enable
tMIN_TOFF has to be setto longer time than length
of ringing period
VDS = VCS
ISEC
tMIN_TON
Exception Timer – LLC featureException timer is special feature
for LLC type SMPS. It
is mainly targeted to operation under light / medium load,where
secondary side SMPS current shape is not sine, but itcontains part
of capacitive peak optionally with no current
part followed by distorted sine. Examples of current shapeis
shown in Figure 76. This figure shows different currentshapes at
different loading. Lower loading makes shapemore distorted from
ideal sine.
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Figure 76. Current Shapes in LLC Examples
t
ISEC
Problematic shapes may cause prematurely SR transistorturn−off,
because CS voltage may get to zero or to positivevoltage (due to
low current or high dI / dt and parasiticinductance). Sensed
voltage drop can be seen in Figure 77.This situation is valid for
SR mosfet with RDSON = 1 mΩ andwith package (SMT) parasitic
inductance LPACPAR =0.5 nH. There can be seen that SR transistor
should be turnedoff in time between 0.4 to 1.5 μs, because CS
voltage is
above VTH_CS_OFF threshold. Turn−off process can bemasked by
min_ton blanking interval, but in this case isneeded to set it at
least to 1.5 μs that can make issue duringvery light load where
current flows just short time and longmin_ton may cause reverse
current from output capacitorsback to transformer and may change
soft switchingcondition to hard switching at primary side.
-10
-8
-6
-4
-2
2
0
0 1 2 3 4 5 6
V DS[m
V]
7
0,00
0,50
1,00
1,50
2,00
3,00
2,50
0 1 2 3 4 5 6
ISE
C[A
]
7
Drop just atRDSon
Drop at RDSonand parasiticinductance
VTH_CS_OFF
Figure 77. Sensed Voltage Drop at SR Transistor in LLC during
Light / Medium Load
t[μs] t[μs]
To early SR transistor turn−off is not issue just fromefficiency
point of view, but also from system stability pointof view. When
load is decreased, feedback loop asks primaryside for lower power
that changes secondary side currentshape and SR driver can be
turned off shortly after min_toffelapses. This causes lower
efficiency transfer to secondaryside and output voltage starts to
decrease. Feedback loop
asks for more power, secondary current shape changes andSR
driver starts to conduct whole period again that improvesenergy
transfer efficiency and output voltage starts toincrease. This has
to be again regulated by feedback loop andeverything starts from
begin and make SMPS oscillationsthat can be accompany with audible
noise.
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Figure 78. LLC System Oscillation due to Short SR Transistor
Conduction
ISEC
VDS1
VDRV1
MIN_TON Bodydiode conducts
VDS2
VDRV2
MIN_TON
Bodydiode conducts
Bodydiode conducts Bodydiode conducts
CS voltage goes above 0 V after
min_ton = SR is turned off
Regulation loop increases transferred power because thanks to
low
SR conduction angle lot of power is lose at body diode
Regulation loop decreases transferred power because thanks to
SR
it is too much voltage at the output
Operation of new feature is shown in Figure 79. Currentshape
makes drop at SR transistor with 1 mΩ and 0.5 nHshown as VDS that
is sensed at CS pin and on and offcomparators decide about SR
operation based on thisvoltage. Driver is turned on and exception
timer is startedwhen VCS drops below VCS_TH_ON. During
minimumon−time blanking interval off comparator is not active.
CSpin voltage is above VTH_CS_OFF after minimum on−timeelapses so
driver is turned−off and because exception timerdoesn’t elapse,
min_ton blanking interval is started. Duringthis time on comparator
output is blanked. Reason is to avoidquick driver turning on and
off that would just increaseconsumption. When min_ton blanking
interval elapses CS
voltage is again below VTH_CS_ON and exception timer isnot
elapsed, driver can be turned on again simultaneouslywith minimum
on−time interval. Driver is turned off againalmost at the end of
conduction phase, but this is correct turnoff. Min−ton blanking
interval doesn’t start, becauseexception timer elapsed before so SR
controller waits forVCS > VCS_TH_RESET to start minimum off time
blankingtimer.
Exception timer length is given as multiple of minimumon time
interval. It should be not set to longer time than
tEXC �1
3 � fSWMAX (eq. 3)where fSWMAX is maximum LLC switching
frequency.
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Figure 79. Exception Timer Operation
Body diode drop(not in scale)
Driver can beturned on again up
to this point
VDS = VCS
VTH_CS_OFF
VTH_CS_ON
Min ON−time
ISEC
VTH_CS_RESET
Min OFF−time
DRV
Min ON−time_2
Exception timer
∼0.25 A
∼2.7 A
Light Load DetectionLight load detection feature is feature
which task is to
decrease SR controller consumption during time when SRtransistor
switching is not needed. This is usually during noload and light
load condition when static SR controllerconsumption starts to play
role. Goal is to disable controllerduring no switching time to
eliminate static consumptionand turn−on SR transistor as soon as
possible whenswitching comes.
Internal simplified block diagram is shown in Figure 80.Main
parts of this system are comparator at CS input that
informs about CS voltage lower than zero (body diode or
SRtransistor conducting), LLD timer with set able nominaltime and
possibility to reduce it to one half and finally D flipflop with
Disable signal output. Nominal time can be set byresistor at LLD
pin connected to ground or internally duringproduction. Recommended
resistor values are shown inTable 6. In case of very noisy system,
capacitor in parallelto LLD resistor may be used. Capacitor value
impactsstart−up time, because capacitor has to be charged
abovedisable threshold by internal LLD current source.
Figure 80. LLD Internal Block Diagram
VCSLLD
CSRESET ELAPSED
LLDTimer
Set maxto tLLD
Set maxto tLLD / 2
tim > 1/4 tLLD
S
RQ
Disable
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Table 6. PIN FUNCTION DESCRIPTION
LLD setting tLLD [�s] IC disabled 70 130 280 540 1075 LLD
disabled
RLLD [k�] 470*
*floating pin allowed, small cap for noise robustness
improvement recommended
Logic function is also described by bubble diagram inFigure 81.
LLD timer is running every time when CS pinvoltage is positive
(body diode and or transistor notconducting). If conduction doesn’t
come sooner than LLDtimer elapses, DISABLE flag is set (IC is sent
into lowconsumption mode), LLD timer length is changed to tLLD /2
(this adds some hysteresis in system and helps keepingoverall
system stable) and timer is also reset. SR controllerwaits for
falling edge at CS pin (begin of new conductioncycle). When CS goes
negative, disable mode is deactivated
and IC starts to wake up (takes tLLD_DIS_REC, system wakeup is
controlled same as exit from disable mode by TRIG /DIS pin). End of
conduction phase (CS voltage goespositive) starts LLD timer. If
next conduction phase comesshortly after first (pulses in skip
burst) so shortly than tLLD/ 4 just LLD timer is reset. LLD timer
length is set back totLLD only when new conduction phase comes
after previousin time between tLLD / 4 to tLLD / 2. This situation
happenswhen load is slowly increased and skip bursts come
moreoften.
Figure 81. LLD Operation Bubble Diagram
Reset TIMDISABLE = 0
LLD_CMP &TIM CNT < tLLD
LLD_CMP
Start
Reset TIMDISABLE = 0
tTIM = tLLD
DISABLE = 1tTIM = 1/2 tLLD
Reset TIM
LLD TIM isRUNNING
1/4
Example of LLD operation with flyback convertor can beseen in
Figure 82. SMPS works under heavy load from point0 to 1 where
switching pulses comes regularly at highfrequency that resets LLD
timer soon after begin ofcounting. Load is significantly decreased
to light load atpoint 1 so primary controller turns to skip mode.
LLD timerelapses during skip so controller enters disable mode
withvery low consumption and change LLD timer maximum totLLD / 2.
Switching pulse in skip comes at time 3, this resetsLLD timer and
starts IC wake−up. Controller is waked upfully before point 4 and
turns−on SR transistor. There isagain no switching from 4 to 6 and
thanks to it, LLD timerelapses at point 5 and controller enters
disable mode again.Disable mode is ended at time 6, because new
cycle comes.SR controller wakes−up and next pulse in skip burst
isconducted via SR transistor. Time between 7 and 8 is delaybetween
skip burst. Time is still less than tLLD / 4, LLD timer
interval is not changed. Pulse at time 8 is fully conducted
viaSR transistor, because controller was not in disable modebefore
pulse came. No switching period between 9 and 11 islonger than tLLD
/ 2 that changes LLD timer setting to tLLD.This is because shorter
delay between skip burst meanshigher load. Pulses are transferred
via SR transistor at time11 and 12, because disable mode was not
activated. Load isbeing decreased again between time 12 to 15 so at
time 15SR controller enters disable mode and LLD timer time
isreduced again to tLLD / 2. Second pulse in skip burst is
againtransferred via turned on SR transistor. Disable mode
isactivated after tLLD / 2 at time 18. Load is sharply changedat
time 19 that means LLD timer is reset each pulse andtimers time is
kept at tLLD / 2. Load is removed at time 20 anddisable is
activated at time 21. Suitable LLD timer setting forflyback type of
SMPS is 540 or 1075 μs (for special type280 μs).
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Figure 82. LLD Operation with Flyback SMPS
VDS = VCS
DRV
DIS
ICC
LLD timtLLD
t1 t2 t4 t1 t2
t1t2
t4
0 1 42 3 5 6 7 8 9 101112 13 14 15 16 17 18 19 20 21 22
t2
Example of LLD operation with LLC convertor can beseen in Figure
83. SMPS works under heavy load from point0 to 3. Both LLD timers
are reset each cycle before LLDtimer reaches tLLD / 4 and disable
mode is not activated.SMPS load decreases at point 3 and goes into
skip. LLDtimers elapse during no switching time and change LLDtimer
time to tLLD / 2. When skip burst comes at time 6channel 2 starts
to wake up, channel 1 starts to wake up attime 7. Both channels are
ready to conduct via SR transistor
at time 8 respectively 9. Skip burst ends at time 12, LLDtimers
elapse at time 13 and 14 (reached tLLD / 2) and SRcontrollers enter
disable mode. Controllers wake up at time15 and 16 same as was in
time 6 and 7. SMPS goes into skipin time 21, but load is connected
soon and SMPS starts tooperate under higher load from time 22. LLD
timers reachtime higher than tLLD / 4 but lower than tLLD / 2 so
LLDtimers maximum is set to tLLD. LLD timer setting for LLCmay be
set to lower times.
Figure 83. LLD Operation with LLC SMPS
VDS1 = VCS1
DRV1
LLD tim1
0 1 42 3 5 6 7 8 9 10 11 12 13 14 15 16 17 19 20 21 22
VDS2 = VCS2
DRV2
DIS1DIS2
LLD tim2tLLD1
tLLD1
23 24 25 26 27 28
t1
t2
t4
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Operation flowFollowed bubble diagram at Figure 84 shows
overall
operation flow. Black bubbles are fundamental parts ofsystem.
States for dV / dt feature are colored by blue colorand states for
LLC feature (exception timer) are in red. LLC
and dV / dt features are never activated both at same
time.Operation starts in bubble start where system comes whenVCC is
higher than UVLO level and / or disable mode isactivated (by LLD or
TRIG / DIS pin).
Figure 84. Overall Operation Bubble Diagram
Power dissipation calculationIt is important to consider the
power dissipation in the
MOSFET driver of a SR system. If no external gate resistoris
used and the internal gate resistance of the MOSFET isvery low,
nearly all energy losses related to gate charge aredissipated in
the driver. Thus it is necessary to check the SRdriver power losses
in the target application to avoid overtemperature and to optimize
efficiency.
In SR systems the body diode of the SR MOSFET startsconducting
before SR MOSFET is turned−on, because thereis some delay from
VTH_CS_ON detect to turn−on the driver.On the other hand, the SR
MOSFET turn off process alwaysstarts before the drain to source
voltage rises up
significantly. Therefore, the MOSFET switch alwaysoperates under
Zero Voltage Switching (ZVS) conditionswhen in a synchronous
rectification system.
The following steps show how to approximately calculatethe power
dissipation and DIE temperature of the NCP4306controller. Note that
real results can vary due to the effectsof the PCB layout on the
thermal resistance.
Step 1 – MOSFET gate to source capacitance:During ZVS operation
the gate to drain capacitance does
not have a Miller effect like in hard switching systemsbecause
the drain to source voltage does not change (or itschange is
negligible).
Figure 85. Typical MOSFET Capacitances Dependency on VDS and VGS
Voltages
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Ciss � Cgs � Cgd (eq. 4)
Crss � Cgd (eq. 5)
Coss � Cds � Cgd (eq. 6)
Therefore, the input capacitance of a MOSFET operatingin ZVS
mode is given by the parallel combination of the gateto source and
gate to drain capacitances (i.e. Ciss capacitancefor given gate to
source voltage). The total gate charge,Qg_total, of most MOSFETs on
the market is defined for hardswitching conditions. In order to
accurately calculate thedriving losses in a SR system, it is
necessary to determine thegate charge of the MOSFET for operation
specifically in aZVS system. Some manufacturers define this
parameter asQg_ZVS. Unfortunately, most datasheets do not provide
thisdata. If the Ciss (or Qg_ZVS) parameter is not available thenit
will need to be measured. Please note that the inputcapacitance is
not linear (as shown Figure 85) and it needsto be characterized for
a given gate voltage clamp level.
Step 2 – Gate drive losses calculation:Gate drive losses are
affected by the gate driver clamp
voltage. Gate driver clamp voltage selection depends on thetype
of MOSFET used (threshold voltage versus channelresistance). The
total power losses (driving loses andconduction losses) should be
considered when selecting thegate driver clamp voltage. Most of
today’s MOSFETs for SRsystems feature low RDS_ON for 5 V VGS
voltage. The
NCP4306 offers both a 5 V gate clamp and a 10 V gateclamp for
those MOSFET that require higher gate to sourcevoltage.
The total driving loss can be calculated using the selectedgate
driver clamp voltage and the input capacitance of theMOSFET:
PDRV_total � VCC � VCLAMP � CG_ZVS � fSW (eq. 7)
Where:VCC is the NCP4306 supply voltageVCLAMP is the driver
clamp voltageCg_ZVS is the gate to source capacitance of theMOSFET
in ZVS modefsw is the switching frequency of the target
application
The total driving power loss won’t only be dissipated inthe IC,
but also in external resistances like the external gateresistor (if
used) and the MOSFET internal gate resistance(Figure 86). Because
NCP4306 features a clamped driver,it’s high side portion can be
modeled as a regular driverswitch with equivalent resistance and a
series voltagesource. The low side driver switch resistance does
not dropimmediately at turn−off, thus it is necessary to use
anequivalent value (RDRV_SIN_EQK) for calculations. Thismethod
simplifies power losses calculations and stillprovides acceptable
accuracy. Internal driver powerdissipation can then be calculated
using equation 10:
Figure 86. Equivalent Schematic of Gate Drive Circuitry
VCC
VCC − VCLAMP−
+
DRV
GNDRDRV_SINK_EQ
RDRV_SOURCE_EQRG_EXT
SR MOSFET
RG_INT
CG_ZVS
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PDRV_IC �12� Cg_ZVS � VCLAMP2 � fSW �� RDRV_SINK_EQRDRV_SINK_EQ
� RG_EXT � Rg_int�� Cg_ZVS � VCLAMP � fSW � (VCC � VCLAMP) (eq.
8)
� � 12� Cg_ZVS � VCLAMP2 � fSW �� RDRV_SOURCE_EQRDRV_SOURCE_EQ �
RG_EXT � Rg_int� (eq. 9)
Where:RDRV_SINK_EQ is the NCP4306 driver low side
switchequivalent resistance (1.6 Ω)RDRV_SOURCE_EQ is the NCP4306
driver high sideswitch equivalent resistance (7 Ω)RG_EXT is the
external gate resistor (if used)Rg_int is the internal gate
resistance of the MOSFET
Step 3 – IC consumption calculation:In this step, power
dissipation related to the internal IC
consumption is calculated. This power loss is given by theICC
current and the IC supply voltage. The ICC currentdepends on
switching frequency and also on the selected mintON and tOFF
periods because there is current flowing outfrom the MIN_TON and
MIN_TOFF pins. The mostaccurate method for calculating these losses
is to measurethe ICC current when CLOAD = 0 nF and the IC is
switchingat the target frequency with given min_tON and
min_tOFFadjust resistors. IC consumption losses can be calculated
as:
PCC � VCC � ICC (eq. 10)
Step 4 – IC die temperature arise calculation:The die
temperature can be calculated now that the total
internal power losses have been determined (driver lossesplus
internal IC consumption losses). The SO8 packagethermal resistance
is specified in the maximum ratings tablefor a 35 μm thin copper
layer with no extra copper plates onany pin (i.e. just 0.5 mm trace
to each pin with standardsoldering points are used).
The die temperature is calculated as:
TDIE � (PDRV_IC � PCC) � R�J�A � TA (eq. 11)
Where:PDRV_IC is the IC driver internal power dissipationPCC is
the IC control internal power dissipation R J−A is the thermal
resistance from junction to ambientTA is the ambient
temperature
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OPN coding tableNCP4306 OPN is built from prefix of NCP4306
and
postfix that consist of seven letters. Meaning of these
lettersare shown in table 7.
Table 7. OPN CODING TABLE
NCP4306xxxxxxx
Postfix Index Parameter Postfix Parameter
1 Pinout A MIN_TON, MIN_TOFF, LLD, TRIG / DIS − 8 pins
B MIN_TON, LLD
C MIN_TOFF, LLD
D MIN_TON, MIN_TOFF
E MIN_TOFF, TRIG / DIS
F MIN_TON, TRIG / DIS
G TRIG / DIS, LLD
H None
2 DRV A DRV CLMP = 10 V
B DRV CLMP = 5 V
3 dV / dt + exception A None
D Flyback (dV / dt) − 100 V / μs
H LLC exception − multiplier 4
4 MIN_TON A 130 ns
B 220 ns
C 310 ns
D 400 ns
E 500 ns
F 600 ns
G 800 ns
H 1000 ns
I 1200 ns
J 1400 ns
K 1700 ns
L 2000 ns
Z External
5 MIN_TOFF A 0.9 μs
B 1.0 μs
C 1.1 μs
D 1.2 μs
E 1.4 μs
F 1.6 μs
G 1.8 μs
H 2.0 μs
I 2.2 μs
J 2.4 μs
K 2.6 μs
L 2.9 μs
M 3.2 μs
N 3.5 μs
O 3.9 μs
Z External
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Table 7. OPN CODING TABLE (continued)
NCP4306xxxxxxx
Postfix Index ParameterPostfixParameter
6 LLD A 68 μs
B 130 μs
C 280 μs
D 540 μs
E 1075 μs
F Disabled
Z External
7 Reserved A −
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PACKAGE DIMENSIONS
SOIC−8 NBCASE 751−07
ISSUE AK
SEATINGPLANE
14
58
N
J
X 45�
K
NOTES:1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.2. CONTROLLING DIMENSION: MILLIMETER.3.
DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBARPROTRUSION SHALL BE 0.127 (0.005)
TOTALIN EXCESS OF THE D DIMENSION ATMAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEWSTANDARD IS 751−07.
A
B S
DH
C
0.10 (0.004)
DIMA
MIN MAX MIN MAXINCHES
4.80 5.00 0.189 0.197
MILLIMETERS
B 3.80 4.00 0.150 0.157C 1.35 1.75 0.053 0.069D 0.33 0.51 0.013
0.020G 1.27 BSC 0.050 BSCH 0.10 0.25 0.004 0.010J 0.19 0.25 0.007
0.010K 0.40 1.27 0.016 0.050M 0 8 0 8 N 0.25 0.50 0.010 0.020S 5.80
6.20 0.228 0.244
−X−
−Y−
G
MYM0.25 (0.010)
−Z−
YM0.25 (0.010) Z S X S
M� � � �
1.520.060
7.00.275
0.60.024
1.2700.050
4.00.155
� mminches
�SCALE 6:1*For additional information on our Pb−Free strategy
and soldering
details, please download the ON Semiconductor Soldering
andMounting Techniques Reference Manual, SOLDERRM / D.
SOLDERING FOOTPRINT*
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PACKAGE DIMENSIONS
ÉÉÉÉ
TSOP−6CASE 318G−02
ISSUE V
2 3
456
D
1
eb
E1
A1
A0.05
NOTES:1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994.2.
CONTROLLING DIMENSION: MILLIMETERS.3. MAXIMUM LEAD THICKNESS
INCLUDES LEAD FINISH. MINIMUM
LEAD THICKNESS IS THE MINIMUM THICKNESS OF BASE MATERIAL.4.
DIMENSIONS D AND E1 DO NOT INCLUDE MOLD FLASH,
PROTRUSIONS, OR GATE BURRS. MOLD FLASH, PROTRUSIONS, ORGATE
BURRS SHALL NOT EXCEED 0.15 PER SIDE. DIMENSIONS DAND E1 ARE
DETERMINED AT DATUM H.
5. PIN ONE INDICATOR MUST BE LOCATED IN THE INDICATED ZONE.
c
*For additional information on our Pb−Free strategy and
solderingdetails, please download the ON Semiconductor Soldering
andMounting Techniques Reference Manual, SOLDERRM / D.
SOLDERING FOOTPRINT*
DIMA
MIN NOM MAXMILLIMETERS
0.90 1.00 1.10A1 0.01 0.06 0.10b 0.25 0.38 0.50c 0.10 0.18 0.26D
2.90 3.00 3.10E 2.50 2.75 3.00
e 0.85 0.95 1.05L 0.20 0.40 0.60
0.25 BSCL2−0� 10�
1.30 1.50 1.70E1
E
RECOMMENDED
NOTE 5
LCM
H
L2
SEATINGPLANE
GAUGEPLANE
DETAIL Z
DETAIL Z
0.606X
3.200.956X
0.95PITCH
DIMENSIONS: MILLIMETERS
M
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