Mosfet Driver Theory

IXYS Corporation; 3540 Bassett Street; Santa Clara, CA 95054; Tel: 408-982-0700; Fax: 408-496-0670IXYS Semiconductor GmbH; Edisonstr. 15; D-68623; Lampertheim, Germany; Tel: +49-6206-503-0; Fax: +49-6206-503627

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MOSFET/IGBT DRIVERSTHEORY AND APPLICATIONS

By Abhijit D. Pathak

© 2001 IXYS Corporation

APPLICATION NOTE AN0002

1. Introduction

1.1. MOSFET and IGBT Technology.1.2. MOSFET Models and critical parameters1.3. Turn-on and Turn-off phenomenon and their explanations1.4. Power losses in Drivers

2. Types of Drivers

2.1. IC Gate Drivers2.2. Techniques available to boost current outputs2.3. Techniques available to generate negative bias during turn-off2.4. Need for under-voltage protections2.5. Overload and Short Circuit Protection

3. Isolation Techniques

3.1. Employing Charge-pump and Bootstrap Techniques3.2. Examples of use of Opto-couplers in practical Driver Circuits3.3. Examples using transformers in practical Driver Circuits

4. IXYS Line of MOSFET/IGBT Drivers

4.1. Technical details of all IXYS Drivers4.2. Features and Advantages of IXYS Drivers4.3. Applying IXYS Drivers in various topologies

5. Practical Considerations

6. Conclusion


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1. INTRODUCTION

Modern Power Electronics makes generous use of MOS-FETs and IGBTs in most applications and, if the presenttrend is any indication, the future will see more and moreapplications making use of MOSFETs and IGBTs.

Although sufficient literature is available on characteris-tics of MOSFETs and IGBTs, practical aspects of drivingthem in specific circuit configurations at different powerlevels and at different frequencies require that design en-gineers pay attention to a number of aspects.

An attempt is made here to review this subject with someillustrative examples with a view to assist both experienceddesign engineers and those who are just initiated into thisdiscipline .

1.1 MOSFET AND IGBT TECHNOLOGY

Due to the absence of minority carrier transport, MOS-FETs can be switched at much higher frequencies. Thelimit on this is imposed by two factors: transit time of elec-trons across the drift region and the time required to chargeand discharge the input Gate and ‘Miller’ capacitances.

IGBT derives its advantages from MOSFET and BJT. Itoperates as a MOSFET with an injecting region on itsDrain side to provide for conductivity modulation of theDrain drift region so that on-state losses are reduced,especially when compared to an equally rated high volt-age MOSFET.

As far as driving IGBT is concerned, it resembles a MOS-FET and hence all turn-on and turn-off phenomena com-ments, diagrams and Driver circuits designed for drivingMOSFET apply equally well to an IGBT. Therefore, whatfollows deals only with MOSFET models.

N+

Co

CpC CG DN+

GATE E lectrode

SOURCEM ETALIZATION

P-BASER B

DBCN-DRIFT

Depletion boundaries

Fig. (1A) M O SFET ce ll internal structure

Fig. (1B) Cross sectiona l view of N-Channel M OSFET showing various inter-junction capacitances

C

C

C G S

D S

G D

N+P

N-

N+

N+P

SO URCE G ATE

PARAS ITIC BJT

DRAIN

DRAIN BO DY DEP LET IO N LAY ER

INTERNAL BO DY D IO DE

O VG S

Actua lLinearized

gm = ID

DR

AIN

CU

RR

EN

T

G S(th)V

G SV

ID

ID

VG S

Fig. (2) Transfer characte ristics of a power M OSFET

1.2 MOSFET MODELS AND CRITICAL PARAMETERS

Fig. (1A) shows the internal cell structure of a DMOS

MOSFET. As can be seen, the Gate to Source Capaci-tance consists of three components, namely, Cp, the com-ponent created by the Gate Electrode over the P-baseregion; C

N+, due to the overlap of the Gate Electrode above

the N+ source region and, CO, arising due to the proximity

of the Gate Electrode to the source metallization. In fact,all these are added to yield C

GS, which we call Gate-to-

Source Capacitance. It is this total value of capacitancethat needs to be first charged to a critical threshold volt-age level V

GS(th), before Drain Current can begin to flow.

The Gate-to-Drain capacitance, CGD

, is the overlap ca-pacitance between the Gate electrode and the N-drift Drainregion. C

GD is sometimes referred to as the ‘Miller’ ca-

pacitance and contributes most to the switching speedlimitation of the MOSFET. The junction capacitance be-tween the drain to the P-Base region is C

DS. The P-Base

region of the MOSFET is shorted to the N+ source. Fig.(2) shows curve of I

D (Drain Current) versus V

GS (Gate Source


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Voltage). The graph has a slope ( ID/ V

GS) equal to g

m,

which is called transconductance. Please note that the

actual relationship between VGS

and ID is shown by dotted

line and it can be observed that in the vicinity of VGS(th)

, therelationship between V

GS and I

D is parabolic in nature:

ID= K [V

GS-V

GS(th)]2 Eq.1.1

However, for Power MOSFETs, it is appropriate to con-sider the relationship to be linear for values of V

GS above

VGS(th).

The manufacturer’s data sheet value of VGS(th)

is speci-fied at 25

oC.

Fig. (3A) shows a symbol of N-Channel MOSFET and anequivalent model of the same with three inter-junction para-sitic capacitances, namely: C

GS, C

GD and C

DS. I have shown

all these as variable as they indeed are. For example theC

GD, decreases rapidly as the Drain to Source voltage rises,

as shown in Fig. (3B). In Fig. (3B), the high value of CGD

iscalled C

GDh, while the low value of C

GD is termed C

GDl. Fig.

(1B) shows another cross-sectional view of a MOSFETwith all these capacitances. In addition, It also shows theinternal body diode and the parasitic BJT.

1.3 Turn-on and Turn-off Phenomena

1.3.1 Turn-on Phenomenon

To understand Turn-on and Turn-off phenomena of the PowerMOSFET, we will assume clamped inductive switching asit is the most widely used mode of operation. This is shownin Fig. (4A) and Fig. (4B). A model of MOSFET is shownwith all relevant components, which play a role in turn-onand turn-off events. As stated above, MOSFET’s Gate toSource Capacitance C

GS needs to be charged to a critical

voltage level to initiate conduction from Drain to Source. Afew words of explanation will help understand Fig. (4A)and Fig. (4B). The clamped inductive load is being shownby a current source with a diode D connected antiparallelacross the inductor. The MOSFET has its intrinsic internalGate resistance, called R

Gint. As described above, the in-

ter-junction parametric capacitances (CGS

, CGD

and CDS

)are shown and connected at their proper points. V

DD repre-

sents the DC Bus voltage to the Drain of the MOSFETthrough the clamped inductive load. The Driver is suppliedby Vcc of value Vp and its ground is connected to thecommon ground of V

DD and is returned to the Source of

the MOSFET. The output from the Driver is connected tothe Gate of the MOSFET through a resistor R

Gext.

Now when a positive going pulse appears at the input ter-minal of the Driver, an amplified pulse appears at the out-put terminal of the Driver with an amplitude Vp. This is fedto the Gate of the MOSFET through R

Gext. As one can see

the rate of rise of voltage, VGS

, over Gate and Source termi-nals of the MOSFET is governed by value of the total resis-tance in series (R

dr+R

Gext+R

Gint) and total effective value of

capacitance (CGS

+CGD

). R

dr stands for the output source

impedance of the Driver. Rgext

is the resistance one gener-ally puts in series with the Gate of a MOSFET to controlthe turn-on and turn-off speed of the MOSFET.

Fig. (3A) Sym bol and equivalent circuit o f a MO SFET

Symbol ofN-Channel M OSFET

D

G

S

G

CG D

G SC

GR

D

S

C DSR DS(on)int

V DSG SV

Fig. (3B) C variation Vwith respect to DS

DSV =

G DC

G DC

CG D

h

l

G D

Fig (4A) A M OSFET being tu rned on by a driver in a clam ped inductive load.

G ext.O

t

i/p R

o/p

DRIVERdr

GI

R

V

+V

CC

DD

G D

G S

G

CG s

G in t.

I

R

S

C

C

IG DD

D

DS


4

The waveforms drawn in Fig. (5) show variation of differentparameters with respect to time, so as to clearly explainthe entire turn-on sequence. In Fig. (4A) and Fig. (4B), theFree Wheeling Diode D is assumed to be ideal with zeroreverse recovery current. The waveforms shown in Fig. 5are based on this assumption.

From time zero to t1, (C

GS+C

GDl) is exponentially charged

with a time constant T1=(Rdr+R

Gext+R

Gint)x(C

GS+C

GDl), until

Gate-to-source voltage reaches VGS(th)

. In this time period,neither the Drain voltage nor the Drain current are affected,i.e. Drain voltage remains at V

DD and Drain current has not

commenced yet. This is also termed turn-on delay. Notethat between 0 to t

1, as V

GS rises, I

GS falls exponentially,

more or less like a mirror image of VGS

, because from thepoint of view of circuit analysis, it is an RC Circuit.

After time t1, as the Gate-to-Source voltage rises above

VGS(th)

, MOSFET enters linear region as shown in Fig. (2).At time t

1, Drain current commences, but the Drain to

Source voltage VDS

is still at VDD

. However, after t1, I

D builds

up rapidly. As can be seen in Fig. (3B), from time t1 to t

2,

CGD

increases from C GDl

to CGDh

and current available fromthe Driver is diverted to charge this increased value ofC

GDh. As we shall see later, this is the real test of a Driver-

how fast it can charge CGDh

in addition to CGS.

Between t1 and t

2, the Drain current increases linearly with

respect to VGS

. At time t2, the Gate to Source voltage

enters the Miller Plateau level. At time t2, the Drain voltage

begins to fall rapidly, while the MOSFET is carrying fullload current. During the time interval, t

2 to t

4, V

GS remains

clamped to the same value and so does IGS

. This is calledthe Miller Plateau Region. During this interval most of thedrive current available from the driver is diverted to dis-

charge the CGD

capacitance to enhance rapid fall of Drainto Source voltage. Only the external impedance in serieswith V

DD limits drain current

.

Beyond t4, V

GS begins to exponentially rise again with a

time constant T2 = (R

dr+R

Gext+R

Gint)x(C

GS+C

GDh). During this

time interval the MOSFET gets fully enhanced, the finalvalue of the V

GS determining the effective R

DS(on). When V

GS

reaches its ultimate value, VDS

attains its lowest value,determined by V

DS= I

DSx R

DS(on) .

In Fig. (5), A1 represents area of IG curve from time t

1 to t

2.

This represents charge on (CGS

+CGD

), as it is the integra-tion of Gate current over a time period.

Similarly A2 repre-

sents charge on CGD,

because it is an integration of IG with

respect to time from t2 to t

3, during which time the Miller

effect is predominant.

If one considers diode ‘D’ not to be ideal, then the reverserecovery of the diode will influence the turn on behaviorand the waveforms would look like what is drawn in Fig.(6). As the diode undergoes reverse recovery, you cansee a hump in the waveform of V

GS as well as I

D. This

occurs at and around time t2 .

Fig. (4B) A MO SFET being turned off by a driver in a clam ped inductive load.

+V

V

dr

t

i/p

O

RG ext.R

CC

DD

G

CG S

CG in t.R G D

S

C

D

D

DS

TIME = t

DR

AIN

CU

RR

EN

T

(0 ,0) t1 t2 t3 t4

I

I (t)

DR

AIN

SO

UR

CE

VO

LTA

GE

V

TIME = t

GA

TE

CU

RR

EN

T

I

TIME = t

GA

TE

-SO

UR

CE

VO

LTA

GE

T IME = t

O/P

OF

DR

IVE

R

VD R

G S

G S (th )

D S

D

D

D S

D S

TIME = t

V (t)

V (on)

A1A2

I (t)

V

V

V (t)=Vp(1-e )-t/T1

-t/T2V (t)=Vp(1-e )

Vp

(a)

(b )

(c )

(d )

(e )

Fig. (5) MO SFET turn on sequence

G S

G S

GG


5

1.3.2. The Turn-off Phenomenon:

The turn-off phenomenon is shown in Fig. (7). As can beexpected, when the output from the Driver drops to zero forturning off MOSFET, V

GS initially decays exponentially at

the rate determined by time constant T2=

(Rdr+R

Gext+R

Gint)x(C

GS+C

GDh) from time 0 to t

1; however, af-

ter t4, it decays exponentially at the rate determined by T1

= (Rdr+RGext+RGint)x(CGS+CGDl

). Please note that thefirst delay in the turn off process is required to dischargethe C

ISS capacitance from its initial value to the Miller Pla-

teau level. From t = 0 to t = t1, the gate current is flowing

through CGS

and CGD

capacitances of MOSFET. Notice thatthe drain current I

D remains unchanged during this time

interval, but the Drain Source voltage VDS

just begins torise. From t

1 to t

2 , V

DS rises from I

D x R

DS(on) towards its

final off state value of VDS(off)

, where it is clamped to the DCBus voltage level by the diode in the clamped inductiveswitching circuit being studied. This time interval also cor-responds to the Miller region as far as the gate voltage isconcerned as mentioned above, which keeps V

GS constant.

During the next time interval, the VGS

begins to fall furtherbelow V

GS(th). C

GS is getting discharged through any exter-

nal impedance between Gate and Source terminals. The

MOSFET is in its linear region and Drain current ID drops

rapidly towards zero value. Remember that the Drain Volt-age V

DS was already at its off state value V

DS(off) at the

beginning of this interval. Thus at t4, the MOSFET is fully

turned off.

Manufacturer’s Data Sheet of MOSFET gives values ofC

ISS, C

RSS and C

OSS. The following relationships help re-

late these to inter-junction parasitic capacitances de-scribed so far:

CGD = CRSS

CGS = CISS - C

RSS Eq. 1.2

COS = COSS - C

RSS

As CGD

and CDS

capacitances are dependent on VDS

, DataSheet values are valid only at specified test conditions.To compute average effective values of these capaci-tances, one can use following formulae, which determinesthese based on required charge to establish voltagechanges across these capacitances:

Effective CGD = 2(CRSS specified)(VDS.specified/VDS.off)1/2

Effective COSS= 2(COSS specified)(VDS.specified/VDS.off)1/2

Eq.1.3

TIME = t

DR

AIN

CU

RR

EN

T

(0 ,0) t1 t2 t3 t4

II (t)

DR

AIN

SO

UR

CE

VO

LTA

GE

V

TIME = t

GAT

E C

UR

RE

NT

If id

TIME = t

GAT

E-S

OU

RC

E V

OLT

AG

E

T IME = t

O/P

OF

DR

IVE

R

VD R

G S

TIME = t

V (t)

VDS (on)

Ifid(t)

V

V

Vp

Fig. (6) MO SFET tu rn on sequence showing the effect o f body diode reverse recovery.

G S (th )

D S

D S

D

DTIME = t

DR

AIN

CU

RR

EN

T

I

GAT

E C

UR

RE

NT

I

T IME = t

GAT

E-S

OU

RC

E V

OLT

AG

E

T IME = t

O/P

OF

DR

IVE

RV D R

G S

G S (th )

TIME = t

V

V

LEVEL Vp

Fig. (7) MO SFET turn o ff sequence

-t/T2e

-t/T1e

DR

AIN

SO

UR

CE

VO

LTA

GE

V

TIME = t0 t1 t2 t3 t4

-VE+V E

G

D

D S


6

1.4 POWER LOSSES IN DRIVERS AND DRIVEN MOS-FET/IGBT

For determining the power loss in a Driver while driving apower MOSFET, the best way is to refer to the GateCharge Q

G vs. V

GS curve for different values of V

DS(off).

PGATE

= VCC

*Qg*fsw

Eq. 1.4

wherein Vcc is the Driver’s supply voltage, Qg is the total

Gate Charge of the MOSFET being driven and fsw

is theswitching frequency. It is prudent then to choose a MOS-FET with lower value of Qg

and it is here that IXYS’ series

of low Gate Charge MOSFETs with a suffix ‘Q’ are pre-ferred because they as well as the drivers incur lowerlosses.

As far as switching losses in a MOSFET are concerned,as can be seen in Fig. (5), Fig. (6) and Fig. (7), there aresome short time-intervals, during which finite V

DS and fi-

nite ID coexist, albeit momentarily. When this happens

during turn-on, the actual integration:

VDS

(t) ID(t)

dt Eq. 1.5

is defined as Turn-On switching energy loss. Likewise,during turn-off, when finite values of I

D and V

DS coexist,

integration of:

VDS(t)ID(t)dt Eq. 1.6

is called Turn-off switching energy loss in a MOSFET.Amongst the responsible parameters determining theseswitching energy losses, C

ISS, C

OSS and C

RSS affect the turn-

on and turn-off delays as well as turn-on and turn-off times.

For an IGBT, it would be similarly shown that:

VCE

(t)IC(t)dt Eq. 1.7

represents switching energy loss. Needless to emphasizethat the time interval for these integrals would be the ap-propriate time during which finite values of I

D and V

DS or

VCE

and IC coexist in a MOSFET or IGBT respectively. Av-

erage switching energy lost in the device can be com-puted thusly:

MOSFET: Ps = 1/2*VDS *ID *fsw*(ton+toff) Eq. 1.8IGBT: Ps = 1/2*VCE *IC *fsw*(ton+toff) Eq. 1.9

Main emphasis in modern Power Electronics is on reduc-ing total losses dissipated in devices and subsystems forhigher operating efficiency and achieving more compactdesigns, reducing volume and weight of resultant systems.

Thus, operation at higher and higher switching frequenciesis now a necessity, and as a result, switching losses pre-dominate in power-loss-budget in semiconductor switches.Reducing switching losses then becomes the single mostcrucial goal. Keeping this goal in mind, the entire line ofIXYS MOSFET/IGBT Drivers are designed to facilitate thedesign of drive circuits that yield fast rise and fall times.

2. TYPES OF DRIVERS

2.1 IC DRIVERS

Although there are many ways to drive MOSFET/IGBTsusing hard wired electronic circuits, IC Drivers offer con-venience and features that attract designers. The fore-most advantage is compactness. IC Drivers intrinsicallyoffer lower propagation delay. As all important parametersare specified in an IC Driver, designers need not gothrough time consuming process of defining, designingand testing circuits to drive MOSFET/IGBTs.

2.2 TECHNIQUES AVAILABLE TO BOOST CURRENTOUTPUTS

Totem pole stage with N-Channel and P-Channel MOS-FETs can be used to boost the output from an IC Driver.The disadvantage is that the signal is inverted and alsothere exists shoot through when common gate voltage isin transition.

Totem pole arrangement using matched NPN-PNP tran-sistors, on the other hand, offer many advantages, whileboosting the output currents from IC Drivers. Shootthrough phenomenon is absent in this case. The pair oftransistors protects each other’s base-emitter junctionsand handle current surges quite well. One such arrange-ment is shown in Fig. (14). Here Q1 is a NPN transistor,while Q2 is a matched PNP transistor with appropriatecollector current rating and switching speed to satisfy Driverequirement for the High Power IGBT. Another featureadded is –ve bias for guaranteed fast switch-off even inelectrically noisy environment. This is done, by utilizingpower supply with +15 and –5 Volts output, whose com-mon ground is connected to the IGBT emitter.

IXDD408 is a very high speed IC Driver with extremelyshort rise and fall times and propagation delays. Its Vccrating is 25 VDC and can actually deliver 8 Amperes peakoutput current. The arrangement shown in Fig. (14) doesa few more things in addition to boosting the output cur-rent still higher. It allows one to choose different Turn-Onand Turn-Off times by choosing different values of R

gon

and Rgoff.

It allows for incorporating –ve bias for reasonsexplained above. A pair of 18V Zener diodes with their cath-odes connected together, protects the Gate-Emitter Junc-


7

tion of IGBT from voltage spikes.

2.3 TECHNIQUES TO GENERATE –Ve BIAS DURINGTURN-OFF

Importance of –ve bias during turn-off for practically allsemiconductor switches cannot be overemphasized, asone may recall from the days of Bipolar transistors. Thishelps to quickly remove any charge on the C

GS and C

GD in

the case of MOSFETs and IGBTs, thus considerably ac-celerating turn-off.

It is important to understand that turn-on speed of a MOS-FET or IGBT can be increased only up to a level matchedby the reverse recovery of rectifiers or diodes in a powersupply, because in an inductive clamped load (most com-mon), turn-on of a MOSFET or IGBT coincides with turn-off (or reverse recovery completion) of the rectifier diode.Any turn-on faster than this does not help. Too fast a turn-on could also cause oscillation in the Drain or Collectorcurrent. However, it is always beneficial to have a Driverwith intrinsic low turn-on time and then be able to tailorthis with a series gate resistor.

Turn-off phenomenon, on the other hand, does not haveto wait for any other component in the subsystem. It ishere that any enhancement technique can be utilized. Al-though IXYS drivers themselves feature extremely lowturn-on and turn-off times, arrangement to provide –vebias during turn-off helps still faster turn-off and preventsfalse turn on even in electrically noisy environment.

Fig. (12C) demonstrates one way of generating –ve biasduring turn-off. Fig. (18) shows how to generate –ve biasin a transformer coupled Drive circuit arrangement. HereZener diode can be chosen of appropriate voltage for giv-ing that much –ve bias (plus one diode drop) during turn-off. Another unique feature of circuit in Fig.(18) is its abil-ity to maintain exact pulse wave shape across Gate andsource. In Fig. (16) a method of using isolated DC to DCconverter with outputs of +15 and –5 V is used to powerIXDD414, while by connecting isolated ground of this DCto DC Converter to the emitter of the IGBT being driven,–5 V of –ve bias during turn-off is ensured.

2.4 NEED FOR UNDER-VOLTAGE PROTECTION

Fig. (2) shows a transfer characteristics (ID vs. V

GS) of a

MOSFET. As can be seen for values of VGS

below VGS(th)

the drain current is negligible, but in this vicinity, the de-vice is in its linear (Ohmic) region and concurrent appli-cation of large values of V

DS could cause considerable

amount of localized heating of the junction. In short, whena MOSFET is being used as a switch, any operation in itslinear region could cause overheating or device failure.

Bringing the MOSFET quickly into its saturation from itsoff-state is the Driver’s job. And, if Vcc is below the mini-mum required value, linear operation can ensue to thedetriment of MOSFET. I hasten to add, however, that mostPWM ICs, controller ICs and microcomputer ICs havethis protection feature built-in and, if sharing the sameVcc bus, the Driver IC gets the benefit of this functionbeing implemented elsewhere in the subsystem.

2.5 OVERLOAD/SHORT CIRCUIT PROTECTION

Any operation of MOSFET/IGBT outside the Safe Oper-ating Area (SOA) could cause overheating and eventualdevice failure and should be prevented by an electronicactive monitoring and corrective arrangement.

Load or current sensing could be done by either a HallEffect Sensor or by a Shunt resistor in series with source/emitter terminal. The voltage picked up, which is propor-tional to current, is low pass filtered and then comparedto a preset limit. The comparator output could initiate turn-off of MOSFET/IGBT. A circuit to detect overload/shortcircuit is shown in Fig. (16), where the output FAULT willgo low when it occurs.

All IXDD series of Drivers have an ENABLE pin, which,when driven low, say, by the FAULT output from this com-parator, puts the final N-Channel and P-channel MOS-FETs of the IXDD Driver in its TRISTATE mode. This notonly stops any output from the Driver, but also providesan environment for implementing soft turn-off. There aretwo ways of doing this. Just by connecting a resistor ofappropriate value from Gate to source/emitter, the C

GS

gets discharged through this resistor and, depending onthe value of the connected resistor, soft turn-off of anyduration can be achieved. Another way, as shown in theFig. (9), is to use a signal MOSFET Q1 to pull down theGate, when short circuit is detected. The resistor in se-ries with this signal MOSFET determines the time dura-tion of this soft turn-off. Soft turn-off helps protect IGBT/MOSFET from any voltage transients generated due toLdI

C/dt (or LdI

D/dt) that could otherwise bring about ava-

lanche breakdown. The PC board layout for this circuit isshown in Fig. (10).

For an IGBT, Desat detection (Desat = Desaturation offorward voltage drop) is a method used for short circuit/overload protection. When short circuit/overload occurs,the forward voltage drop of the IGBT (V

CE) rises to dispro-

portionately high values. One must ignore the initial turn-on rise in V

CE, when output from Driver has still not risen

to high enough value. Nevertheless, when VCE

rises to alevel of, say, 7 Volts, in presence of sufficient Gate Drivevoltage, it means the collector current I

C has risen to a


8

disproportionately high value, signaling overload. When avoltage level higher than 6.5 Volts is detected, Gate signalcan be softly turned off, resulting into soft turn-off of theIGBT. Fig. (16) shows how Desat feature can be wired intoa total Driver Circuit, using also other features, such asOpto-isolation and –ve turn-off bias.

3. HIGH SIDE DRIVING TECHNIQUES

3.1 EMPLOYING CHARGE-PUMP AND BOOTSTRAPMETHODS

For driving the upper MOSFET/IGBT in a phase leg em-ployed in a bridge topology, a buck converter or a 2-transitor forward converter, low side drivers cannot be useddirectly. This is because the source/emitter of upper MOS-FET/IGBT is not sitting at ground potential.

Fig. (12A) shows how a charge pump creates a higherVcc to be used for the driver IC for the Upper MOSFET/IGBT. Here the pair of N-Channel and P-Channel MOS-FETs acts as switches, alternately connecting incomingsupply voltage to output through capacitors and Schottkydiodes, isolating it and almost doubling it. Switching fre-quency in several hundred Kilohertz is used and, there-fore, low ripple isolated output voltage is available as DCSupply for the Driver of Upper MOSFET/IGBT. Fig. (12C)illustrates how one IXDD404 can be used as charge pump,delivering 350 mA, and one IXDD408 as a Driver giving +/-8 Amps, in conjunction with IXBD4410 and IXBD4411, fordriving a phase leg of two IXFX50N50 MOSFETs. Fig. (12D)shows how a charge pump delivering as much as 500 mAcan be constructed using one IXDD404; and by utilizingone IXDD414, one can boost the output from IXBD4410and IXBD4411 to +/- 14Amps for driving Size 9 high powerMOSFETs and IGBTs or even MOSFET/IGBT modules inphase leg configuration.

Another method is the Bootstrap Technique as shown inFig. (12B). The basic bootstrap building elements are thelevel shift circuit, bootstrap diode DB, level shift transistorQ1, bootstrap capacitor CB and IXDD408 or IXDD414.The bootstrap capacitor, IXDD408/IXDD414 driver andthe gate resistor are the floating, source-referenced partsof the bootstrap arrangement. The disadvantages of thistechnique are longer turn-on and turn-off delays and 100%duty cycle is not possible. Additionally the driver has toovercome the load impedance and negative voltagepresent at the source of the device during turn-off.

3.2 ACHIEVING GALVANIC ISOLATION BY USINGOPTO-COUPLERS TO DRIVE UPPER MOSFET/IGBT

For driving high side MOSFET/IGBT in any topology, opto-couplers can be used with following advantages:

1. They can be used to give a very high isolation voltage;2500 to 5000 Volts of isolation is achievable by use ofproperly certified opto-couplers.

2. Signals from DC to several MHz can be handled byopto-couplers.

3. They can be easily interfaced to Microcomputers orother controller ICs or any PWM IC.

One disadvantage is that the opto-coupler adds its ownpropagation delay. Another disadvantage of using an opto-coupler is that separate isolated power supply is requiredto feed the output side of the opto-coupler and the driverconnected to it. However, isolated DC to DC Converterswith few thousand Volts of isolation are readily available.These can be used to supply isolated and regulated +ve15 V and –ve 5V to the output side of the opto-couplerand the Driver IC for driving Upper MOSFET/IGBT as isshown in Fig. (16) and Fig. (17). As can be seen, identicalchain of opto-coupler, Driver and DC to DC Convertersare used for even lower IGBTs. This is to guarantee iden-tical propagation delays for all signals so that their time ofarrival at the Gate of IGBT bear the same phase relation-ships with one another as when they originated in the mi-crocomputer.

3.3 USE OF TRANSFORMERS TO OBTAIN GALVANICISOLATION IN DRIVING UPPER MOSFET/IGBT

Using transformers to achieve galvanic isolation is a veryold technique. Depending on the range of frequencies be-ing handled and power rating (voltage and current ratingsand ratios), transformers can be designed to be quite ef-ficient. The gate drive transformer carries very small av-erage power but delivers high peak currents at turn-onand turn-off of MOSFET/IGBT.

While designing or choosing a Gate Drive transformer, thefollowing points should be kept in mind:1. Average power being handled by the transformer

should be used as a design guideline. Margin of safetyshould be taken into account, keeping in mind maxi-mum volt-second product and allowing for worst casetransients with maximum duty ratio and maximum in-put voltage possible.

2. Employing bifilar winding techniques to eliminate anynet DC current in any winding. This is to avoid coresaturation.

3. If operation in any one quadrant of B-H loop is cho-sen, care should be taken for resetting the core.

Advantages of employing transformers for Gate Drive are:

1. There is no need for any isolated DC to DC Converter


9

for driving an upper MOSFET/IGBT .2. There is practically no propagation delay time in a trans-

former to carry signals from primary side to the sec-ondary side.

3. Several thousand volts of isolation can be built-in be-tween windings by proper design and layouts.

The disadvantages of using transformers for Gate Driveare:

1. They can be used only for AC signals.2. Large duty ratios cannot be handled by the trans-

former without being saturated by net DC, unless ACcoupling capacitors are employed in series.

Two examples of gate Drive circuits, using transformersfollow. In Fig. (15), a phase shift controller outputs its sig-nals to the IXDD404 Dual Drivers, which in-turn, feed thetransformers. The secondary windings of these transform-ers are coupled to the Gates of upper and lower MOS-FETs in a “H” Bridge topology. Fig. (18) shows anothertransformer coupled Gate Drive circuit employing DC re-store technique to maintain same waveshape of originalsignal with added feature of -ve bias offered using a Zenerin series with a fast diode across secondary.

4.0 IXYS LINE OF MOSFET/IGBT DRIVERS

IXYS Corporation offers the following MOSFET/IGBT Driv-ers:

1. IXBD4410 and IXBD4411 Half Bridge Driver Chipset2. IXDD404 Dual 4 Amp Ultrafast Driver3. IXDD408 Single 8 Amp Ultrafast Driver4. IXDD414 Single 14 Amp Ultrafast Driver5. IXDD415 Dual 15 Amp Ultrafast Driver

4.1 IXBD4410 and IXBD4411 Half Bridge MOSFET/IGBT Driver Chipset

This chipset is best suited to applications in Half Bridge,Full Bridge and 3 Phase Bridge topologies. Here theIXBD4410 is wired as a full-featured Low-Side Driver, whileIXBD4411 is wired as a full-featured High-Side Driver. To-gether, they make up a stand alone Driver System forPhase leg of any of the above mentioned Bridge Configu-rations. The suggested wiring diagram is shown in Fig.(11). Likewise, the wiring diagram is to be repeated foreach phase leg and hence one needs two such cards for“H” Bridge and three such cards for 3-Phase Bridge.

As can be seen in this schematic, to obtain galvanic iso-lation, it uses one ferrite core transformer for sending drivesignals to IXBD4411 and another ferrite core transformerfor receiving fault and status signals from IXBD4411. T1

represents both these transformers housed in one IC typepackage. To avoid saturation, capacitors are placed in se-ries with each primary winding to which AC (time-varyingsignals) are transmitted. 1200 Volts of isolation barrier isbuilt in.

Both IXBD4410 and IXBD4411 are feature-rich Drivers. Theseinclude:

1. Undervoltage and overvoltage lockout protection for Vcc;2. dV/dt immunity of greater than ± 50 V/ns;3. Galvanic isolation of 1200 Volts (or greater) between low side and high side;4. On-chip negative gate-drive supply to ensure MOSFET/IGBT turn-off even in electrically noisy environment;5. 5 volt logic compatible HCMOS inputs with hysteresis;6. 20ns rise and fall times with 1000 pF load and 100 ns rise and fall times with 10000 pF load;7. 100 ns of propagation delay;8. 2 Ampere peak output Drive Capability;9. Automatic shutdown of output in response to overcurrent and/or short-circuit;10.Protection against cross conduction between upper and lower MOSFET/IGBT;11.Logic compatible fault indication from both low and high-side drivers.

Higher current MOSFET/IGBTs require higher drive cur-rents, especially for operating them at high switching fre-quencies. For these applications, one can use IXDD408or IXDD414, either in stand-alone mode or in conjunctionwith IXBD4410 and IXBD4411. It is easy to realize nowthat one can easily get all the facilities of feature-richIXBD4410 and IXBD4411 and when higher drive currentcapability is called for, use them in conjunction withIXDD408 or IXDD414. In case of a low-side MOSFET/IGBT, it is simple to use IXDD408 or IXDD414 alone. Onesuch example is given in Fig. (11).

For driving upper MOSFET/IGBT of a phase leg, one ofthe approaches is to employ a charge pump. Two suchapplication circuit schematics are shown in Fig. (12C) andFig. (12D). In Fig. (12C) output from IXBD4410/4411 isboosted up to ± 8 Amps using IXDD408 and the chargepump output is boosted to 350 mA, using one driver ofIXDD404 for driving IXFK48N50 (rated at Id=48 Amps andVd = 500 Volts). In Fig. (12D), the output from IXBD4410/4411 is boosted up to ±14 Amps by IXDD414 to drive aSize 9 MOSFET IXFN80N50 (rated at Id = 80 Amps andVd = 500 Volts). IXDD404 can still adequately provide500 mA for the charge pump circuit.


10

4.2 GENERAL REMARKS ABOUT IXDD SERIES OFDRIVERS

The most important strength of these Drivers is their abilityto provide high currents needed to adequately drive today’sand tomorrow’s large size MOSFETs and IGBTs. This ismade possible by devoting a large portion of the silicon diearea to creating high current (NMOS and PMOS) outputstage. Another important feature of these Drivers is thatthere is no cross conduction, thus giving almost 33% lowertransition power dissipation.

In addition, all these Drivers incorporate a unique facilityto disable output by using the ENABLE pin. With the ex-ception of the IXDD408, the ENABLE pin is tied high in-ternally. When this pin is driven LOW in response to de-tecting an abnormal load current, the Driver output entersits Tristate (High Impedance State) mode and a soft turn-off of MOSFET/IGBT can be achieved. This helps pre-vent damage that could occur to the MOSFET/IGBT dueto LdI/dt overvoltage transient, if it were to be switched offabruptly, “L” representing total inductance in series withDrain or Collector. A suggested circuit to accomplish thissoft turn off upon detecting overload or short circuit isshown in Fig. (9 ). It is also possible to do this by an inde-pendent short circuit/overload detect circuit, which couldbe a part of the PWM or other controller IC. All one needsto do is to take output signal (FAULT) from such a circuitand feed it into the ENABLE pin of Driver. A resistor R

P

connected across Gate and Source or Gate and Emitter(as the case may be) would ensure soft turn-off of theMOSFET/IGBT, turn-off time being equal to R

P(C

GS +C

GD)

Detailed specifications of these IXDD series of Driversare available on web site www.ixys.com under heading:“ICs”. All IXDD series of IXYS MOSFET/IGBT Drivers areLow-Side Drivers, but with techniques covered in this Ap-plication Note, one can use them as High-Side Drivers aswell. Evaluation Printed Circuit Boards are also available.

4.2.1 IXDD404

IXDD404 is a dual 4 Ampere Driver, which comes in handyin many circuits employing two MOSFETs or IGBTs. Itcould also be used for two MOSFETs/IGBTs connectedin phase leg, provided the voltage level is below 25 VDC.While using High Voltage DC Supplies for driving Phaseleg, H-Bridge or 3 Phase Bridge Circuits, some techniquefor achieving galvanic isolation of upper MOSFET/IGBTDrivers is required, in addition to making provision for iso-lated power supplies.

Fig. (15) shows an interesting application for IXDD404 ina Phase Shift PWM Controller application, in which gal-vanic isolation is obtained by using ferrite core Gate DriveTransformers. Note that this Controller operates at a fixed

frequency. Turn-off enhancement is achieved by usinglocal PNP transistors.

For a vast number of low and medium current MOSFETsand IGBTs, IXDD404 provides a simple answer for driv-ing them.

4.2.2 IXDD408 AND IXDD414

IXDD408 is eminently suitable for driving higher currentMOSFETs and IGBTs and IXDD414 can drive larger sizeMOSFET/IGBT Modules. Many circuit schematics apply-ing these in various topologies are possible and some ofthese are shown in different figures in this application note.

The 5 pin TO-263 surface mount version can be soldereddirectly on to a copper pad on the printed circuit board forbetter heat dissipation. It is possible then to use thesehigh current drivers for very high frequency switching ap-plication, driving high current MOSFET modules for a highpower converter/inverter.

Fig. (8) shows a basic low side driver configuration usingIXDD408 or IXDD414. C1 is used as a bypass capacitorplaced very close to pin No. 1 and 8 of the Driver IC. Fig..(13) shows a method to separately control the turn onand turn off times of MOSFET/IGBT. Turn-on time can beadjusted by R

gon, while the turn-off time can be varied by

Rgoff

. The diodes in series are fast diodes with low forwardvoltage drop. The 18V, 400mW Zener diodes protect theGate-Emitter junction of the IGBT. A careful layout of thePCB, making shortest possible length between pin Nos.6and 7 and IGBT Gate and providing generous coppersurface for a ground plane, helps achieve fast turn-onand turn-off times without creating oscillation in the Drain/Collector current.

Fig. (13) shows another arrangement and includes amethod for faster turn-off using a PNP transistor placedvery close to the MOSFET Gate and Source. It is a goodpractice to tie the ENABLE pin of drivers to Vcc through a10K resistor. This ensures that the driver always remainsin its ENABLED mode except when driven low due to aFAULT signal. Again this FAULT signal puts these twodrivers into their TRISTATE output mode.

Fig. (14) shows a method to boost output from IXDD408to a much higher level for driving very high power IGBTmodule. Here the turn-on and turn-off times can be var-ied by choosing different values of resistors, R

gon and R

goff.

To provide –ve bias of 5 Volts, the IGBT emitter is groundedto the common of +15V and –5V power supply, whichfeeds +15V and –5 v to the IXDD408. Notice that the in-coming signals must also be level shifted.


11

Fig. (16) shows an IXDD414 driving one IGBT of a Con-verter Brake Inverter (CBI) module. Here all protection fea-tures are incorporated. For High Temperature cutoff, abridge circuit is used with the CBI module’s thermistor.Comparator U3 compares voltage drop across the ther-mistor to the stable voltage from the Zener diode. P1 canbe used to preset the cutoff point at which the comparator’soutput goes low. This is fed into the Microcomputer asOVERTEMP signal.

Short circuit protection is provided by continuously moni-toring the voltage drop across a SHUNT. Note that oneend of SHUNT is connected to the power supply groundGND1. The voltage picked up from this SHUNT is ampli-fied by a low noise Op Amp and is then compared to thestable voltage from the same Zener. When short circuitoccurs, the comparator output (FAULT) goes low. 1%metal film resistors are used throughout in both these cir-cuits to ensure precision and stability. C3 and C4 help inoffering low pass filtering to avoid nuisance tripping.

Principle of DESAT sensing for detecting overload on anIGBT has been explained before in section 2.5 above. Inthe case of AC Motor Drive, each IGBT has to be pro-tected from overload using separate DESAT sensing. Fig.(16) and Fig. (17) show the connection for each IGBT.DESAT sensing is done on the isolated side of each opto-coupler, while the resultant FAULT signal is generated onthe common input side with respect to GND1. Each FAULTsignal is open collector type and hence can be tied to-gether with other FAULT signals from other opto-coupler orfrom other comparators. The Microcomputer will stop out-put drive signals when either FAULT or OVERTEMP signalgoes low. When this happens, notice that IXDD414 offersa -ve bias of -5V to guarantee turn-off conditions, even inpresence of electrical noise. -5V is applied to gate of eachIGBT during turn-off even under normal operating condi-tions. After fault is cleared, the Microcomputer can issuea RESET signal for resuming normal operation.

5.0 PRACTICAL CONSIDERATIONS

When designing and building driver circuits for MOSFET/IGBT, several practical aspects have to be taken care ofto avoid unpleasant voltage spikes, oscillation or ringingand false turn-on. More often than not, these are a resultof improper or inadequate power supply by-passing, lay-out and mismatch of driver to the driven MOSFET/IGBT.

As we understand now, turning MOSFET/IGBT on andoff amounts to charging and discharging large capacitiveloads. Suppose we are trying to charge a capacitive loadof 10,000 pF from 0 to 15 VDC (assuming we are turningon a MOSFET) in 25 ns, using IXDD414, which is a 14amp ultra high speed driver.

I = VxC / t Eq. 5.1

I = (15-0)x10000x10-12 / 25x10-9 = 6 A

What this equation tells us is that current output from driveris directly proportional to voltage swing and/or load ca-pacitance and inversely proportional to rise time. Actuallythe charging current would not be steady, but would peakaround 9.6 Amps, well within the capability of IXDD414.However, IXDD414 will have to draw this current from itspower supply in just 25 ns. The best way to guaranteethis is by putting a pair of by-pass capacitors (of at least10 times the load capacitance) of complementary imped-ance curves in parallel, very close to the VCC pin ofIXDD414. These capacitors should have the lowest pos-sible ESR (Equivalent Series Resistance) and ESL(Equivalent Series Inductance). One must keep the ca-pacitor lead lengths to the bare minimum.

Another very crucial point is proper grounding. Driversneed a very low impedance path for current return toground avoiding loops. The three paths for returning cur-rent to ground are: 1. Between IXDD414 and the logicdriving it; 2. Between IXDD414 and its own power supply;3. Between IXDD414 and the source/emitter of MOSFET/IGBT being driven. All these paths should be extremelyshort in length to reduce inductance and be as wide aspossible to reduce resistance. Also these ground pathsshould be kept distinctly separate to avoid returning groundcurrent from the load to affect the logic line driving theIXDD414. A good method is to dedicate one copper planein a multilayered PCB to provide a ground surface. Allground points in the circuit should return to the same physi-cal point to avoid generating differential ground poten-tials.

With desired rise and fall times in the range of 25 to 50ns, extreme care is required to keep lengths of currentcarrying conductors to the bare minimum. Since everyinch of length adds approximately 20 nH of inductance, adi/dt of 240 Amps/microsecond (used in the example cal-culation for Eq. 5.1) generates a transient LdI/dt voltage of4.8 volts per inch of wire length, which subtracts from thedriver’s output. The real effect will be a significant increasein rise time for every tiny increase in conductor length fromoutput pin of driver to the Gate lead of MOSFET/IGBT. Forexample, one extra inch of conductor length could increaserise time from 20 ns to 70 ns, in a ultra high speed gatedrive circuit. Another detrimental effect of longer conductorlength is transmission line effect and resultant RFI/EMI.

It is prudent to also keep in mind the fact that every MOS-FET/IGBT has some inductance depending on the pack-age style and design. The lower this value, the better is


12

the switching performance, as this inductance is, in effect,in series with the source/emitter and the resulting negativefeedback increases switching times. IXYS MOSFET/IGBTsare housed in packages, which have extremely low intrin-sic inductance.

While applying driver IC for any application, it is also nec-essary to compute power dissipated in the driver for a worstcase scenario. The total power dissipated in the driver ICis a sum of the following:

1. Capacitive load power dissipation;2. Transition power dissipation;3. Quiescent power dissipation.

For all IXDD series of drivers, transition power dissipa-tion is absent due to a unique method (Patent pending) todrive the output N-Channel and P-Channel MOSFETs, prac-tically eliminating cross conduction.

As described under section 1.4, a MOSFET/IGBT driverincurs losses. Let us derive formulae to compute thispower loss in a driver:

PD(on) = D x ROH x Vcc x Qg x fsw Eq. 5.2 ROH + RGext + RGint

PD(off) = (1-D) x ROL x Vcc x Qg x fsw Eq. 5.3 ROL + RGext+ RGint

where: ROH = Output resistance of driver @ output HighROL = Output resistance of driver @ output Lowfsw = Switching frequency

RGext = resistance kept externally in series with Gate of MOSFET/IGBT RGint = Internal mesh resistance of MOSFET/ IGBT D = Duty Cycle ( Value between 0.0 to 1.0) Qg= Gate Charge of MOSFET/IGBT

Total loss PD = PD(on) + PD(off)

Note also that in general, RGint

is small and can be ne-glected and that ROH = ROL for all IXDD drivers. Conse-quently, if the external turn-on and turn-off gate resistorsare identical, the total driver power dissipation formula sim-plifies to:

PD = PD(on) + PD(off) = ROH x Vcc x Qg x fsw Eq. 5.4 ROL + RGext

Let us review some examples.

1. Assuming that, we are driving an IXFN200N07 for a Telecom power supply application or for a UPS/Inverter application at a switching frequency of 20 kHz

2. RGext = 4.7 Ohms and gate supply voltage is 15V.

On page No. 2 of the IXDD408 Data sheet, we read thevalue of ROH = 1.5 Ohms ( Maximum). For Qg, refer toData Sheet of the IXFN200N07 from www.ixys.com and goto Gate Charge vs. V

GS curve and look for value of Qg at

Vcc = 15 V. You can read it as 640 nC. Substituting thesevalues into Eq. 5.4 yields:

PD = 1.5 x 15 x 640 x 20,000 x 10-9

1.5 + 4.7PD = 46.45 mW

Assuming an ambient of 50 oC in the vicinity of IXDD408PI,the power dissipation capability of IXDD408PI must bederated by 7.6mW/oC, which works out to be 190 mW.The maximum allowable power dissipation at this tem-perature becomes 975-190=785 mW. However, as cal-culated above, we will be dissipating only 46.45 mW sowe are well within the dissipation limit of 785 mW.

If one increases fsw to 500 kHz for a DC to DC Converterapplication, keeping other parameters the same as above,now the dissipation would be 1.16 W, which exceeds thespecification for IXDD408PI. So in this case, it is recom-mended to use either the IXDD408YI (TO-263 package)or IXDD408CI (TO-220) package. Both these packagescan dissipate about 17 W with proper heat sinking ar-rangement. The TO-263 is a surface mount package andcan be soldered onto a large pad on a copper surface ofa PCB for achieving good heat transfer. For TO-220 pack-age, a heat sink can be employed.

Let us take another example of a boost converter, usingIXFK55N50 at V

DS = 250 VDC and at I

D = 27.5 Amps. As-

suming fsw = 500 kHz, Vcc = 12 V. From the curve ofGate Charge for IXFK55N50 in the Data Sheet one candetermine that Qg = 370 nC. Let us set RGext = 1.0 Ohm.We use IXDD414YI or IXDD414CI here, which can dissi-pate 12W. Here typical value of ROH = ROL = 0.6 Ohm.Substituting the above values in our equation, we com-pute the power dissipation to be:

PD = 0.6 x 12 x 370 x 500 kHz x 10-9

0.6 + 1.0 + 0.0PD = 0.83 W.

With adequate air circulation, one may be able to use thePDIP Package.

For the third example, considering driving the large sizeMOSFET module VMO 580-02F at fsw = 250 kHz. LetVcc = 10 V, ROH = ROL = 0.6 Ohm, RGext= 0.0 Ohm. Weread that Qg =2750 nC at Vcc = 10 V off the VMO 580-02F data sheet. Now:


13

PD = 0.6 x 10 x 2750 x 250kHz x 10-9

0.6+ 0.0 + 0.0PD = 6.86 W

IXDD414YI (TO-263) or IXDD414CI (TO-220) can easily drivethis load provided adequate heatsinking and proper air flowis maintained. Comments above for mounting TO-263 and/or TO-220 packages apply here as well. For derating use0.1 W/oC. So for an ambient temperature of 50 oC, it worksout to be 2.5 W. As the limit of IXDD414YI or IXDD414CI is12 W, subtracting 2.5 W from this yields 9.5 W. So 6.86W is still possible. Thermal Impedance (Junction to Case)is 0.55oC/W for TO-263 and TO-220, hence a rise in casetemperature should be within limit. If we increase Vcc to15V, conduction losses in MOSFET could reduce due tolower RDS(on), but obtaining the same rise and fall timeswill incur more power loss in driver due to increased Vccand Qg. If that happens, approach described in Fig. (14)can be employed.

6. CONCLUSION

With proliferating applications of modern power electron-ics worldwide, faster, more efficient and more compactMOSFETs and IGBTs are replacing older solid state andmechanical devices. The design of newer and more effi-cient techniques to turn these solid state devices on andoff is a subject that requires thorough study and under-standing of the internal structure and dynamic processesinvolved in the working of MOSFET/IGBTs.

Main emphasis in modern Power Electronics is to reducetotal losses dissipated in devices and subsystems for higheroperating efficiency and achieving more compact designs,reducing volume and weight of resultant systems. Thus,operation at higher and higher switching frequencies is nowa necessity, and as a result, switching losses predomi-nate in the power-loss-budget in semiconductor switches.Reducing switching losses then becomes the single mostcrucial goal. Keeping this goal in mind, the entire line ofIXYS MOSFET/IGBT Drivers are designed to facilitate thedesign of drive circuits that yield fast rise and fall times,matching or often exceeding the speeds of the drivenMOSFET/IGBTs. Extremely low propagation delay timefacilitates implementation of fast overload/short circuit pro-tection.

With the advent of IC Drivers for these fast MOSFET/IGBTs, the designer is relieved of the tedious task of de-signing elaborate driver circuits. Nevertheless, under-standing these newer ICs, their strengths and limitations,is of paramount importance. Different configurations forparticular topologies call for specific application knowl-edge. Illustrations are the best way to explain theory and

applications of these IC drivers. Practical use of these ICdrivers call for great care for achieving near theoreticalresults.

References:

1. B. Jayant Baliga, “Power Semiconductor Devices”,PWS Publishing Company, Boston, MA (1996)

2. Ned Mohan, Tore M. Undeland, William P. Robbins:“POWER ELECTRONICS: Converters, Applicationsand Design”, John Wiley & Sons, New York (1994)

3. Power Supply Design Seminar - 2001 series, UnitrodeProducts from Texas Instruments.

Recommended for further reading:

1. George J. Krausse, “Gate Driver Design for Switch-Mode Applications and the DE-SERIES MOSFET Tran-sistor”, Directed Energy, Inc. Application Note availablefrom<www.directedenergy.com>.

DISCLAIMER:

Although information furnished herein is believed to beaccurate and reliable, IXYS Corporation assumes noresponsibility for its use; nor for any infringement of pat-ents or other rights of third parties which may resultfrom its use.


14

X2

34 4 51

4 6

DD

7

1 I 8I/P +C9

R7

Vcc : 3V to 25VDC

U5

VDC

Ld=10 Hm

Rd=0.1 O hm

U4 U5R5

C7

U2

U2U5

MAINS

T1 D1

D2

C1C2

+ +7815

1

2 +C2

3 +15V TO

LM-317

R1

P1

1 3

2

P2

R3R2

Z1

R6

R9 Rs=0.005 O hm

Ls=20nH

M OSFET

C4

M ODULEVM O580-02F

U32

4

5+

-C3

+15V

R4

Fig. (9). Evalua tion circuit to test IXDD408/IXDD414 for soft turn off.

Q1

ALL ICs

(Representingstray inductance)

C 1,C 3 : 22 M FD , 25VD C Tantalum capacitors

D 1

R G

R p ZD 1ZD 2

R S

C SC

E

GQ

+15V D C

H .V.D C

C 1

R 1

12

3

4 5

6

78

V

TIM E

IN PU TIC 2

D 2

D 3

C 2IC 1

C 3

T1

+15V D C

+

C .T.

C 2 : 2200 M FD , 35VD C E lectro lytic capacitorsT1 : 220 VAC to 15-0-15 VAC , 15VA contro l transfo rm er

110VAC TO 15-0-15

D 2,D 3 : IN 4002

IC 2 : IXD D 408 or IXD D 414D 1 : IN 5817

IC 1 : 7815 R egu lator

C s,R s : Snubber netw ork to reduce IGB T sw itching R p : 2K2, ¼ W, 5%

R G 1 : 3 .3 ohm s to 27 ohm s ZD 1,ZD 2 : 18V, 400M W Z EN ER S

C s=0.1 M FD , R s=10 to 33 ohm s

Fig. (8) C ircuit schem atic show ing how to use IXD D 408 or IXD D 414 to drive an IG BT

losses. Value depends upon fsw. Suggest:

depending on Turn-O N speed

LOAD

Q : IX LF19N 250

R 1 : 10K , ¼ w

A


15

Bill of Materials for Fig.(9)

Resistors:R1: 240R2: 560R3: 10KR4: 5KR5: 1MegR6: 1K5R7: Rg-T.B.D.R8: 1Meg

Capacitors:

C1: 1000mF;35VWDCC2: 22mF, 63 VWDCC3: 1pF, silver dipped micaC4: 100pF silver dipped micaC5: 0.1mF, 35WDC TantalumC6: 0.1mF, 35VWDC TantalumC7: 1pF silver dipped micaC8: 0.1mF, 35VWDC TantalumC9: 0.1mF, 35VWDC TantalumC10: 0.1mF, 35VWDC Tantalum

Diodes:D1: 1N4002 or BA 159D2: 1N4002 or BA 159

Zener Diodes:1. Z1: 1N821

Voltage Regulators:1. 78152. LM317T

Transistors:1. Q1: 2N7000

ICs:U1: IXDD408PI or IXDD414PIU2: CD4001U3: LM339U4: CD4011U5: CD4049U6: IXDD408YI or IXDD414YINote: Either use either U1 or U6 but not both.

Trimmers:P1: 5K, 3006P Bourns or SpectrolP2: 1K, 3006P Bpurns or Spectrol


16

Com

pone

nt S

ide

IXDD408-EV-A IXDD408-EV-A

IXDD414EV-A IXDD414EV-A

Solder Side


IXDD408-EV-A

IXDD408-EV-A

CN

1

P2

P1

U1

C1

D2

D1

R1

C12

C2

T1

C7

C5

U4

C6

T2

C11

U2

C10R3R2

Z1R5CN

2

U5

C8

Q1

C4

C3

R6

R7

R9R4

U3

C9

R8

31 2

CN

3


IXD

D40

8-E

V-A

IXD

D40

8-E

V-A

Fig. (10). +ve and -ve and component layout with silk sceen diagram.


17

Fig

. 11.

IXB

D44

10/4

411

Eva

luat

ion

Boa

rd S

chem

atic

.


18

Fig. (12A) Basic Charge Pum p Doubler

Vcc+

-

Q1

Q2

+-

C1

SD1

C2 -

+

SD2

18VZener

+2xVcc -V SD2

ON

OFFt

f sw~~ 400 KHz

ON

OFFt

F ig . (12B) Basic bootstrap ga te d rive techn ique

1

4

3

2

8

5

7

44

80

41

DD or

XDD

X

6

I I

Q 1

R1

Q 4

DB

Rgext

LOAD

D1

V DC

CB

I/P

t

V

CF

Vcc = +15V

L

P1 R4

+

D2

~~ 500VD C

C2

Q 1 : IXTU01N 100DB : DSEP9-06CRD1 : DSEC60-12AD2 : 1N 5817C1 : 20M FD,25VC2 : 20M FD,1000 Volts ,C SI 10DC 0020CB : 10M FD ,25VD CP1 : 5K Trim pot

R3

C1

0

15V

R2

R5

Rg 1 .0 Ohm to 4 .7 Ohmext:

CF : G E A28F5601 0.1M FD,1000 Volts

R1 : 1KR2 : 10KR3 : 2KR4 : 10KR5 : 1 O hmL : 5µH ,DALE IH-5


19

Fig

. (12

C).

Boo

stin

g ou

tput

gat

e dr

ive

to +

/-8

A a

nd c

harg

e pu

mp

outp

ut to

350

mA

for

400k

Hz

switc

hing

of s

ize

9 de

vice

sw

ith th

e IX

BD

4410

/441

1 ga

te d

river

chi

p se

t.


20

Fig

. (12

D).

Boo

stin

g ou

ptpu

t gat

e dr

ive

to +

/-14

A a

nd c

harg

e pu

mp

outp

ut to

500

mA

for

400k

Hz

switc

hing

of s

ize

9 de

vice

sw

ith th

e IX

BD

4410

/441

1 ga

te d

river

chi

p se

t.

IXY

S C

orporation; 3540 Bassett S

treet; Santa C

lara, CA

95054; Tel: 408-982-0700; Fax: 408-496-0670

IXY

S S

emiconductor G

mbH

; Edisonstr. 15; D

-68623; Lampertheim

, Germ

any; Tel: +49-6206-503-0; F

ax: +49-6206-503627

21

1

2

3

4

8

7

6

5

IXDD408

or

4

41

DDXI

F ig (13 ) Tu rn -o ff enhancem ent m ethods.

Vcc

I/P

D

C F R g+

LOAD

H.V.DC

I/P 2 7D D

F

or

4 8

3 40

D

54

41

D6

R g

Q 2

1XI

XI

8C

+

V cc

O

DA

H .V.D C

L

DV

t

on

Z1

Z2

Z1

Z2

D on

offRg

F ig . (14). Techn ique to boos t curren t outpu t and provide -ve b ias to ach ieve faste r tu rn o ff fo r h igh pow er M O S F ET and IG B T M odu les

D

04 8

3

2

4D

5

6

7 R b

1 IX 8

+

FC

+V cc=+15V

Q 2

LO A D H .V.D C

V = -5VE E

V

+15V

t

-5V

Q 1R g on

R g off

IG B T

0.0V

I/P

Z 2

Z 1 1. Z1 ,Z2 : 18V,400m W Zener d iodes2. D : 1N 58213. Q 1 : D 44VH 104. Q 2 : D 45VH 105. R b : 10 O hm s,1 /4w,1%

IXY

S C


treet; Santa C

lara, CA

95054; Tel: 408-982-0700; Fax: 408-496-0670

IXY

S S

emiconductor G

mbH

; Edisonstr. 15; D

-68623; Lampertheim

, Germ

any; Tel: +49-6206-503-0; F

ax: +49-6206-503627

22

Fig. (15) Transfo rmer coupled Gate D rive a rrangem ent for "H" Bridge in a Phase Shift PW M Controller at fixed Sw itching frequency.

D

04 4

3

2

4D

5

6

7

1 IX 8

72

4

3

4 5

40

D6

1XD

I8

P h aseS h iftP W MC on tro lle rICU 1

A

B

C

D

V cc=16 V D C

D 1

R 6

R 5

R 8

R 7

Q 1

T1

R 1

R G 1

G 4R

R 4

D 4

Q 4

Z1

Z2

Z8

Z7

FC+

L O A D

T3

T2

Z5

Z6

Z 4

Z 3

D .C . S U P P LY ~ 3 0 0 to 37 5 V.D .C .~+V E

Q 2

D 2

R 2

G 2R

M 1 M 2

M 4 M 3

Q 3R 3

R G 3

D 3

D .C .S U P P LYC O M M O N

SU GGESTE D PARTS: 1. U1 : T.I. UC38792. T1,T2 : Coilcraft Part No. SD250-33. Q1 ,Q 2,Q 3,Q 4 : 2N2905A4. D1,D2,D3,D4 : DSEP8-02A IXYS H iPerFRED5. T3 : O UTPUT Transform er6. CF : 22MFD,35 V W DC Tan talum Capacitor7. R1,R2,R3,R4 : 560 Ohm s, ¼ w,1% Metal film8. M 1,M 2,M 3,M 4 : IXFN55N50 IXYS H iPerFET or IXFN80N50 IXYS H iPerFET9. Z1,Z2,...Z8 : 18V, 400mW Zener diodes.

10. R G1,RG2,RG 3,RG 4 : 3.3 O hm s, ¼ w, 1% M eta l F ilm resistors.11. R5,R6 ,R7,R8 : 10K , ¼ w, 5%


23

S in eW a veP W M

S ig na lsfo r

3 -P h as eIn ve rte r

D y na m icB rake

MIC

RO

CO

NT

RO

LLE

R

U 1

F ig. (16) 3-Phase AC M otor d rive schem atic show ing how IX YS C B I (C onverter-B rake-Inverter) M odule can be driven by IXD D 414 us in g opto-coup lers. A ll p ro tec tion features are incorporated.

FA U LT

T 1

T 2

T 3

T 4

T 5

T 6

T 7

R E S E T

O V E RT E M P

G N D 1

+ 5.0 V

G N D 1

1

2

3

4

5

6

7

8

1 6

1 5

1 4

1 3

1 2

11

1 0

9U 6

H C P L-316J

R E S E T

FA U LT R 5

V in

V in

+

-

Vcc 1G N D 1

VL E D 1

L E D 1V

+

-

VE

L E D 2+V

D E S AT

cc 2V

cV

O U TV

E EV

E EVR 6

-5V

IX D D 41 4 P IU 2

ccV O/P

GR

OU

ND

I/P EN

AB

LE

IS O LATE DD C TO D C

C O N V E R T E R

+ 15V C O M -5V

+ +

C 1 C 2

G N D 1 + 5.0V

+ 5.0 V

1 2 3

D 1 6D 1 4D 1 2

D 11 D 1 3 D 1 5

1 4

7

FL

CF

11

1 6

D 1 5

2 22 1

1 5

D 1

D 2

1 2

1 7

1 8

D 4

D 3

R g

D 5

D 6

1 3

1 9

2 0

T 1 T 3 T 5

T 6T 4T 2

IX Y S 's (C B I)C O N V E R T E RB R A K EIN V E RT E RM O D U LE

U 3

-

+

NTC

R 4

R 3

8

9

R 2

Z 1

U 4

-

+ +

-U 5

S H U N T

R 7

C 3

R 8

G N D 1 C 4

R 9

2 3 2 4

P 1

+ 5V

R 11FA U LT

R 1

R 1 2

R 1 3

R 1 0

R

D d

D


24

D12

D14

D16

D11

D13

D15

U V W

3ø M

AIN

S

MC

B

1

2

3

DC

to D

C

IXD

D41

4

+15

V-5

V

HC

PL3

16J

16 15

T1

F R T1

NO

TE

S: 1

. ALL

F =

FA

ULT

SIG

NA

LS A

RE

TIE

D T

OG

ET

HE

R (

BE

ING

OP

EN

CO

LLE

CT

OR

) A

ND

FE

D IN

TO

MIC

RO

CO

MP

UT

ER

.2.

ALL

R =

RE

SE

T S

IGN

ALS

AR

E T

IED

TO

GE

TH

ER

AN

D F

ED

TO

HC

PL-

316J

.3.

OV

ER

TE

MP

AN

D O

VE

RLO

AD

/SH

OR

T C

IRC

UIT

FA

ULT

SIG

NA

LS A

RE

GE

NE

RA

TE

D A

S P

ER

FIG

(16)

OV

ER

TE

MP

IS A

LSO

FE

D IN

MIC

RO

CO

MP

UT

ER

.

FIG

(17)

IXY

S C

ON

VE

RT

ER

, BR

AK

E IN

VE

RT

ER

(C

BI)

MO

DU

LE B

EIN

G D

RIV

EN

BY

IXD

D41

4 W

ITH

OP

TO

-CO

UP

LER

AN

D D

ES

AT,

OV

ER

TE

MP

AN

D S

HO

RT

CIR

CU

IT/O

VE

RLO

AD

PR

OT

EC

TIO

NS

.

+5V

D1

2122

LF

2324

CF

HC

PL3

16J

IXD

D41

4

14

DC

to D

C+5V

T7

7

T3

18 17

IXD

D41

4

HC

PL3

16J

+15V

-5V

D3

+5V

DC

to D

CT

3D

C to

DC

HC

PL3

16J

IXD

D41

4

T5

+15

V-5

V

1920

D5

+5V

T5

T2

-5V

IXD

D41

4

HC

PL3

16J

DC

to D

C

HC

PL3

16J

T4

1011

DC

to D

C

IXD

D41

4

+15

V+1

5V-5

V-5

V

+5V

D2

T2

+15V

T6

HC

PL3

16J

-5V

T4

12

+5V

+15

V

D4

IXD

D41

4

DC

to D

C+5V

13

T6

D6

+

M

4

5

6 MIC

RO

CO

NT

RO

LLE

R

T1

T2

T3

T4

T5

T6

T7

RF

OV

ER

TE

MP

U V W

F F F

GN

D1

SH

UN

T

T7

RF

RF RF

RF

RF

RF

Rg

Rg

Rg

Rg

Rg

Rg

Rg

RE

GIS

TOR

BR

AK

ED

YN

AM

IC

DE

SAT

DE

SAT

DE

SA

TD

ES

AT

DE

SA

TD

ES

AT

3 A

.C.

MO

TO

R

+5V

Dd

DR

Dd

D

Dd

D

Dd

D

Dd

D

Dd

DR

RRR R


25

Bill of Materials for Fig. (16) and Fig. (17)

R1, R3, R5, R10, R11: 10K, 1/4W, 1% MFRR2: 560 Ohms, 1/4W, 1% MFRR4: 2.2 MegOhms, 1/4W, 5%R6: 100 Ohms, 1/4 W, 1% MFRR7: 20K, 1/4W, 1% MFRR8, R9: 61.9K, 1/4W, 1% MFRR12, R13: 1.24K, 1/4W, 1% MFRRg: T.B.D. based on Ton and Toff & size of IGBTRD:100 Ohms,1/4 w, 5%P1: 10K, multi turn trimpot, Bourns 3006P or

SpectrolC3, C4: 33 pF, silver dipped micaDd: General Semiconductor make,Type: RGP02-20E, 0.5 A, 2000 V, trr: 300 nsZ1: Zener LM336, 2.5 VoltU3, U5 : LM339 ComparatorU4 : LM-101 Op AmpSHUNT : 75 mV @ full load currentLF: Gapped D C Choke for filtering rectified

powerCF: Electrolytic Filter Capacitor with very low

ESR & ESL and screw type terminals tohandle high ripple current. Voltage ratingis based on DC Bus plus AC ripple Voltage

CBI Module: IXYS Corporation Type Nos:MUBW 50-12A8 or any MUBW module fromCBI 1, CBI 2 or CBI 3 series.Microcontroller: T.I. TMS320F240 with embedded software for AC Drive, using brake feature.IXDD414 Driver chip: 7 are required to imple-ment the A.C.Drive, using Brake feature.HCPL316J( Opto-coupler) : 7 are required toimplement the A.C.Drive With Brake feature.Isolated DC to DC Converter: 7 are requiredwith specified isolation.

IXY

S C


treet; Santa C

lara, CA

95054; Tel: 408-982-0700; Fax: 408-496-0670

IXY

S S

emiconductor G

mbH

; Edisonstr. 15; D

-68623; Lampertheim

, Germ

any; Tel: +49-6206-503-0; F

ax: +49-6206-503627

26

FIG (18) A Transform er coupled G ate Drive c ircuit em ploying D .C . restore techn ique and show ing how to generate -ve bias during turn-off.

H72

4

3

5

OR6

1 8

Vcc

T1

H.V.D.C.

H .V.D.C.

LOAD

D2

Z1

T3

T2

I/P

8

D

04D

IX

4

D

14D

IX

R1 C1

D1

C2

RpD2

Z1Rp

C2

D2

Z1Rp

C2

D2Rp

Z1

C2

CO MMON

Q4

Q1 Q 2

Q 3

1 8

L

I/P

4 4

4 8

30

4 51

6

2 DD

XDDOR

7

X

D1

R1 C1

I I

Vcc

+

+C F

t

V

t

V

R H

C F

R H


27

N.A. Corporate SalesSanta Clara, CAPH: 408-982-0700FX: 408-748-9788

AlabamaNew Age Electronics, Inc.Huntsville, AL256-885-0755

AlaskaAdvanced Technical Mktg,Redmond, WA425-869-7636

ArizonaEDSTempe, AZ602-438-1547

EDSTucson, AZ520-322-6945

ArkansasNova MarketingMounds, OK918-827-5560

California (North)N. A. Corporate Sales408-982-0700

California (South)Select ElectronicsSanta Ana, Ca714-648-0181

SCASan Diego, CA858-792-1101

ColoradoAKI EnterprisesDenver, CO303-756-0700

ConnecticutSJ-New EnglandNaugatuck, CT203-723-4707

DelawareS-J Mid-AtlanticMt. Laurel, NJ856-866-1234

FloridaCDMBoynton Beach, FL561-740-3335

CDMOrlando, FL407-522-5808

CDMPalm Harbor, FL727-937-1814

CDMTampa, FL813-886-9721

GeorgiaNew Age Electronics, Inc.Norcross, GA919-872-2562

IdahoAdvanced Technical Mktg.Beaverton, OR503-643-8307

Illinois (North)Martan, Inc,Schaumburg, IL847-330-3200

Illinois (South)AEM, Inc.St. Louis, MO314-298-9900

IndianaR.O. Whitesell & Assoc.Ft. Wayne, IN219-489-0588

R.O. Whitesell & Assoc.Indianapolis, IN317-876-9000

R.O. Whitesell & Assoc.Kokomo, IN765-457-9127

IowaAEM, Inc.Cedar Rapids, IA319-377-1129

KansasAEM, Inc.Olathe, KS913-393-4100

KentuckyR.O. Whitesell & Assoc.Lexington, KY859-277-4904

LouisianaTechnical Marketing, Inc.Carrollton, TX972-387-3601

MaineSJ-New EnglandN.Billerica, MA978-670-8899

MarylandSJ ChesapeakeFalls Church, VA703-533-2233

MassachusettsSJ-New EnglandN.Billerica, MA978-670-8899

MichiganR.O. Whitesell & Assoc.Farmington Hills, MI248-473-5454

R.O. Whitesell & Assoc.Clarkston, MI248-620-9640

R.O. Whitesell & Assoc.Jenison, MI616-457-1717

MinnesotaR.W. Marshall Co., IncSt. Paul, MN651-287-0011

MississippiNew Age Electronics, Inc.Huntsville, AL256-885-0755

MissouriAEM, Inc.St. Louis, MO314-298-9900

MontanaAKI EnterprisesDenver, CO303-756-0700

NebraskaAEM, Inc.Lincoln, NE402-475-4660

Nevada (North)N.A. Corporate Sales408-982-0700

Nevada (South)EDSTempe, AZ602-438-1547

New HampshireSJ-New EnglandN.Billerica, MA978-670-8899

New Jersey (North)SJ Associates, Inc.Rockville Center, NY516-536-4242

New Jersey (South)S-J Mid-AtlanticMt. Laurel, NJ856-866-1234

New MexicoEDSTucson, AZ520-322-6945

New York (Metro)SJ Associates, Inc.Rockville Center, NY516-536-4242

North CarolinaNew Age Electronics, Inc.Raleigh, NC919-872-2562

North DakotaR.W. Marshall Co., Inc.St. Paul, MN651-287-0011

OhioR.O. Whitesell & Assoc.Dayton, OH937-298-9546

R.O. Whitesell & Assoc.Cleveland, OH216-661-2613

R.O. Whitesell & Assoc.Columbus, OH614-888-9396

OklahomaNova MarketingMounds, OK918-827-5560

OregonAdvanced Technical Mktg.Beaverton, OR503-643-8307

Pennsylvania (East)S-J Mid-AtlanticMt. Laurel, NJ856-866-1234

Pennsylvania (West)R.O. Whitesell & Assoc.Cleveland, OH216-661-2613

Rhode IslandSJ - New EnglandN.Billerica, MA978-670-8899

South CarolinaNew Age Electronics, Inc.Raleigh, NC919-872-2562

South DakotaR.W. Marshall Co., Inc.St. Paul, MN651-287-0011

TennesseeNew Age Electronics, Inc.Huntsville, AL256-885-0755

TexasTechnical Marketing, Inc.Austin, TX512-343-6976

Technical Marketing, Inc.Carrollton, TX972-387-3601

Technical Marketing, Inc.Houston, TX713-783-4497

Texas (Panhandle)EDS520-385-6945

UtahAKI EnterprisesAmerican Fork, UT801-763-0780

VermontSJ - New EnglandN.Billerica, MA978-670-8899

VirginiaSJ ChesapeakeFalls Church, VA703-533-2233

WashingtonAdvanced Technical Mktg.Redmond, WA425-869-7636

Washington D.C.SJ ChesapeakeFalls Church, VA703-533-2233

West VirginiaR.O. Whitesell & Assoc.Columbus, OH614-888-9396

WisconsinMartanMequon, WI262-241-4955

WyomingAKI EnterprisesDenver, CO303-756-0700

Puerto RicoCDMMeredita787-844-3840

CanadaCMT-Renmark, Inc.Mississauga, Ontario905-612-0900

CMT-Renmark, Inc.Ottawa, Ontario613-596-4200

CMT-Renmark, Inc.Pointe Claire, Quebec514-694-6088

VancouverCMT-Renmark, Inc.Calgary, Alberta403-225-3033

MexicoEDSTucson, AZ520-322-6945

Argentina, BrazilColombia(Stocking Reps)

Hitech ElectronicaBrazil+5511-882-4000

Etek de Colombia S.A.Colombia+571-635-3700

Etek Electronics(U.S. Office)Miami, FL+305-513-3982

North & South American Sales Representatives


28

AustraliaBraemac PTY. Ltd.1/59-61 Burrows RoadAlexandria NSW 2015Phone: (02) 9550 6600Fax: (02) 9550 6377

AustriaABB KomponentenDivision der Industrie &Gebäudesysteme GmbHWienerbergstraße 11BBusiness Park ViennaA-1810 WienPhone:01-60 10 9-6153Fax: 01-60 10 9-8600

ChinaGangli Electronics Co.Rm. 722; 7/F InternationalBldg; No. 20 Ixnfeng StreetChongzhou CityPhone: 86-519-8116388Fax: 86-519-8104942

HIP Electronics Corporation9/F, Lanyang Mansion200th Beidajie StreetChangzhou, Jiansu, 213003Phone: 0086-519-6800000Fax: 0086-519-6688597

Sunguard Electronics Ltd.Room 3117, EverbrightInternational Trust MansionNo. 11, Zhong GuanCunSouth RoadDistrict Handian; BeijingPhone: (86) 10 6847 2670Fax: (86) 10 6847 3759

Sunguard Electronics Ltd.Room 5-16, Post Bldg. 79Xiao Xin StreetSha Ping Ba DistrictChong QingPhone: (86) 23 6533 4678Fax: (86) 23 6533 4678

Sunguard Electronics Ltd.10-D, Block B, Citic Pen-XMetropolis No. 1089 Pudong AvenueShanghaiTel: (86) 21 6853 3587Fax: (86) 21 6853 3334

Sunguard Electronics Ltd.Room 7C, 7/F Hao GongBldg.; 5 Yan Nan Rd;ShenzhenPhone: (86) 755 3228 553Fax: (86) 755 3219 505

DenmarkC-88 A/SSavsvinget 7DK-2970 HørsholmPhone:+45 70 10 48 88Fax: +45 70 10 48 89

FinlandTeamtron OYTikkurilantie 68 C01300 VantaaPhone:358-9-8571770Fax: 358-9-85177600

FranceARCEL2, rue des Aulnes69410 Champagne au Mont-d'OrPhone:478 35 02 21Fax: 478 35 69 54

Eurocomposant S.A.144 Avenue Joseph Kessel78960 Voisins le BretonneuxPhone:1-30 64 95 15Fax: 1-30 43 68 27

GermanyIXYS Semiconductor GmbHEdisonstr. 1568623 LampertheimPhone: 06206/503 293Fax: 06206/503 627

Bernd MeyerFurkastr. 6712107 BerlinPhone: 030/74 79 25 30Fax: 030/74 79 25 31

Dietrich SchurichtGmbH & Co. KGRichtweg 3228195 BremenPhone: 0421 36 54 54Fax: 0421 36 54 291

Hubert Schroeter KGSaseler Bogen 122393 HamburgPhone: 040/6 00 006-0Fax: 040/6 00 006-30

Future ElectronicsDeutschland GmbHOskar-Messter-Straße 2585737 IsmaningPhone: 089/95 72 70Fax: 089/95 72 71 29

Peter KrausSorpestraße 246049 OberhausenPhone: 0208/88 33 99 2Fax: 0208/88 33 99 3

Reiner NeufRhönstr. 397791 ObersinnPhone: 09356/97 20 20Fax: 09356/97 21 21

Hartmut HauberWahlwiesenstraße 371711 SteinheimPhone: 07144/28 15 03/04Fax: 07144/28 15 05

MSC Vertriebs GmbHIndustriestraße 1676297 StutenseePhone: 07249/910-0Fax: 07249/910-215

Future ElectronicsDeutschland GmbHOskar-Messter-Straße 2585737 IsmaningPhone: 089/95 72 70Fax: 089/95 72 71 29

GreeceMarkides Michelis & Co8, Markris Street, Aegaleo122 41 AthensPhone:1-5 98 01 45Fax: 1-5 90 98 33

HongkongKARIN Electronic Supplies1/F, Karin Bldg166 Wai Yip StreetKwun Tong, KowloonPhone: 852-2389-8252Fax: 852-2343-6479

Rêvel Electronics Co. Ltd.Unit 2801A, Level 28Tower 2, Metroplaza, 223Hing Fong RoadKwai Chung, New TerritoriesPhone: 852-2389-8891Fax: 852-2389-2448

Sunguard Electronics Ltd.Room 1801, APEC Plaza49 Hoi Yuen RoadKwun Tong, Hong KongPhone: (852) 2191 3181Fax: (852) 2191 3562

IndiaPankaj ElectronicsB-276, Chittranjan ParkNew Dehli-110019Phone: 91 11 6211049Fax: 91 11 6433494

IndonesiaSee Winedge, Singapore

IsraelAvnet Gallium Co. LTD11 Hasadna St., P.O.B. 2552IL 43650; RA’ANANAPhone: 972 9 74 82 182Fax: 972 9 74 84 046

ItalyABB DACOM SpAViale Edison, 5020099 Sesto San GiovanniMilanPhone:02-26 232.125Fax: 02-26 232.144

JapanJ Rep Corporation2-1 Toyotsuchu564-0051 Suita OsakaPhone: 0081 6 6368 2111Fax: 0081 6 6368 2114

Unidux Inc.5-1-21 Kyonan-choMusashino-shi, Tokyo 180Phone: 0422-32-4500Fax: 0422-32-2050

KoreaKisung International Co. Ltd.DA-3425,JoongangYootong,1258Kuro-Dong, Kuro-Ku, SeoulPhone: 82-2 2634-9364Fax: 82-2 2631-9363

Daan Electronic Co., Ltd.Daan Bldg, 4F22-1 Singye-DongYongsan-Ku, SeoulPhone:82-2-718-8033Fax: 82-2-718 1160

MalaysiaSee Winedge, Singapore

Netherlands / BeneluxABB Componenten B.V.Lylantse Baan 92908 LG Capelle a/d IJsselPhone:010-2 58 22 50Fax: 010-4 58 65 59

New ZealandBraemac, Pty. Ltd.48 Enterprise StreetBirkenhead, AucklandPhone: 64-9-486-5200Fax: 64-9-486-4460

NorwayHenaco A/STrondheimsveien 436Po. box 126, Kalbakken0902 Oslo 9Phone:2-216 21 10Fax: 2-225 77 80

PhillipinesSee Revel ElectronicsHong Kong

PolandP.U.H. VoltexUl. Michaelowskiego 2059-300 LubinPhone:+48 76 841 1289Fax: +48 76 846 0068

SingaporeWinedge Electronics Ltd.BLK 4008 #03-13, Ang MoKio Ave 10Techplace 1, SingaporePhone: 65-455-6311Fax: 65-455-6811

South AfricaAvnet Kopp (Pty) LtdP. O. Box 3853, Rivonia,2128 South Africa.Phone: +27 11 444 2333Fax: +27 11 444 7778

Advanced Product Technol-ogy (Pty) LtdP. O. Box 832, 2160Ferndale, South Africa.Phone: +27 11 792-6010Fax: +27 11 792-987

SpainRectificadores Guasch S.A.Componentes y Electronicade PotenciaCiutat de Granada, 8008005 BarcelonaPhone:93-3 09 88 91Fax: 93-3 00 18 41

SwedenPelcon Electronics ABGirovägen 1317562 JärfällaPhone:+46 8 506 498 70Fax: +46 8 506 498 99

SwitzerlandABB Normelec AGBadenerstr. 7908048 Zürich-AltstettenPhone:1-4 35 67 17Fax: 1-4 35 66 06

TaiwanIndustrade Co., Ltd.6F, No. 64, Section 2Chung Cheng RoadShihlin, Taipei, Taiwan 111Phone: 886-2-2836 90 11Fax: 886-2-2835 30 37

ThailandSee Winedge, Singapore

TurkeyÖzdisan Elektronik A.S.Galata KULESI SokakNo: 34Karaköy/IstanbulPhone:90-212-249 98 06Fax: 90-212-244 59 43

United KingdomIXYS Semiconductor GmbHProvidence HouseForest Road, BinfieldBracknellBerkshire RG 12 5HPPhone:01344-48 28 20Fax: 01344-48 28 10

GD Rectifiers Ltd.Victoria GardensBurgess HillWest Sussex RH159NBPhone:01444-24 34 52Fax: 01444-87-07-22

Larontrol Ltd.Unit K4 Brookside AvenueRustington Trading EstateLittlehamptonWest Sussex BN16 3LFPhone:01903-77 11 60Fax: 01903-77 20 73

Future ElectronicsPoyle Road, ColnbrookBerkshire SL3 OEZPhone: 01753-68 70 00Fax: 01753-68 91 00

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