Design of High Frequency, Low Phase Noise LC Digitally ...

electronics

Article

Design of High Frequency, Low Phase Noise LCDigitally Controlled Oscillator for 5G IntelligentTransport Systems

Vytautas Macaitis * and Romualdas Navickas

Micro and Nanoelectronics Systems Design and Research Laboratory, Vilnius Gediminas Technical University,10257 Vilnius, Lithuania; [email protected]* Correspondence: [email protected]; Tel.: +370-5-274-4768

Received: 9 November 2018; Accepted: 3 January 2019; Published: 8 January 2019

Abstract: This paper presents the design, simulation, and measurements of a low power, low phasenoise 10.25–11.78 GHz LC digitally controlled oscillator (LC DCO) with extended true single phaseclock (E-TSPC) frequency divider in 130 nm complementary metal–oxide–semiconductor (CMOS)technology for 5G intelligent transport systems. The main goal of this work was to design the LCDCO using a mature and low-cost 130 nm CMOS technology. The designed integrated circuit (IC)consisted of two parts: the LC DCO frequency generation and division circuit and an independentfrequency divider testing circuit. The proposed LC DCO consisted of the following main blocks:the high Q-factor inductor, switched-capacitors block, cross-coupled transistors, and the currentcontrol block. Inductors with switched-capacitors block formed an LC tank. The designed E-TSPCfrequency divider consisted of eight blocks connected in a series; each block increased the divisionratio by a factor of two. The frequency of the input signal was divided in the region from two to256 times using the designed divider. The main parameters of the designed E-TSPC divider andthe LC DCO measurements were given as follows: LC DCO achieved a wide tuning range from10.25 GHz to 11.78 GHz (1.53 GHz, 15.28% bandwidth); phase noise at 1 MHz offset frequency fromLC DCO lowest carrier frequency was −113.42 dBc/Hz; phase noise at 1 MHz offset frequency fromLC DCO highest carrier frequency was −110.51 dBc/Hz; The average power consumption of thedesigned LC DCO core and E-TSPC divider were 10.02 mW and 97.52 mW, respectively; the figure ofmerit (FOM) and the extended FOMT values of the proposed LC DCO were −183.52 dBc/Hz and−187.20 dBc/Hz, respectively. These FOM and FOMT results were achieved due to very low phasenoise (−113.52 dBc/Hz) and a wide frequency tuning range (15.28%). The total layout area includingthe pads was 1.5 mm × 1.5 mm, with the largest part of the layout occupied by the proposed LCDCO (193 µm × 311 µm). The largest part of the LC DCO was occupied by the inductor 184 µm ×184 µm. The manufactured chip was packed into a quad flat no-leads (QFN) 20 pads package.

Keywords: digitally controlled oscillator; voltage controlled oscillator; divider; phase locked loop;DPLL; 5G; intelligent transportation systems

1. Introduction

The automotive industry is evolving through interconnected and autonomous vehicles thatbring reduced traffic congestion, reduced environmental impact, and reduced capital costs. Vehicleto everything (V2X) technology is the key to intelligent transportation systems (ITS) based on 5Gwireless network evolution. The ITS combines many different types of information and communicationtechnologies to create intelligent wireless networks. Such a smart ITS network is needed for reasonssuch as improving traffic safety (real-time weather monitoring, warning sign notifications, speed

Electronics 2019, 8, 72; doi:10.3390/electronics8010072 www.mdpi.com/journal/electronics

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limitation, emergency alerts), reducing infrastructure damage (overloaded machines can be identifiedand eliminated from traffic participation, thus avoiding damage to the road), and traffic control (trafficmanagement can be adaptive depending on traffic flow). These benefits of an ITS can make traffic safer,faster, and more convenient. To create an ITS network, transmitters are required that generate signalsof a certain frequency using a frequency synthesizer based on the LC digitally controlled oscillator(LC DCO).

Most LC DCOs around 10 GHz are designed using 65 nm and newer technology, resulting in asignificant increase in the cost of the final product. The used area of silicon is not reduced by muchsince most of the design is occupied by an inductor. The basic idea behind this article was to design astandard N-channel metal oxide semiconductor (NMOS) architecture, low power, low occupancy, lowphase noise, wide tuning range—and most importantly—cheap DCO using low-cost 130 nm technology.The application of this low-cost DCO in the constantly evolving intelligent transport systems wasthe main goal. The main difficulty was getting the required parameters and performance using amature low-cost 130 nm IBM radio frequency (RF) integrated circuit (IC) technology. High-quality RFtransistors and high-quality factor Q copper inductors were used in this technology with correlatingresults of the simulations and measurements. In this technology, copper and aluminum wiring witha thick top metal was used. The 130 nm technology is a semiconductor technology for low-cost andhigh-performance wireless applications. Unfortunately, specific and more detailed information aboutthis technology is under a non-disclosure agreement.

The aim of this work was to create an LC DCO whose output signal’s frequency range is suitablefor USA and Japan 5G ITS bands. The LC DCO output signal frequency should correspond to the 2.5LTE B24 and 600 MHz USA 5G band, as well as the —4.4–4.9 GHz Japan 5G band [1]. DCO may beused in other communication systems.

The LC DCO alone could not generate a stable frequency, thus additional blocks were used thatcollectively formed a digital or analog phase locked loop (PLL), which was applied in multibandmultistandard radio frequency (RF) transceivers.

As one of the most important blocks of any wireless multiband, the multistandard transceiver isthe PLL. The main PLL applications are signal synthesis, synchronization, modulation, demodulation,as well as signal tracking. There are two main types of PLLs: conventional PLL (CPLL) and all digitalPLLs (ADPLL). The CPLLs main advantages are frequency synthesis with good performance and ahigh frequency range, but the disadvantages are the high power consumption and large chip area.ADPLLs are characterized by low power consumption, small area, and scalability across differenttechnology nodes. The blocks of these PLLs are presented in the following paragraphs.

The basic PLL block diagram is shown in Figure 1a. In a typical PLL, a voltage controlled oscillator(VCO) generates unstable output signal frequency FOUT. The transceiver cannot function properlywhen the frequency of the generated signal is unstable, thus an additional reference signal (FREF)is used to stabilize the frequency. The FOUT adjustment is as follows: the feedback divider (DIV)generates a comparison frequency using the VCO output signal FOUT divided by coefficient N; phasefrequency detector (PFD) compares the phase/frequency of two signals with frequencies FREF andFOUT/N at its input. The purpose of the PFD is to detect the phase difference between FREF and FOUT/N.It is considered that the PLL is locked when the FREF signal is equal to the signal FOUT/N. The mainpurpose of a charge pump (CP) is to convert the logic states of the phase frequency detector into analogsignals suitable to control the VCO. The output of the charge pump is connected to an analog low passfilter (ALPF) that integrates the charge pump output current to an equivalent VCO control voltage [2].

The basic ADPLL synthesizer block diagram is shown in Figure 1b. In the digital controlledoscillator, DCO replaces the VCO. The time-to-digital converter (TDC) with the phase detector (PD)replaces the phase frequency detector and the charge-pump of the ADPLL. The digital low pass filter(DLPF) replaces the analog low pass filter of the PLL. The main advantages of the ADPLL are less chiparea, elimination of additional external control voltage (mostly, the ALPF is realized on the printedcircuit board (PCB) side, the DLPF is integrated inside the chip), the chip cost is lower, and, with

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digitalized block controls, all system programming is simplified [3]. The main advantages of LC DCOversus LC VCO are the lower occupied area, lower phase noise, lower power consumption, and theability to digitize further deployable systems. The following paragraphs outline the main architecturesof LC DCO.Electronics 2019, 8 FOR PEER REVIEW 3

(a) (b)

Figure 1. Basic block diagrams of: (a) conventional phase locked loop (CPLL) synthesizer; (b) all digital phase locked loop (ADPLL) synthesizer.

The basic ADPLL synthesizer block diagram is shown in Figure 1b. In the digital controlled oscillator, DCO replaces the VCO. The time-to-digital converter (TDC) with the phase detector (PD) replaces the phase frequency detector and the charge-pump of the ADPLL. The digital low pass filter (DLPF) replaces the analog low pass filter of the PLL. The main advantages of the ADPLL are less chip area, elimination of additional external control voltage (mostly, the ALPF is realized on the printed circuit board (PCB) side, the DLPF is integrated inside the chip), the chip cost is lower, and, with digitalized block controls, all system programming is simplified [3]. The main advantages of LC DCO versus LC VCO are the lower occupied area, lower phase noise, lower power consumption, and the ability to digitize further deployable systems. The following paragraphs outline the main architectures of LC DCO.

The basic architectures of N-channel metal oxide semiconductor (NMOS) LC DCO cores are presented in Figure 2a,b. The current source connected to the ground architecture (Figure 2a) has the lowest sensitivity to noise on the ground line but the highest sensitivity to power supply source noise. This architecture is characterized by lower flicker noise in the output compared to architecture, as shown in Figure 2b. This is achieved due to a more correct output signal waveform. Reduced flicker noise causes lower distortion of the output signal. Using this architecture’s output signal swing can be doubled compared to the used supply voltage and can be used for phase noise reduction [4].

(a) (b)

Figure 1. Basic block diagrams of: (a) conventional phase locked loop (CPLL) synthesizer; (b) all digitalphase locked loop (ADPLL) synthesizer.

The basic architectures of N-channel metal oxide semiconductor (NMOS) LC DCO cores arepresented in Figure 2a,b. The current source connected to the ground architecture (Figure 2a) hasthe lowest sensitivity to noise on the ground line but the highest sensitivity to power supply sourcenoise. This architecture is characterized by lower flicker noise in the output compared to architecture,as shown in Figure 2b. This is achieved due to a more correct output signal waveform. Reduced flickernoise causes lower distortion of the output signal. Using this architecture’s output signal swing can bedoubled compared to the used supply voltage and can be used for phase noise reduction [4].

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(a) (b)

Figure 1. Basic block diagrams of: (a) conventional phase locked loop (CPLL) synthesizer; (b) all digital phase locked loop (ADPLL) synthesizer.

The basic ADPLL synthesizer block diagram is shown in Figure 1b. In the digital controlled oscillator, DCO replaces the VCO. The time-to-digital converter (TDC) with the phase detector (PD) replaces the phase frequency detector and the charge-pump of the ADPLL. The digital low pass filter (DLPF) replaces the analog low pass filter of the PLL. The main advantages of the ADPLL are less chip area, elimination of additional external control voltage (mostly, the ALPF is realized on the printed circuit board (PCB) side, the DLPF is integrated inside the chip), the chip cost is lower, and, with digitalized block controls, all system programming is simplified [3]. The main advantages of LC DCO versus LC VCO are the lower occupied area, lower phase noise, lower power consumption, and the ability to digitize further deployable systems. The following paragraphs outline the main architectures of LC DCO.

The basic architectures of N-channel metal oxide semiconductor (NMOS) LC DCO cores are presented in Figure 2a,b. The current source connected to the ground architecture (Figure 2a) has the lowest sensitivity to noise on the ground line but the highest sensitivity to power supply source noise. This architecture is characterized by lower flicker noise in the output compared to architecture, as shown in Figure 2b. This is achieved due to a more correct output signal waveform. Reduced flicker noise causes lower distortion of the output signal. Using this architecture’s output signal swing can be doubled compared to the used supply voltage and can be used for phase noise reduction [4].

(a) (b)

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(c) (d)

Figure 2. Simplified architecture of N-channel metal oxide semiconductor (NMOS) LC digitally controlled oscillator (LC DCO) cores: (a) current source (CS) connected to ground (VSS); (b) CS connected to supply (VDD). Simplified architecture of P-channel metal oxide semiconductor (PMOS) LC DCO cores: (c) CS connected to VSS; (d) CS connected to VDD.

The basic architectures of P-channel metal oxide semiconductor (PMOS) LC DCO cores are presented in Figure 2b,c. In order to obtain the same operating frequency, a PMOS-based LC DCO core would require a transistor that is twice as large as the one in an NMOS-based LC DCO core. This increases both the occupied chip area and the power consumption but improves phase noise due to the lower PMOS transistor flicker noise in comparison to that of NMOS devices.

Hybrid NMOS and PMOS transistor architectures are also possible. Such architecture consists of a PMOS transistor pair connected to the power supply while an NMOS transistors pair is connected to the ground. This results in two pairs of transistors, allowing half of the energy to be consumed by the same negative resistance. This architecture receives output swing close to the supply voltage, thus the generated signal is limited by the voltage supply, which depends on the technology of the integrated circuits. With a low output signal swing, the phase noise gets worse and further signal processing becomes more complex.

We decided to use the Figure 2a LC DCO design architecture with an NMOS LC DCO core when the current source (CS) connected to the ground in order to fulfill the requirements of reducing the occupied space but maintaining a double output signal voltage swing when compared to the supply voltage (the signal generated by the LC DCO is processed by other blocks). The LC DCO’s operating frequency is high, around 10 GHz, as NMOS transistors are faster than the PMOS, thus a higher frequency can be achieved when using NMOS architecture.

A previous work titled “CMOS Technology based LC VCO Review” contains a review and a comparison of 44 different LC oscillators published in papers [5]. In order to quantify the comparison of the latter LC oscillators, an FOMT was used. According to the acquired LC oscillators FOMT results, the best overall performance was achieved when the circuit was implemented in the 130 nm CMOS process, and the second best was achieved when using the 65 nm CMOS process. Therefore, in order to design an LC DCO with low phase noise, low power consumption, and wide tuning range while also reducing the overall price, the 130 nm CMOS process was chosen.

2. Design of the LC DCO and Frequency Divider

The simplified block diagram of the proposed LC DCO and frequency divider is presented in Figure 3. The block diagram consisted of the following components: LC tank based DCO, frequency divider, and output buffer. The designed IC consisted of two parts: the LC DCO frequency generation and division circuit and an independent frequency divider testing circuit. Due to the limited number of IC package pads, only two divider outputs (division by 64 and 256) were connected to be measured externally. In order to check the frequency divider separately, it was included as an independent block but with different division ratios (division by 16 and 32) [6].

Figure 2. Simplified architecture of N-channel metal oxide semiconductor (NMOS) LC digitallycontrolled oscillator (LC DCO) cores: (a) current source (CS) connected to ground (VSS); (b) CSconnected to supply (VDD). Simplified architecture of P-channel metal oxide semiconductor (PMOS)LC DCO cores: (c) CS connected to VSS; (d) CS connected to VDD.

The basic architectures of P-channel metal oxide semiconductor (PMOS) LC DCO cores arepresented in Figure 2b,c. In order to obtain the same operating frequency, a PMOS-based LC DCOcore would require a transistor that is twice as large as the one in an NMOS-based LC DCO core. Thisincreases both the occupied chip area and the power consumption but improves phase noise due tothe lower PMOS transistor flicker noise in comparison to that of NMOS devices.

Hybrid NMOS and PMOS transistor architectures are also possible. Such architecture consists ofa PMOS transistor pair connected to the power supply while an NMOS transistors pair is connected to

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the ground. This results in two pairs of transistors, allowing half of the energy to be consumed by thesame negative resistance. This architecture receives output swing close to the supply voltage, thus thegenerated signal is limited by the voltage supply, which depends on the technology of the integratedcircuits. With a low output signal swing, the phase noise gets worse and further signal processingbecomes more complex.

We decided to use the Figure 2a LC DCO design architecture with an NMOS LC DCO core whenthe current source (CS) connected to the ground in order to fulfill the requirements of reducing theoccupied space but maintaining a double output signal voltage swing when compared to the supplyvoltage (the signal generated by the LC DCO is processed by other blocks). The LC DCO’s operatingfrequency is high, around 10 GHz, as NMOS transistors are faster than the PMOS, thus a higherfrequency can be achieved when using NMOS architecture.

A previous work titled “CMOS Technology based LC VCO Review” contains a review and acomparison of 44 different LC oscillators published in papers [5]. In order to quantify the comparisonof the latter LC oscillators, an FOMT was used. According to the acquired LC oscillators FOMT results,the best overall performance was achieved when the circuit was implemented in the 130 nm CMOSprocess, and the second best was achieved when using the 65 nm CMOS process. Therefore, in orderto design an LC DCO with low phase noise, low power consumption, and wide tuning range whilealso reducing the overall price, the 130 nm CMOS process was chosen.

2. Design of the LC DCO and Frequency Divider

The simplified block diagram of the proposed LC DCO and frequency divider is presented inFigure 3. The block diagram consisted of the following components: LC tank based DCO, frequencydivider, and output buffer. The designed IC consisted of two parts: the LC DCO frequency generationand division circuit and an independent frequency divider testing circuit. Due to the limited numberof IC package pads, only two divider outputs (division by 64 and 256) were connected to be measuredexternally. In order to check the frequency divider separately, it was included as an independent blockbut with different division ratios (division by 16 and 32) [6].

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Figure 3. Simplified block diagram of the proposed LC DCO and frequency divider. DCO—digitally controlled oscillator, ÷N—frequency divider, BUF—output buffer, F—frequency divider input signal, F/N—frequency divider signal frequency divided by coefficient N.

2.1. LC Core Based DCO

The schematic of the proposed LC DCO is shown in Figure 4. The proposed DCO consisted of these main blocks: the highest available Q-factor inductor (L)—which was dependent on IC technology, a switched-capacitors block, cross-coupled transistors (M1, M2), and the current control block. The inductor with a switched-capacitors block formed an LC tank. Under real conditions, the generated signal in this circuit would be suppressed. To avoid this effect, the NMOS M1 and M2 transistors were connected to the circuit. They created a negative resistance to eliminate LC tank losses so that the circuit could generate constant fluctuations.

Figure 4. Schematic of the proposed LC DCO.

The size of the cross-coupled transistors M1 and M2 were chosen so that the output signal’s swing would be not less than the supply voltage. With such a level of output signal, successful further signal processing was possible. The sizes of the cross-coupled transistors are presented in the table in Figure 4.

2.1.1. Switched-Capacitor Block

Figure 3. Simplified block diagram of the proposed LC DCO and frequency divider. DCO—digitallycontrolled oscillator, ÷N—frequency divider, BUF—output buffer, F—frequency divider input signal,F/N—frequency divider signal frequency divided by coefficient N.

2.1. LC Core Based DCO

The schematic of the proposed LC DCO is shown in Figure 4. The proposed DCO consisted ofthese main blocks: the highest available Q-factor inductor (L)—which was dependent on IC technology,a switched-capacitors block, cross-coupled transistors (M1, M2), and the current control block. Theinductor with a switched-capacitors block formed an LC tank. Under real conditions, the generatedsignal in this circuit would be suppressed. To avoid this effect, the NMOS M1 and M2 transistors were

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connected to the circuit. They created a negative resistance to eliminate LC tank losses so that thecircuit could generate constant fluctuations.

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Figure 3. Simplified block diagram of the proposed LC DCO and frequency divider. DCO—digitally controlled oscillator, ÷N—frequency divider, BUF—output buffer, F—frequency divider input signal, F/N—frequency divider signal frequency divided by coefficient N.

2.1. LC Core Based DCO

The schematic of the proposed LC DCO is shown in Figure 4. The proposed DCO consisted of these main blocks: the highest available Q-factor inductor (L)—which was dependent on IC technology, a switched-capacitors block, cross-coupled transistors (M1, M2), and the current control block. The inductor with a switched-capacitors block formed an LC tank. Under real conditions, the generated signal in this circuit would be suppressed. To avoid this effect, the NMOS M1 and M2 transistors were connected to the circuit. They created a negative resistance to eliminate LC tank losses so that the circuit could generate constant fluctuations.

Figure 4. Schematic of the proposed LC DCO.

The size of the cross-coupled transistors M1 and M2 were chosen so that the output signal’s swing would be not less than the supply voltage. With such a level of output signal, successful further signal processing was possible. The sizes of the cross-coupled transistors are presented in the table in Figure 4.

2.1.1. Switched-Capacitor Block

Figure 4. Schematic of the proposed LC DCO.

The size of the cross-coupled transistors M1 and M2 were chosen so that the output signal’sswing would be not less than the supply voltage. With such a level of output signal, successful furthersignal processing was possible. The sizes of the cross-coupled transistors are presented in the tablein Figure 4.

2.1.1. Switched-Capacitor Block

Frequency tuning was performed by changing the effective capacitance of the switched-capacitorblock. The frequency was gradually changed by activating or deactivating arrays of certain sizecapacities. In this design, a 6-bit switched-capacitor array was used where each segment was controlledindependently and could be toggled to reach the required capacity. All En bit (zero to five) switches inthe proposed LC DCO were implemented as NMOS transistors.

The DCO tuning capacitance not only consisted of a switched-capacitors block to change theoscillator frequency, but it also included the parasitic capacitances. The main parasitic capacitancesand blocks that affected the DCO frequency were the capacitances of the inductor, the active elements,and any load connected to the DCO output. Therefore, the computer simulations were carried out tocalculate the exact sizes of the transistors of the switched-capacitors block. The sizes of the transistorsused in this block are presented in Figure 4.

The above mentioned NMOS transistors drains and sources were shorted, which acted as one pinof the capacitor as the gate acted as another pin of the capacitor. Controls of the switched-capacitorblock were routed to the chip output, thus the number of control bits was limited by the numberof chip pads. There were only six pads left for this purpose since the other pads were used for theconnection of input and output signals to supply voltage and ground. This resulted in 64 (26) differentDCO control values. To control the frequency more precisely, an additional switched-capacitor blockwith smaller capacitance transistors could be used.

2.1.2. Current Control Block

The current control block was of high importance in the proposed DCO and was used for LC DCOcore current regulation. The swing of the DCO output signal varied, so this block was intended tooptimize swing within the normal range. This block could adjust LC DCO power consumption to the

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optimum and had a current mirror as the base element [7]. The sizes of the current control transistorsM3 and M4 were chosen in such a way that the output current was double that of the control current.The sizes of the current control transistors are presented in the table in Figure 4.

2.1.3. Frequency Divider

A large part of the overall PLL synthesizer power consumption was drained by the frequencydivider. True single phase clock (TSPC) dividers used lower power compared to the current mode logic(CML) frequency divider, but the main disadvantage of the TSPC was limited operating frequency.However, with the advancement of CMOS technology, the TSPC working frequency increasedsignificantly. Extended TSPC (E-TSPC) designs were superior to the TSPC structure because E-TSPCflip-flops were utilized, which contained six transistors and were suitable for high frequency operationwith low power consumption [8].

The circuit of the divide by two (N/2) stage is shown in Figure 5a. The circuit operated in twomodes: pre-charge and evaluation. In the pre-charge mode—and before the operation of the fallingedge of the clock (CLK pin)—the clock was in high state and the output (OUT pin) was in low state.The node N2 was pre-charged in low state, and when the ratio of M1 was greater than M2, N1 hada high value. M2 and M4 were in the cut area in the falling edge of clock. M1 was still connected,thus the output OUT charged more rapidly than node N2, and M2 remained in the cut area. In thepre-charge mode, node N2 had the low state and node OUT had the high state. In the evaluation mode,M2 and M3 were connected on the rising edge of the clock. Both M3 and M4 were connected andoperated in the triode region, and the node N1 discharged rapidly and subsequently turned M3 on [9].The proposed divider consisted of eight series connected blocks (each block increased the dividingfactor by two). The proposed divider allowed division of the frequency of the input signal from two to256 times. (Figure 5b). The ETSPC divider consisted of three transistors branches. The output level ofthe branch was determined by the ratio of PMOS and NMOS transistor sizes. This meant that staticpower dissipation existed in the ETSPC flip-flops. Transistor sizes were selected so that the dividerused low power without affecting the operation of itself. The sizes of the transistors used in this blockare presented in the table in Figure 5a [10].Electronics 2019, 8 FOR PEER REVIEW 7

(a) (b)

Figure 5. Extended true single phase clock (E-TSPC) frequency divider: (a) divide by two stage schematic; (b) block diagram.

3. Simulation and Measurements

The proposed LC-DCO and frequency divider were designed in 130 nm RF CMOS technology in the Cadence IC software package. The design and simulation process was as follows. Firstly, a basic electrical schematic was designed and its simulations were carried out. The schematic was adjusted if the results obtained after the simulations did not meet the required conditions. Upon obtaining the required results, layout design was performed. Parasitics were extracted from the designed layout, then simulation of the layout was performed. In the absence of the required results, the electric schematic and layout were once again corrected, the layout was extracted, and simulations were performed. After obtaining the required simulation results, the main blocks were connected, which were also connected to the pads. After these steps, the whole system was verified. After making the chip, the basic parameters were measured and compared with the parameters obtained during the simulation. Simulations were performed in nominal conditions using 1.8 V supply voltage and 40 °C temperature. The main parameters of the simulations and measurements are presented in Table 1.

To evaluate the overall performance of the frequency oscillators, an FOM (1) and FOMT (2), including the tuning range, was used:

0( ) 20log 10log10 1F PFOM PN ff mW

= Δ − + Δ ⋅ , (1)

0( ) 20log 10log10 1T

F F PFOM PN ff mW

⋅Δ = Δ − + Δ ⋅ , (2)

where PN(Δf) is the phase noise at an offset frequency Δf, F0 is the carrier frequency, P is the power consumption in mW, and ΔF is a percentage of the frequency tuning range [5]. The proposed LC DCO and E-TSPC divider schematics, post-layout, and chip main parameters are presented in Table 1.

Table 1. Main parameters of the proposed LC DCO and E-TSPC divider.

Parameter Schematic Layout Measurements VDD, V 1.8 1.8 1.8

Technology, nm 130 FMIN, GHz 10.7 9.54 9.25 FMAX, GHz 11.6 10.56 10.78

F0, GHz 11.15 10.05 10.01

Figure 5. Extended true single phase clock (E-TSPC) frequency divider: (a) divide by two stageschematic; (b) block diagram.

3. Simulation and Measurements

The proposed LC-DCO and frequency divider were designed in 130 nm RF CMOS technology inthe Cadence IC software package. The design and simulation process was as follows. Firstly, a basicelectrical schematic was designed and its simulations were carried out. The schematic was adjusted if

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the results obtained after the simulations did not meet the required conditions. Upon obtaining therequired results, layout design was performed. Parasitics were extracted from the designed layout,then simulation of the layout was performed. In the absence of the required results, the electricschematic and layout were once again corrected, the layout was extracted, and simulations wereperformed. After obtaining the required simulation results, the main blocks were connected, whichwere also connected to the pads. After these steps, the whole system was verified. After making thechip, the basic parameters were measured and compared with the parameters obtained during thesimulation. Simulations were performed in nominal conditions using 1.8 V supply voltage and 40 Ctemperature. The main parameters of the simulations and measurements are presented in Table 1.

Table 1. Main parameters of the proposed LC DCO and E-TSPC divider.

Parameter Schematic Layout Measurements

VDD, V 1.8 1.8 1.8Technology, nm 130

FMIN, GHz 10.7 9.54 9.25FMAX, GHz 11.6 10.56 10.78

F0, GHz 11.15 10.05 10.01∆F, GHz; % 0.9; 8.7 1.02; 10.15 1.53; 15.28

PN@1 MHz, dBc/Hz −116.21 −114.79 −113.42PDCO, mW 8.73 9.15 10.02PDIV, mW 90.5 93.21 97.52

FOM, dBc/Hz −187.74 −185.22 −183.52FOMT, dBc/Hz −185.88 −185.35 −187.20

VDD—supply voltage; technology—integrated circuit (IC) technology; FMIN—lowest LC DCO output signalfrequency; FMAX—highest LC DCO output signal frequency; F0—central LC DCO output signal frequency; ∆F—LCDCO frequency tuning range; PN@1MHz—LC DCO phase noise at 1 MHz offset frequency at lowest LC DCOcarrier frequency; PDCO—LC DCO power consumption; PDIV—E-TSPC divider power consumption; FOM—figureof merit; FOMT—figure of merit including the tuning range.

To evaluate the overall performance of the frequency oscillators, an FOM (1) and FOMT (2),including the tuning range, was used:

FOM = PN(∆ f )− 20 log(

F0

∆ f · 10

)+ 10 log

(P

1mW

), (1)

FOMT = PN(∆ f )− 20 log(

F0 · ∆F∆ f · 10

)+ 10 log

(P

1mW

), (2)

where PN(∆f ) is the phase noise at an offset frequency ∆f, F0 is the carrier frequency, P is the powerconsumption in mW, and ∆F is a percentage of the frequency tuning range [5]. The proposed LC DCOand E-TSPC divider schematics, post-layout, and chip main parameters are presented in Table 1.

Figure 6 describes the tuning range of the proposed LC DCO. The LC DCO achieved a widetuning range from 10.07 GHz to 11.16 GHz during schematic simulations. The tuning range reducedto 9.54–10.56 GHz during the post-layout simulation. The measured tuning range was 9.25–10.78 GHz,which was better than both the schematic and the post-layout simulation, and reached a 1.53 GHzbandwidth, which was 0.51 GHz wider than the results obtained in the post-layout simulation.

The frequency spectrum measurement results of the proposed LC DCO divided by a 64 outputwhen operating at the lowest and highest operating frequencies are shown in Figure 7. From the latterpicture (a), it is seen that the lowest output signal frequency (when dividing by 64) was 144.6 MHz.This resulted in the proposed LC DCO generating a 144.6 MHz × 64 = 9.25 GHz frequency signal. Thehighest output signal frequency (when dividing from 64) was 168.4 MHz, which is shown in Figure 7b.The LC DCO generated a 168.4 MHz × 64 = 10.78 GHz frequency signal.

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∆F, GHz; % 0.9; 8.7 1.02; 10.15 1.53; 15.28 PN@1 MHz, dBc/Hz −116.21 −114.79 −113.42

PDCO, mW 8.73 9.15 10.02 PDIV, mW 90.5 93.21 97.52

FOM, dBc/Hz −187.74 −185.22 −183.52 FOMT, dBc/Hz −185.88 −185.35 −187.20

VDD—supply voltage; technology—integrated circuit (IC) technology; FMIN—lowest LC DCO output signal frequency; FMAX—highest LC DCO output signal frequency; F0—central LC DCO output signal frequency; ∆F—LC DCO frequency tuning range; PN@1MHz—LC DCO phase noise at 1 MHz offset frequency at lowest LC DCO carrier frequency; PDCO—LC DCO power consumption; PDIV—E-TSPC divider power consumption; FOM—figure of merit; FOMT—figure of merit including the tuning range.

Figure 6 describes the tuning range of the proposed LC DCO. The LC DCO achieved a wide tuning range from 10.07 GHz to 11.16 GHz during schematic simulations. The tuning range reduced to 9.54–10.56 GHz during the post-layout simulation. The measured tuning range was 9.25–10.78 GHz, which was better than both the schematic and the post-layout simulation, and reached a 1.53 GHz bandwidth, which was 0.51 GHz wider than the results obtained in the post-layout simulation.

Figure 6. Tuning range of the proposed LC DCO.

The frequency spectrum measurement results of the proposed LC DCO divided by a 64 output when operating at the lowest and highest operating frequencies are shown in Figure 7. From the latter picture (a), it is seen that the lowest output signal frequency (when dividing by 64) was 144.6 MHz. This resulted in the proposed LC DCO generating a 144.6 MHz × 64 = 9.25 GHz frequency signal. The highest output signal frequency (when dividing from 64) was 168.4 MHz, which is shown in Figure 7b. The LC DCO generated a 168.4 MHz × 64 = 10.78 GHz frequency signal.

(a) (b)

Figure 6. Tuning range of the proposed LC DCO.

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∆F, GHz; % 0.9; 8.7 1.02; 10.15 1.53; 15.28 PN@1 MHz, dBc/Hz −116.21 −114.79 −113.42

PDCO, mW 8.73 9.15 10.02 PDIV, mW 90.5 93.21 97.52

FOM, dBc/Hz −187.74 −185.22 −183.52 FOMT, dBc/Hz −185.88 −185.35 −187.20

VDD—supply voltage; technology—integrated circuit (IC) technology; FMIN—lowest LC DCO output signal frequency; FMAX—highest LC DCO output signal frequency; F0—central LC DCO output signal frequency; ∆F—LC DCO frequency tuning range; PN@1MHz—LC DCO phase noise at 1 MHz offset frequency at lowest LC DCO carrier frequency; PDCO—LC DCO power consumption; PDIV—E-TSPC divider power consumption; FOM—figure of merit; FOMT—figure of merit including the tuning range.

Figure 6 describes the tuning range of the proposed LC DCO. The LC DCO achieved a wide tuning range from 10.07 GHz to 11.16 GHz during schematic simulations. The tuning range reduced to 9.54–10.56 GHz during the post-layout simulation. The measured tuning range was 9.25–10.78 GHz, which was better than both the schematic and the post-layout simulation, and reached a 1.53 GHz bandwidth, which was 0.51 GHz wider than the results obtained in the post-layout simulation.

Figure 6. Tuning range of the proposed LC DCO.

The frequency spectrum measurement results of the proposed LC DCO divided by a 64 output when operating at the lowest and highest operating frequencies are shown in Figure 7. From the latter picture (a), it is seen that the lowest output signal frequency (when dividing by 64) was 144.6 MHz. This resulted in the proposed LC DCO generating a 144.6 MHz × 64 = 9.25 GHz frequency signal. The highest output signal frequency (when dividing from 64) was 168.4 MHz, which is shown in Figure 7b. The LC DCO generated a 168.4 MHz × 64 = 10.78 GHz frequency signal.

(a) (b)

Figure 7. Frequency spectrum measurement results of the proposed LC DCO divide-by-64 output(a) when operating at the lowest operating frequency, and (b) when operating at highest operatingfrequency. Measurements were performed using Keysight Technologies N9010A EXA signal analyzer.

The phase noise at the designed LC DCO output (divided by 64 when operating at the lowestfrequency) is shown in Figure 8. From this, it is visible that the phase noise at 1 MHz offset frequencyfrom the DCO lowest (En bit = 0) carrier frequency was −130.42 dBc/Hz output of the frequencydivider when operating at the lowest frequencies.

Electronics 2019, 8 FOR PEER REVIEW 9

Figure 7. Frequency spectrum measurement results of the proposed LC DCO divide-by-64 output (a) when operating at the lowest operating frequency, and (b) when operating at highest operating frequency. Measurements were performed using Keysight Technologies N9010A EXA signal analyzer.

The phase noise at the designed LC DCO output (divided by 64 when operating at the lowest frequency) is shown in Figure 8. From this, it is visible that the phase noise at 1 MHz offset frequency from the DCO lowest (En bit = 0) carrier frequency was −130.42 dBc/Hz output of the frequency divider when operating at the lowest frequencies.

Figure 8. Phase noise measurement results of the proposed DCO divide-by-64 output when operating at the lowest operating frequency (144.6 MHz). Measurements were performed using Keysight Technologies N9010A EXA signal analyzer.

It was noted that the phase noise at 1 Mhz offset from frequency divider’s output frequency when the DCO was operating at the lowest frequency (En bit = 0, FMIN = 144.6 MHz) was −130.42 dBc/Hz at the divider output. The phase noise at 1 MHz offset frequency from the DCO highest carrier frequency (En bit = 63, FMAX = 168.4 GHz) was −128.13 dBc/Hz at the divider output. These parameters were obtained by measuring the phase noise at the divide-by-64 output. This theory revealed that at each divider cascade, phase noise improved by 3 dBc/Hz [11]. Therefore, it was possible to estimate the real LC DCO phase noise at the input of the first of the six divider cascades (F/64): −130.42 + (3 × 6) = −113.42 dBc/Hz at 1 MHz offset from the lowest LC DCO carrier frequency (FMIN = 9.25 GHz) and −128.51 + (3 × 6) = −110.51 dBc/Hz at 1 MHz offset from the highest LC DCO carrier frequency (FMAX = 10.78 GHz). During the post-layout simulations, the following phase noise values were obtained: −114.79 dBc/Hz at 1 MHz offset from the lowest LC DCO carrier frequency and −113.62 dBc/Hz at 1 MHz offset from the highest LC DCO carrier frequency.

Table 2 shows the proposed LC DCO performance compared with other reported LC DCOs. The table shows that all compared DCOs were designed using 65 nm technology, while this paper’s proposed LC DCO, which was designed using 130 nm technology, lead to reduced production cost of the chip. The lowest supply voltage was used in [12] and was equal to 1.1 V, which affected the lowest power consumption of the LC DCO core −3.3 mW. All of the reviewed and reported DCOs had a carrier frequency that varied around 10 GHz. The widest DCO tuning range was [13] DCO (15.12%). The lowest phase noise result was in the proposed LC DCO and reached –113.42 dBc/Hz at 1 MHz offset frequency from the lowest carrier frequency. It was possible to achieve such low phase noise due to the high quality inductor. The FOM and FOMT values of the proposed LC DCO were –183.52 dBc/Hz and –187.20 dBc/Hz, respectively. These results were achieved due to very low phase noise (–113.52 dBc/Hz) and wide frequency tuning range (15.28%).

Table 2. Performance of the proposed LC DCO compared with other LC DCOs.

Figure 8. Phase noise measurement results of the proposed DCO divide-by-64 output when operatingat the lowest operating frequency (144.6 MHz). Measurements were performed using KeysightTechnologies N9010A EXA signal analyzer.

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It was noted that the phase noise at 1 Mhz offset from frequency divider’s output frequency whenthe DCO was operating at the lowest frequency (En bit = 0, FMIN = 144.6 MHz) was −130.42 dBc/Hz atthe divider output. The phase noise at 1 MHz offset frequency from the DCO highest carrier frequency(En bit = 63, FMAX = 168.4 GHz) was −128.13 dBc/Hz at the divider output. These parameters wereobtained by measuring the phase noise at the divide-by-64 output. This theory revealed that at eachdivider cascade, phase noise improved by 3 dBc/Hz [11]. Therefore, it was possible to estimate thereal LC DCO phase noise at the input of the first of the six divider cascades (F/64): −130.42 + (3 × 6) =−113.42 dBc/Hz at 1 MHz offset from the lowest LC DCO carrier frequency (FMIN = 9.25 GHz) and−128.51 + (3 × 6) = −110.51 dBc/Hz at 1 MHz offset from the highest LC DCO carrier frequency (FMAX

= 10.78 GHz). During the post-layout simulations, the following phase noise values were obtained:−114.79 dBc/Hz at 1 MHz offset from the lowest LC DCO carrier frequency and −113.62 dBc/Hz at 1MHz offset from the highest LC DCO carrier frequency.

Table 2 shows the proposed LC DCO performance compared with other reported LC DCOs.The table shows that all compared DCOs were designed using 65 nm technology, while this paper’sproposed LC DCO, which was designed using 130 nm technology, lead to reduced production costof the chip. The lowest supply voltage was used in [12] and was equal to 1.1 V, which affected thelowest power consumption of the LC DCO core −3.3 mW. All of the reviewed and reported DCOs hada carrier frequency that varied around 10 GHz. The widest DCO tuning range was [13] DCO (15.12%).The lowest phase noise result was in the proposed LC DCO and reached –113.42 dBc/Hz at 1 MHzoffset frequency from the lowest carrier frequency. It was possible to achieve such low phase noisedue to the high quality inductor. The FOM and FOMT values of the proposed LC DCO were –183.52dBc/Hz and –187.20 dBc/Hz, respectively. These results were achieved due to very low phase noise(–113.52 dBc/Hz) and wide frequency tuning range (15.28%).

Table 2. Performance of the proposed LC DCO compared with other LC DCOs.

Parameter [12] [13] [14] [15] This Work **

VDD, V 1.1 1.2 1.2 2 1.8Technology, nm 65 65 65 65 130

FMIN, GHz 9.87 13.69 9.60 10.60 9.25FMAX, GHz 10.92 15.93 11.60 11.70 10.78

F0, GHz 10.40 14.81 10.60 11.15 10.01∆F, % 10.10 15.12 9.87 9,87 15.28

PN@1 MHz,dBc/Hz −102.00 −99.00 −101.00 −116.00 −113.42

P, mW 3.30 19.80 29.00 36.00 10.02FOM, dBc/Hz −177.15 −169.44 −166.88 –181.38 −183.52

FOMT, dBc/Hz −177.24 −173.04 −172.40 −181.26 −187.20

** Parameters of IC measurement.

4. Prototype Test-Bench

The layout of the proposed LC DCO with the E-TSPC frequency divider is shown in Figure 9. Thetotal layout area with pads was 1.5 mm × 1.5 mm. The largest part of the layout in the proposed LCDCO had an area of about 193 µm × 311 µm. The largest part of the LC DCO was the inductor L—184µm × 184 µm. The E-TSPC divider occupied a 304 µm × 44 µm area of the chip. One pad occupied160 µm × 324 µm. Other dimensions presented in Figure 9.

The LC DCO and frequency divider IC in quad flat no-leads (QFN) with a 20 pads packageis presented in Figure 10a. The manufactured chip was packed into a QFN 20 pads package. Themain benefits of this package over the traditional small outline integrated circuit (SOIC), the shrinksmall outline package (SSOP), the thin shrink small outline package (TSSOP), or the thin very smalloutline package (TVSOP) are: the packages are physically smaller and allow the chips to be used onsmall printed circuit boards (PCBs), which reduces the cost and dimensions of the gadget; the QFN

Electronics 2019, 8, 72 10 of 12

package has a smaller area of the route, which reduces the length of interconnections, resulting inreduced parasitic impedances and capacities; improved heat efficiency—in the case of high thermalconductivity, IC is better cooled; and finally, QFN packages have low inductance and capacitance andreduced electrical parasites allow for higher frequencies, which is why this package is suitable for RFsolutions [16,17].

Electronics 2019, 8 FOR PEER REVIEW 10

Parameter [12] [13] [14] [15] This Work ** VDD, V 1.1 1.2 1.2 2 1.8

Technology, nm 65 65 65 65 130 FMIN, GHz 9.87 13.69 9.60 10.60 9.25 FMAX, GHz 10.92 15.93 11.60 11.70 10.78

F0, GHz 10.40 14.81 10.60 11.15 10.01 ∆F, % 10.10 15.12 9.87 9,87 15.28

PN@1 MHz, dBc/Hz −102.00 −99.00 −101.00 −116.00 −113.42 P, mW 3.30 19.80 29.00 36.00 10.02

FOM, dBc/Hz −177.15 −169.44 −166.88 –181.38 −183.52 FOMT, dBc/Hz −177.24 −173.04 −172.40 −181.26 −187.20

** Parameters of IC measurement.

4. Prototype Test-Bench

The layout of the proposed LC DCO with the E-TSPC frequency divider is shown in Figure 9. The total layout area with pads was 1.5 mm × 1.5 mm. The largest part of the layout in the proposed LC DCO had an area of about 193 µm × 311 µm. The largest part of the LC DCO was the inductor L—184 µm × 184 µm. The E-TSPC divider occupied a 304 µm × 44 µm area of the chip. One pad occupied 160 µm × 324 µm. Other dimensions presented in Figure 9.

Figure 9. Layout of the proposed LC DCO and E-TSPC frequency divider with pads and dimensions.

The LC DCO and frequency divider IC in quad flat no-leads (QFN) with a 20 pads package is presented in Figure 10a. The manufactured chip was packed into a QFN 20 pads package. The main benefits of this package over the traditional small outline integrated circuit (SOIC), the shrink small outline package (SSOP), the thin shrink small outline package (TSSOP), or the thin very small outline package (TVSOP) are: the packages are physically smaller and allow the chips to be used on small printed circuit boards (PCBs), which reduces the cost and dimensions of the gadget; the QFN package has a smaller area of the route, which reduces the length of interconnections, resulting in reduced parasitic impedances and capacities; improved heat efficiency—in the case of high thermal conductivity, IC is better cooled; and finally, QFN packages have low inductance and capacitance and reduced electrical parasites allow for higher frequencies, which is why this package is suitable for RF solutions [16,17].

Figure 9. Layout of the proposed LC DCO and E-TSPC frequency divider with pads and dimensions.Electronics 2019, 8 FOR PEER REVIEW 11

(a) (b)

Figure 10. LC DCO and frequency divider IC: (a) in quad flat no-leads (QFN) with 20 pads package, and (b) in printed circuit test board (PCB).

The LC DCO and E-TSPC frequency divider IC in the printed circuit test board is presented in Figure 10b. The PCB test board dimensions were 73 × 50 mm. The PCB had two layers and was 1.6 mm thick. The dielectric layer was made of FR-4, a fiberglass-reinforced epoxy-laminated material used in printed circuit board manufacturing. For PCB pads, hot air solder leveling finish (HASL) was used. The power supply pin headers were located on the right side of the PCBl. The IC control switches were placed on the top of the PCB. The E-TSPC divider input (DIV IN) and outputs (DIV32 and DIV16) were oriented on the right side and the bottom. The presented LC DCO and E-TSPC divider outputs (DCO64 and DCO 256) were oriented on the bottom and the left side. The designed LC DCO and E-TSPC divider IC (IC1) was oriented on the center of the PCB, and the output buffers (IC2 and IC5) were oriented in the bottom of the IC1. SubMiniature version A (SMA) connectors were used for high frequency outputs and inputs (J3, J4, J5, J6, J7), and these connectors had a 50 Ω impedance. SMA was designed for use from 0 Hz to 18 GHz [18].

5. Conclusions

The main goal of this work was to design a low power, low phase noise, wide tuning range, low-area, and low-cost LC DCO and frequency divider IC for ITS systems working on a 5G network. The main difficulty was getting the required parameters and performance using the 130 nm IC technology, but the main advantage of the IC technology was the cheapness of the ICs produced. The proposed LC DCO was designed for USA and Japan 5G frequency bands. The proposed LC DCO output signal frequency divided by four and sixteen could be applied to the USA 5G network (2.5 LTE B24 and 600 MHz USA 5G bands), and the output signal frequency divided by two could be applied to the 4.4–4.9 GHz Japan 5G band.

The designed IC consisted of two parts: the LC DCO frequency generation and division circuit and the divider testing circuit. The LC DCO frequency generation and division circuit in two outputs generated a LC DCO signal frequency divided by 64 and 256 times. The divider testing circuit divided the frequency of the input signal by 16 and by 32. Using the proposed high frequency E-TSPC divider, the output frequency of the LC DCO could be divided from two to 256 times.

Simulations and measurements were performed in nominal conditions using 1.8 V supply voltage and 40 °C temperature. The LC DCO achieved a wide tuning range from 10.07 GHz to 11.16 GHz when schematics were simulated. Frequency varied from 9.54 GHz to 10.56 GHz when the post-layout was simulated, and frequency varied from 9.25 GHz to 10.78 GHz when the chip was measured.

Measured phase noise at 1 MHz offset frequency from the LC DCO lowest carrier frequency was −113.42 dBc/Hz, and phase noise at 1 MHz offset frequency from the LC DCO highest carrier frequency was −110.51 dBc/Hz. During the post-layout simulations, the following phase noises were

Figure 10. LC DCO and frequency divider IC: (a) in quad flat no-leads (QFN) with 20 pads package,and (b) in printed circuit test board (PCB).

The LC DCO and E-TSPC frequency divider IC in the printed circuit test board is presented inFigure 10b. The PCB test board dimensions were 73 × 50 mm. The PCB had two layers and was 1.6 mmthick. The dielectric layer was made of FR-4, a fiberglass-reinforced epoxy-laminated material used inprinted circuit board manufacturing. For PCB pads, hot air solder leveling finish (HASL) was used.The power supply pin headers were located on the right side of the PCBl. The IC control switches wereplaced on the top of the PCB. The E-TSPC divider input (DIV IN) and outputs (DIV32 and DIV16)were oriented on the right side and the bottom. The presented LC DCO and E-TSPC divider outputs(DCO64 and DCO 256) were oriented on the bottom and the left side. The designed LC DCO andE-TSPC divider IC (IC1) was oriented on the center of the PCB, and the output buffers (IC2 and IC5)were oriented in the bottom of the IC1. SubMiniature version A (SMA) connectors were used for highfrequency outputs and inputs (J3, J4, J5, J6, J7), and these connectors had a 50 Ω impedance. SMA wasdesigned for use from 0 Hz to 18 GHz [18].

5. Conclusions

The main goal of this work was to design a low power, low phase noise, wide tuning range,low-area, and low-cost LC DCO and frequency divider IC for ITS systems working on a 5G network.

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The main difficulty was getting the required parameters and performance using the 130 nm ICtechnology, but the main advantage of the IC technology was the cheapness of the ICs produced. Theproposed LC DCO was designed for USA and Japan 5G frequency bands. The proposed LC DCOoutput signal frequency divided by four and sixteen could be applied to the USA 5G network (2.5 LTEB24 and 600 MHz USA 5G bands), and the output signal frequency divided by two could be applied tothe 4.4–4.9 GHz Japan 5G band.

The designed IC consisted of two parts: the LC DCO frequency generation and division circuitand the divider testing circuit. The LC DCO frequency generation and division circuit in two outputsgenerated a LC DCO signal frequency divided by 64 and 256 times. The divider testing circuit dividedthe frequency of the input signal by 16 and by 32. Using the proposed high frequency E-TSPC divider,the output frequency of the LC DCO could be divided from two to 256 times.

Simulations and measurements were performed in nominal conditions using 1.8 V supply voltageand 40 C temperature. The LC DCO achieved a wide tuning range from 10.07 GHz to 11.16 GHzwhen schematics were simulated. Frequency varied from 9.54 GHz to 10.56 GHz when the post-layoutwas simulated, and frequency varied from 9.25 GHz to 10.78 GHz when the chip was measured.

Measured phase noise at 1 MHz offset frequency from the LC DCO lowest carrier frequencywas −113.42 dBc/Hz, and phase noise at 1 MHz offset frequency from the LC DCO highest carrierfrequency was −110.51 dBc/Hz. During the post-layout simulations, the following phase noiseswere obtained: −114.79 dBc/Hz at 1 MHz offset from the lowest LC DCO carrier frequency and−113.62 dBc/Hz at 1 MHz offset from the highest LC DCO carrier frequency.

The measured average power consumption of the proposed LC DCO IC core was 10.02 mW, whichwas higher than it was during both the schematic and the post-layout simulations, which were equalto 8.73 mW and 9.15 mW, respectively. The measured power consumption average of the proposedE-TSPC divider IC was 97.52 mW, which was also higher than it was during both the schematic andthe post-layout simulations, which were 90.05 mW and 93.21 mW, respectively. To optimize powerconsumption, it is possible to limit the current to optimum using the “current control” block.

The FOM and FOMT values of the proposed LC DCO were −183.52 dBc/Hz and −187.20 dBc/Hz,respectively. These results were achieved due to very low phase noise (−113.52 dBc/Hz) and widefrequency tuning range (15.28%).

The total layout area with pads was 1.5 mm × 1.5 mm. The largest part of the layout was occupiedby the proposed LC DCO, with an area of about 193 µm × 311 µm, while the largest part of the LCDCO was the inductor L—184 µm × 184 µm.

Author Contributions: All authors contributed to the present paper with the same effort in finding availableliterature resources, conducting both simulations and measurements, as well as writing the paper.

Funding: This research was funded by the Research Council of Lithuania grant number DOTSUT-235,No. 01.2.2-LMT-K-718-01-0054 as a part of “Design and Research of Internet of Things (IoT) Framework Model andTools for Intelligent Transport Systems” project. The APC was funded by Vilnius Gediminas technical universityFaculty of Electronics.

Acknowledgments: The authors would like to thank the project “Design and Research of Internet of Things (IoT)Framework Model and Tools for Intelligent Transport Systems” team and Vaidotas Barzdenas for organizingprototype chip manufacturing, as well as Aleksandr Vasjanov for observations while writing this paper.

Conflicts of Interest: The authors declare no conflict of interest.

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© 2019 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open accessarticle distributed under the terms and conditions of the Creative Commons Attribution(CC BY) license (http://creativecommons.org/licenses/by/4.0/).