CHAPTER 3 Wideband Code Division Multiple Access (WCDMA…shodhganga.inflibnet.ac.in/bitstream/10603/26343/12/12_chapter_03.pdf · 31 3.2 Implementation of WCDMA Fig 3.1 WCDMA system

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CHAPTER 3

Wideband Code Division Multiple Access (WCDMA)

3.1 Introduction

The new requirements of the third generation systems are listed below:

Bit rates up to 2Mbps.

Variable bit rate to offer band width on demand.

Multiplexing of services with different quality requirements on a single connection,

e.g. speech, video and packet data.

Delay requirements from delay-sensitive real time traffic to flexible best effort

packet data.

Quality requirements from 10% frame error rate to 10-6 bit error rate.

Co-existence of second and third generation systems and inter system handovers for

coverage enhancements and load balancing.

Support of asymmetric uplink and downlink traffic, e.g. web browsing causes more

loading to downlink than to uplink.

High spectrum efficiency.

Co-existence of FDD and TDD modes.

Carrier spacing 5 MHz (normal)

Chip rate 3.84 Mcps

Frame length 10 ms (38400 chips)

Number of slots / frame 15

Number of chips / slot 2560 chips (normal)

Uplink SF 4 to 256

Downlink SF 4 to 512

Channel rate 7.5 Kbps to 960 Kbps

Table 3.1 Some parameters of WCDMA physical layer

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3.2 Implementation of WCDMA

Fig 3.1 WCDMA system model

3.2.1 Channel izat ion Codes

There are certain restrictions as to which of the channelization codes can be

used for a transmission from a single source. Another physical channel may use a

certain code in WCDMA.

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Specification Channelization code Scrambling code

Usage Uplink: Separation of

physical data (DPDCH) and

control channels (DPCCH)

from same terminal

Uplink: Separation of terminal

Downlink: Separation of

downlink connections to

different users within one

cell

Downlink: Separation of sectors

(cells)

Length Uplink: 4 – 256 chips (1.0

– 66.7 µs)

Uplink (1) 10ms = 38400 chips

Uplink (2) 66.7 µs=256 chips

(advanced BS receivers)

Downlink: 512 chips Downlink: 10ms = 38400 chips

Number of codes Number of codes under one

scrambling code =

spreading factor

Uplink: several millions

Downlink : 512

Code family OVSF Long 10ms = Gold code

Short – extended S(2) code family

Spreading Yes, increases Transmission

bandwidth

No, doesn’t affect transmission

bandwidth

Table 3.2: Functionality of the channelization and scrambling codes

the tree if no other physical channel to be transmitted using the same code tree is

using a code that is on an underlying branch, i.e. using a higher spreading factor code

generated from the intended spreading code to be used. Neither can a smaller

spreading factor code on the path to the root of the tree be used. The downlink

orthogonal codes within each base station are managed by the radio network

controller (RNC) in the network.

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The functionality and characteristics of the scrambling and

channelization codes are summarized in Table 3.2 One code tree is used with

one scrambling code on top of the tree. This means that different terminals and

different base stations may operate their code trees totally independent of each

other; there is no need to coordinate the code tree resource usage between

different base stations or terminals.

3.2.2 Uplink spreading and modulation

3.2.2.1 Uplink modulation

In the uplink direction, there are basically two additional terminal-

oriented criteria that need to be taken into account in the definition of the

modulation and spreading methods. The uplink modulation should be designed so

that the terminal amplifier efficiency is maximized and the interference from the

terminal transmission is minimized.

With GSM operation, we are familiar with the occasional audible

interference with audio equipment that is not properly protected. The interference

from GSM has a frequency of 217 Hz, which is determined by the GSM frame

frequency. This interference falls into the band that can be heard by the human

ear. The continuous transmission achieved with an I-Q/code multiplexed control

channel is shown in Figure 3.4. Now, as the pilot and the power control signaling

are maintained on a separate continuous channel, no pulsed transmission

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Fig 3.2 Parallel transmissions of DPDCH and DPCCH when data ispresent/absent (DTX)

occurs. The only pulse occurs when the data channel DPDCH is switched on and

off, but such switching seldom happens. The average interference to other users and

the cellular capacity remain the same as in the time- multiplexed solution. In

addition, the link level performance is the same in both schemes if the energy

allocated to the pilot and the power control signaling is the same.

For the best possible power amplifier efficiency, the terminal transmission

should have as low a peak-to-average (PAR) ratio as possible to allow the terminal

to operate with a minimal amplifier back-off requirement, mapping directly to

the amplifier power conversion efficiency, which in turn, is directly proportional

to the terminal talk time. With I – Q / code multiplexing, also called dual –

channel QPSK modulation, the power levels of the DPDCH and DPCCH are

typically different, especially as data rates increases, and would lead in extreme

cases to BPSK – type transmission when transmitting the branches independently.

This has been avoided by using a complex-valued scrambling operation after the

spreading with channelization codes.

The signal constellation of the I-Q/code multiplexing before complex

scrambling is shown in Figure 3.5. The same constellation is obtained after

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descrambling in the receiver for the data detection G denotes the relative gain factor

between DPCCH and DPDCH branches.

.

Fig 3.3 Constellation of I-Q/code multiplexing before complex scrambling.

The transmission of two parallel channels, DPDCH and DPCCH, leads

to multicode transmission, which increases the peak-to-average power ratio

(crest factor). the transmitter power amplifier efficiency remains the same as

for normal balanced QPSK transmission in general. The complex scrambling

codes are formed in such a way that the rotations between consecutive chips

within one symbol period are limited to ±90°. The full 180° rotation can happen

only between consecutive symbols. This method further reduces the peak-to-

average ratio of the transmitted signal from the normal QPSK transmission.

The efficiency of the power amplifier remains constant, irrespective

of the power difference G between DPDCH and DPCCH. This can be

explained with the signal constellation for the I-Q/code multiplexed control

channel with complex spreading. In the middle constellation with G = 0.5, the

possible constellation points are only circles or only crosses during one symbol

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period. Their constellation is the same as for rotated QPSK. Thus, the signal

envelope variations with complex spreading are

Fig 3.4. I-Q/code multiplexing with complex scrambling

Fig 3.5 Signal constellation for I-Q/code multiplexed control channel with

complex scrambling. G

very similar to QPSK transmission for all values of G. The I – Q /code

multiplexing solution with complex scrambling results in power amplifier

output back-off requirements that remain constant as a function of the power

difference between DPDCH and DPCCH.

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3.2.2.2 Uplink Scrambling Codes

The transmissions from different sources are separated by the

scrambling codes. In the uplink direction, there are two alternatives: short and

long scrambling codes. The long codes with 25 degree generator polynomials are

truncated to the 10 ms frame length, resulting in 38 400 chips with 3.84 Mcps.

The short scrambling code length is 256 chips. The long scrambling codes are used

if the base station uses a Rake receiver (refer Appendix C). The Rake receiver is

described in next section. If advanced multiuser detectors or interference

cancellation receivers are used in the base station, short scrambling codes can be

used to make the implementation of the advanced receiver structures easier. Both

of the two scrambling code families contain millions of scrambling codes, thus,

in the uplink direction, code planning is not needed.

The short scrambling codes have been chosen from the extended S(2)

code family. The long codes are Gold codes. The complex-valued scrambling

sequence is formed in the case of short codes by combining two codes, and in the

case of long codes from a single sequence where the other sequence is the delayed

version of the first one.

The complex-valued scrambling code can be formed from two real-valued

codes C1 and C2 with the decimation principle as:

1 0 2 1(( )(2 ) )scramblingC C w jC k w (3.1)

Where 0 , 1 , 2 , . . .k

0 1

0 1{1 1} , {1 1}

w i th s e q u e n c e s w a n d w

w w (3.2)

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The decimation factor with the second code is 2. This way of creating

the scrambling codes will reduce the zero crossings in the constellation and

will further reduce the amplitude variations in the modulation process.

3.2.2.3 Spreading and Modulation on Uplink Common Channels

The Random Access Channel (RACH) contains preambles that are sent

using the same scrambling code sequence as with the uplink transmission, the

difference being that only 4096 chips from the beginning of the code period are

needed and the modulation state transitions are limited in a different way. The

spreading and scrambling process on the RACH is QPSK-value, thus only one

sequence is used to spread and scramble both the in- phase and quadrature

branches. This has been chosen to reduce the complexity of the required

matched filter in the base station receivers for RACH reception.

The RACH message part spreading and modulation, including

scrambling, is identical to that for the dedicated channel. The codes available for

RACH scrambling use are transmitted on the BCH of each cell.

For the peak-to-average reduction, an additional rotation function is used

on the RACH preamble, given as:

( )4 2( ) ( ) 0,1, 2 , ..., 409 5

j k

b k a k e k

(3.3)

where a(K) is the binary preamble and b(K) is the resulting complex-

valued preamble with limited 90° phase transition between chips. The

autocorrelation properties are not affected by this operation.

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The CPCH spreading and modulation are identical to those of the

RACH in order to maximize the commonality for both terminal and base

station implementation when supporting CPCH. RACH and CPCH

processes will be described in more detail in connection with the physical

layer procedures.

3 .3 Transmi t ter Characteri s t i c s

The pulse shaping method applied to the transmitted symbols is root-raised

cosine filtering with a roll-off factor of 0.22. The same roll-off is valid for both

the terminals and the base stations. There are a few other key RF parameters that are

introduced here and that have an essential impact on the implementation, as well

as on system behavior.

The nominal carrier spacing in WCDMA is 5 MHz but the carrier

frequency in WCDMA can be adjusted with a 200 kHz raster. The central

frequency of each WCDMA carrier is indicated with an accuracy of 200 kHz.

The target of this adjustment is to provide more flexibility for channel spacing

within the operator's band.

3.4 User Data Transmission

For user data transmission in second generation systems, such as the first

versions of GSM, typically only one service has been active at a time, either voice

or low-rate data. From the beginning, the technology base has required that the

physical layer implementation be defined to the last detail without real

flexibility. For example, puncturing patterns in GSM have been defined bit by

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bit, whereas such a definition for all possible service combinations and data rates

is simply not possible for UTRA. Instead, algorithms for generating such

patterns are defined. Signal processing technology has also evolved greatly, thus

there is no longer a need to have items like puncturing on hardware as in the

early days of GSM hardware development. For the circuit switched (CS) traffic

(e.g. speech and video), a transmission dedicated channel needs to be used, while

for packet data, there are additional choices available, RACH and CPCH for the

uplink and FACH and DSCH for the downlink.

3.5 Multiple Input and Multiple Output on WCDMA

Mobile multimedia and high speed internet access are driving the development

of third generation (3G) wireless communication systems toward significantly higher

data rates than those achieved by their second generation (2G) predecessors. To

provide this the system must support a large number of concurrent users. Using

DSCDMA, the UMTS standard allows different data rates to be allocated across the

active users by appropriate selection of spreading factors, with a maximum data rate

of 2 Mb/s. However, assuming QPSK modulation, the total uncoded data rate of the

system is limited to 8 Mb/s. A detailed analysis [12] has shown that these limits are

almost never consistently achieved in DS – CDMA systems using conventional

RAKE receivers due to multiple access interference in multipath environments.

The application of multiple antennas is expected to increase system capacity

through different mechanism. The most straight forward approach is to exploit spatial

diversity with multiple antennas at the transmitter and/or receiver. Different

techniques are used to mitigate the effects of fading and spatially non – white

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interference (in – cell and out – of – cell multiple access interference) thereby

increasing the received SINR. As such, one possible method of spatial diversity is

already defined within the UMTS standard [13]. While these methods allow the

system to Improve the transmission data rates, neither transmitter nor receiver

diversity techniques significantly increases the system capacity. In addition, some of

them are disadvantaged by their required knowledge of the channel at the transmitter.

In 1996 Foschini pointed out [14] that the use of multiple antenna at the

transmitter and at the receiver would allow for a significant increase in system

capacity through the application of spatial multiplexing. It is shown that the capacity

of such a system with M transmitter and N receiver antenna is:

2log det HM M

SNRC I HH

M

(3.4)

where H describes the M×N channel between the M transmit and N receive antennas

(the matrix elements are assumed to be complex and unit variance variables). Under

idealized conditions (e.g., orthogonal channels and M = N) this spatial multiplexing

results in an M fold capacity gain [15].

A rich scattering environment is known to favor spatial multiplexing. We refer

to this form of transmission over multiple antenna systems as BLAST (Bell Labs

Layered space time). So far the concept has been analyzed, demonstrated and verified

for narrow band system in numerous publications [16] – [21]. A study that compares

capacity and BER performance of space time coding, beam forming and BLAST in a

UMTS TDD scenario has been presented in [22] and [23]. In summary, it is

concluded that in typical indoor environments and when no reliable channel state

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information is available at the transmitter (as in the UMTS – FDD downlink) spatial

multiplexing is the method best suited for optimally exploiting the channel capacity.

Further, some theoretical and practical performance aspects will be

highlighted. Next section provides a description of the implementation of the main

components in the system. Furthermore, the testing of the system and corresponding

results are presented and prototype implementation is described. Next section

presents indoor over – the – air BER and capacity measurement results.

3.5.1 MIMO Extended UMTS – FDD downlink.

The design is based on the DS – CDMA principle and uses the UMTS - FDD

chip – rate of 3.84MHz [2, 13]. It’s essential components are the dedicated physical

channel (DPCH), a common pilot channel (CPICH) for the pilot assisted channel

estimation and the primary and secondary synchronization channels (PSCH, SSCH).

The common pilot channels, as well as the user channels, are separated by utilizing

orthogonal variable rate channelization codes, which are derived from Hadamard

matrix (OVSF – Codes). The CPICH uses a length 256 OVSF code, while the length

of the codes for the DPCH is variable to support raw data rates between 31.25kb/s and

2Mb/s in the single antenna case and 125kb/s to 8Mb/s in the proposed 4 × 4 antenna

configuration. After channelization, the channel are scrambled with the same

truncated Gold sequence. Adjacent base stations reside in the same band and are

separated through the use of different Gold sequences. The PSCH and the SSCH are

added after the scrambling and are not orthogonal to the other codes. They are both

required to achieve initial synchronization and to identify the Gold sequence used by

the targeted base station [13].

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The extensions of this basic system to support M transmit and N receive

antenna for spatial multiplexing is straight forward. After channel coding, the

incoming serial data stream of each user in this mode is first demultiplexed into M

parallel streams. Each of the stream uses the same channelization code and is

scrambled with the same Gold sequence before it is transmitted through one of the M

transmit antennas. The total transmitted power is begin conserved by sharing it

equally among the antennas and no additional user codes are required. It should also

be noted that other users, which do not use this scheme, are not affected, as the

codes of the MIMO users are still orthogonal to the other channelization codes in

the system. However, while the legacy system requires only one pilot channel for

the estimation of the 1×1 channel at the mobile, this is not sufficient for the pilot

assisted estimation of the M×N channels of the MIMO systems. Therefore, a

separate pilot channel, again using a length 256 OVSF code is assigned to each

of the transmit antennas.

The synchronization channels utilize only one of the transmit antennas,

as their replication would lead to an inherent beam forming, which is not

intended. The transmitter of the realized system with M = 4 transmit antennas.

Only two users are depicted in this diagram for readability. Many details of the

implemented system are presented in [14] – [18]. It has been shown that large

diversity gains are desired to reduce the effect of fading.

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Fig 3.6 Basic block diagram for MIMO communication

3 . 5 . 2 Signal Model and MIMO rake receiver.

It has been shown that large diversity gains are desired to reduce the effect of

fading. It is therefore concluded that multipath propagation should be exploited

whenever it exits [19]. In general, adding multipath components does not

automatically lead to better performance (see the work in [11, 20, 21] ). In addition,

for practical receiver designs multipath is not necessarily a desirable condition as it

can also lead to serious multiple access and self interference, reducing the effective

system capacity [22]. The severity of the interference depends strongly on the type of

receiver. While a MIMO equalizer (temporal and spatial) approach is a method to

avoid interference and still exploit diversity, its complexity is often prohibitive,

especially in high mobility environments where fast update rates are necessary. A

MIMO receiver offers the ability to exploit the advantages of the multipath

propagation at a reasonable implementation cost. However, this is done at the cost of

additional interference.

It has been shown that the receive diversity scheme is not suitable for the

downlink, as it is difficult and inconvenient to install multiple antennas on handsets

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[106]. The multiple antenna burden is preferably placed at the base station. This is

called transmit diversity. Transmit diversity has gained a lot of attraction and research

in last five years. In transmit diversity the same signal is transmitted from all

antennas. If same signal is transmitted from all the antennas, at the receiver the copies

of this signal add incoherently, and no diversity gain can be achieved. Thus in order to

diversity transmission to work, one must find a transmission scheme where replicas of

the signal combine coherently at the receiver. One of the simplest and most attractive

transmit diversity schemes was proposed by Alamouti [10,109].

S1 S2

G2 = (3.5)

-S*2 S*

1

Here the rows denote time instances and columns denote transmit antennas. At time t1

= 1, S1 and S2 will be transmitted from antennas 1 and 2 respectively, and at time t2, -

S2 *and S2 will be transmitted from antennas 1 and 2 respectively. One can see that

two symbols are transmitted over two time intervals. Hence the code is full rate.

Assuming the fading coefficients as h1 & h2 for the transmitting antenna 1 and 2 is

assumed to be constant over nT = 2 consecutive time slots.

h1 = h1 (T = 1) = h1 (T=2) (3.6)

h2 = h2 (T=1) = h2 (T=2) (3.7)

Hence the received signal is

Y1 = h1x1 + h2x2 + n1 (3.8)

Y2 = -h1x*2 + h2x

*1 + n2 (3.9)

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We observe from the expression that by using two transmit and one receive

antenna, the transmitted signals are effectively multiplied by h12 + h2

2 . Hence if one

of the paths is in fade, the other may still represent the signal with reliability. In fact,

the use of orthogonal STBC changes the probability distribution of the channel to

distribution with lower variance. It can also be observed that STBC in MIMO

channels can be represented as equivalent SISO channel. Tarokh et al[12] extended

Alamouti's 2-transmit diversity scheme to more than two antennas. This new

generalized space-time signaling scheme is known as Space Time Block Codes

(STBC). Space Time Block Codes derive their name from the fact that the encoding is

done in both space and time, a simple a matrix defines their encoder. A space time

block code is defined by the relationship between the k-tuple signal x and the set of

signals to be transmitted from nT antenna over p time periods. Such a relation is given

by p x nT transmission matrix.

S11 S12………….S1nT

G = S21 S22………….S2nT (3.10)

.... … …….. ….

SP1 SP2………..SpnT

Where sij are functions of k-tuple input sequence x1;x2;:::;xk and their

complex conjugates. At time slot 1, sij is transmitted from antenna j. Since k

information bits are transmitted over p time interval, the rate of the code is defined as

R= k/p. At the receiver we can use arbitrary number of receive antennas. The design

does not depend on the number of receive antennas nR. At the receiver, the nR

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receive antennas use maximum likelihood (ML) decoding. In orthogonal STBC and

ML decoding is equivalent to Maximum Ratio Combining (MRC). Assuming perfect

channel side-information (SI) the decoder at antenna j maximizes. Since the block

coding requires only linear processing at the receiver, the decoding can be done

efficiently and quickly. Space Time Block Codes can be constructed for any type of

signal constellation and provide full diversity.

Exploiting multiple antennas at the transmitter and the receiver in wireless

communication systems has recently been proven to provide substantial benefits in

both increasing system capacity and improving its immunity to deep fading in the

channel [1, 2]. To take advantage of these benefits, special space-time coding

techniques are required. A pioneering work in the area of space-time coding for

multiple-input multiple-output (MIMO) wireless channels has been done by Tarokh et

al. in [3], in which two code design criteria have been proposed for flat fading

channels with coherent receivers, and high-performance space-time trellis codes have

been designed. However, these codes suffer from rather high decoding complexity. In

the same year, Alamouti proposed his celebrated space-time block coding (STBC)

scheme for two transmit and multiple receive antennas [4]. The maximum likelihood

(ML) decoder for Alamouti’s code has very low complexity. Inspired by this work,

Tarokh et al. generalized Alamouti’s codes to multiple transmit antennas using the

theories of orthogonal and amicable designs [5]. The codes developed in [5] are

known as orthogonal STBCs (OSTBCs) and are able to provide high performance at

very low decoding complexity. Some other designs of OSTBCs have also been

recently developed in [6–8].

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OSTBC theory can be divided into two almost separate parts. The first part

deals with how OSTBCs can be designed. This aspect of the OSTBC theory is beyond

the scope of our chapter, and interested readers are referred to the relevant literature

[5–9]. The second part of this theory deals with the properties of these codes and

constitutes the focus of our chapter. Interestingly, the main properties of OSTBCs can

be derived from their simple definition proposed in [5] and are almost independent of

the way these codes have been constructed. In this chapter, we provide a background

of the theory of OSTBCs and review recent results that are based on their

constellation space invariance property and similarities between OSTBC models and

signal models used in array processing.

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3.6 Algorithm

Fig 3.7 Flow diagram

3.7 Results

The above shown algorithm is executed using MATLAB and outputs are

tabulated. The graph has been plotted between power in (dB) and frequency (Hz) on

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both the channels (AWGN & Rayleigh). Fig 3.9 represents baseband signaling of

power spectrum. Fig 3.10 represents the modulated signal at the channels plotted

against Amplitude and Time. Fig 3.11 represents the modulated signal spectrum. Fig

3.12 represents Output signal spectrum. Fig 3.13 represents Constellation output for

the signal before and after the channel. Both theoretical and simulated estimation of

AWGN and Rayleigh noise channel are shown in the Fig 3.14. The performance

analysis of 2 x 2 MIMO with different users is shown in the Fig 3.15.

Fig 3.8. Output waveform obtained from various levels of blocks

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Output signal observed along with various parameters in accordance with same input

Fig 3.9. Baseband signal power spectrum for AWGN (Left) and Rayleigh (Right) in

both the graph frequency (Hz) differs

Fig 3.10. Modulated signal at the channel for AWGN (Left) and Rayleigh (right)

Fig 3.11. Modulated signal spectrum for AWGN (Left) and Rayleigh (Right)

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Fig 3.12. Output signal spectrum for the received signal

Fig 3.13. Constellation output for the signal before and after the channel

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3.14 BER analysis of AWGN and Rayleigh channel for both theory and simulated

Fig 3.15 Performance Analysis of 2×2 Mimo Antenna With Different Users

3.8 Summary

The MATLAB® simulation test bed was written and used to extract the

main characteristics of the target application. Once broad guidelines of the design

process were available, a synthesizable VHDL description of the architecture was

written. The design of the architecture was further optimized through iterative post

synthesize simulations and redesign.