All About Op Amplifier-collection

7/23/2019 All About Op Amplifier-collection

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LM101,LM103

AN-32 FET Circuit Applications

Literature Number: SNOA620


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TLH6791

FET

CircuitApplications

AN-32

National SemiconductorApplication Note 32February 1970

FET Circuit Applications

Polycarbonate dielectric

TLH67911

Sample and Hold With Offset Adjustment

The 2N4339 JFET was selected because of its low lGSS(k100 pA) very-low lD(OFF)(k50 pA) and low pinchoff volt-

age Leakages of this level put the burden of circuit perform-ance on clean solder-resin free low leakage circuit layout

TLH67912

TLH67913

Long Time Comparator

The 2N4393 is operated as a Miller integrator The high Y fsof the 2N4393 (over 12000 mmhos 5 mA) yields a stagegain of about 60 Since the equivalent capacitance lookinginto the gate is C times gain and the gate source resistance

can be as high as 10 MX time constants as long as aminute can be achieved

JFET AC Coupled Integrator

This circuit utilizes the m-amp technique to achieve very

high voltage gain Using C1 in the circuit as a Miller integra-tor or capacitance multiplier allows this simple circuit tohandle very long time constants

C1995 National Semiconductor Corporation RRD-B30M115Printed in U S A


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TLH67914

Ultra-High ZINAC Unity Gain Amplifier

Nothing is left to chance in reducing input capacitance The

2N4416 which has low capacitance in the first place isoperated as a source follower with bootstrapped gate bias

resistor and drain Any input capacitance you get with this

circuit is due to poor layout techniques

TLH67915

TLH67916

FET Cascode Video Amplifier

The FET cascode video amplifier features very low input

loading and reduction of feedback to almost zero The2N3823 is used because of its low capacitance and highYfs Bandwidth of this amplifier is limited by R L and load

capacitance

JFET Pierce Crystal Oscillator

The JFET Pierce crystal oscillator allows a wide frequencyrange of crystals to be used without circuit modification

Since the JFET gate does not load the crystal good Q ismaintained thus insuring good frequency stability

2


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TLH67917

FETVM-FET Voltmeter

This FETVM replaces the function ot the VTVM while at the

same time ridding the instrument of the usual line cord Inaddition drift rates are far superior to vacuum tube circuits

allowing a 05 volt full scale range which is impractical with

most vacuum tubes The low-leakage low-noise 2N4340 isan ideal device for this application

TLH67918

HI-FI Tone Control Circuit (High Z Input)

The 2N3684 JFET provides the function of a high input

impedance and low noise characteristics to buffer an op

amp-operated feedback type tone control circuit

3


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Rs-SCALING RESISTORS

TLH679110

Differential Analog Switch

The FM1208 monolithic dual is used in a differential multi-

plexer application where RDS(ON) should be closely

matched Since RDS(ON) for the monolithic dual tracksat better than g1% over wide temperature ranges

(b25 to a125C) this makes it an unusual but ideal choicefor an accurate multiplexer This close tracking greatly re-

duces errors due to common mode signals

TLH679111

Magnetic-Pickup Phono Preamplifier

This preamplifier provides proper loading to a reluctancephono cartridge It provides approximately 25 dB of gain at

1 kHz (22 mV input for 100 mV output) it features S a NN

ratio of better than b70 dB (referenced to 10 mV input at1 kHz) and has a dynamic range of 84 dB (referenced to

1 kHz) The feedback provides for RIAA equalization

4


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TLH679112TLH679113

Variable Attenuator

The 2N3685 acts as a voltage variable resistor with anRDS(ON) of 800X max The 2N3685 JFET will have linear

resistance over several decades of resistance providing anexcellent electronic gain control

Negative to Positive Supply Logic Level Shifter

This simple circuit provides for level shifting from any logicfunction (such as MOS) operating from minus to ground

supply to any logic level (such as TTL) operating from a plusto ground supply The 2N3970 provides a low r ds(ON) and

fast switching times

TLH679114

Voltage Controlled Variable Gain Amplifier

The 2N4391 provides a low RDS(ON)(less than 30X) The

tee attenuator provides for optimum dynamic linear rangefor attenuation and if complete turnoff is desired attenua-

tion of greater than 100 dB can be obtained at 10 MHzproviding proper RF construction techniques are employed

AV em

2e 500 TYPICAL

m eY fs

Yos

TLH679115

Ultra-High Gain Audio Amplifier

Sometimes called the JFET m amp this circuit provides

a very low power high gain amplifying function Since m of a

JFET increases as drain current decreases the lower drain

current is the more gain you get You do sacrifice input

dynamic range with increasing gain however

5


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TLH679116

Level-Shifting-Isolation Amplifier

The 2N4341 JFET is used as a level shifter between two opamps operated at different power supply voltages The

JFET is ideally suited for this type of application becauselD e l S

Burroughs Corp

Trademark of the

TLH679117

IO eVIN

R1

VIN l 0V TLH679118

FET NixieDrivers

The 2N3684 JFETs are used as Nixie tube drivers Their Vpof 2-5 volts ideally matches DTL-TTL logic levels Diodesare used to a a50 volt prebias line to prevent breakdown of

the JFETs Since the 2N3684 is in a TO-72 (4 lead TO-18)package none of the circuit voltages appear on the canThe JFET is immune to almost all of the failure mechanisms

found in bipolar transistors used for this application

Precision Current Sink

The 2N3069 JFET and 2N2219 bipolar have inherently high

output impedance Using R1

as a current sensing resistor toprovide feedback to the LM101 op amp provides a large

amount of loop gain for negative feedback to enhance thetrue current sink nature of this circuit For small current val-ues the 10k resistor and 2N2219 may be eliminated if the

source of the JFET is connected to R1

6


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TLH679119

JFET-Bipolar Cascode Circuit

The JFET-Bipolar cascode circuit will provide full video out-put for the CRT cathode drive Gain is about 90 The cas-code configuration eliminates Miller capacitance problems

with the 2N4091 JFET thus allowing direct drive from the

video detector An m derived filter using stray capacitanceand a variable inductor prevents 45 MHz sound frequencyfrom being amplified by the video amplifier

Polycarbonate dielectric capacitor

TLH679120

Low Drift Sample and Hold

The JFETs Q1 and Q2 provide complete buffering to C1the sample and hold capacitor During sample Q1 is turnedon and provides a path r ds(ON) for charging C1 Duringhold Q1 is turned off thus leaving Q1 ID(OFF) (k50 pA)

and Q2 IGSS (k100 pA) as the only discharge paths Q2serves a buffering function so feedback to the LM101 andoutput current are supplied from its source

7


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TLH679121

Wein Bridge Sine Wave Oscillator TLH679122

JFET Sample and Hold Circuit

The major problem in producing a low distortion constantamplitude sine wave is getting the amplifier loop gain just

right By using the 2N3069 JFET as a voltage variable resis-tor in the amplifier feedback loop this can be easily

achieved The LM103 zener diode provides the voltage ref-erence for the peak sine wave amplitude this is rectifiedand fed to the gate of the 2N3069 thus varying its channel

resistance and hence loop gain

The logic voltage is applied simultaneously to the sampleand hold JFETs By matching input impedance and feed-

back resistance and capacitance errors due to rds(ON) ofthe JFETs is minimized The inherent matched r ds(ON) and

matched leakage currents of the FM1109 monolithic dualgreatly improve circuit performance

TLH679123

VOUT tR2

R1VIN

TLH679124

High Impedance Low Capacitance Wideband Buffer

The 2N4416 features low input capacitance which makes

this compound-series feedback buffer a wide-band unitygain amplifier

High Impedance Low Capacitance Amplifier

This compound series-feedback circuit provides high input

impedance and stable wide-band gain for general purposevideo amplifier applications

8


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TLH679125

Stable Low Frequency Crystal Oscillator

This Colpitts-Crystal oscillator is ideal for low frequencycrystal oscillator circuits Excellent stability is assured be-

cause the 2N3823 JFET circuit loading does not vary withtemperature

TLH679126

0 to 360 Phase Shifter

Each stage provides 0 to 180 phase shift By ganging thetwo stages 0 to 360 phase shift is achieved The 2N3070JFETs are ideal since they do not load the phase shift net-works

TLH679127

DTL-TTL Controlled Buffered Analog Switch

This analog switch uses the 2N4860 JFET for its 25 ohmrONand low leakage The LM102 serves as a voltage buffer

This circuit can be adapted to a dual trace oscilloscope

chopper The DM7800 monolithic IC provides adequateswitch drive controlled DTL-TTL logic levels

TLH679128

20 MHz OSCILLATOR VALUES

C1 j 700 pF L1 e 13 mH

C2 e 75 pF L2 e 10T DIA LONG

VDD e 16V ID e 1 mA

20 MHz OSCILLATOR PERFORMANCE

LOW DISTORTION 20 MHz OSC

2ND HARMONIC b60 dB

3RD HARMONIC l b70 dB

Low Distortion Oscillator

The 2N4416 JFET is capable of oscillating in a circuit where

harmonic distortion is very low The JFET local oscillator

is excellent when a low harmonic content is required for a

good mixer circuit

9


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TLH679129

200 MHz Cascode Amplifier

This 200 MHz JFET cascode circuit features low crossmo-

dulation large-signal handling ability no neutralization andAGC controlled by biasing the upper cascode JFET The

only special requirement of this circuit is that lDSS of the

upper unit must be greater than that of the lower unit

TLH679130

FET Op Amp

The FM3954 monolithic-dual provides an ideal low-offsetlow-drift buffer function for the LM101A op amp The excel-

lent matching characteristics of the FM3954 track well over

its bias current range thus improving common mode rejec-tion

TLH679131

High Toggle Rate High Frequency Analog Switch

This commutator circuit provides low impedance gate driveto the 2N3970 analog switch for both on and off drive condi-

tions This circuit also approaches the ideal gate drive con-ditions for high frequency signal handling by providing a low

ac impedance for off drive and high ac impedance for ondrive to the 2N3970 The LH0005 op amp does the job of

amplifying megahertz signals

10


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TLH679132

4-Channel Commutator

This 4-channel commutator uses the 2N4091 to achieve lowchannel ON resistance (k30X) and low OFF current leak-

age The DM7800 voltage translator is a monolithic device

which provides from a10V to b20V gate drive to theJFETs while at the same time providing DTL-TTL logic com-

patability

TLH679134

Current Monitor

R1senses current flow of a power supply The JFET is usedas a buffer because lD e lS therefore the output monitor

voltage accurately reflects the power supply current flow

11


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TLH679135

Low Cost High Level Preamp and Tone Control Circuit

This preamp and tone control uses the JFET to its best

advantage as a low noise high input impedance device Alldevice parameters are non-critical yet the circuit achieves

harmonic distortion levels of less than 005% with a SN

ratio of over 85 dB The tone controls allow 18 dB of cut and

boost the amplifier has a 1 volt output for 100 mV input atmaximum level

TLH679136lO e

VIN

RlVIN s 0V

Precision Current Source

The 2N3069 JFET and 2N2219 bipolar serve as voltagedevices between the output and the current sensing resis-tor R1 The LM101 provides a large amount of loop gain to

assure that the circuit acts as a current source For smallvalues of current the 2N2219 and 10k resistor may be elimi-

nated with the output appearing at the source of the2N3069

TLH679137

Schmitt Trigger

This Schmitt trigger circuit is emitter coupled and provides

a simple comparator action The 2N3069 JFET places verylittle loading on the measured input The 2N3565 bipolar is a

high hFE transistor so the circuit has fast transition actionand a distinct hysteresis loop

12


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TLH679138

Low Power Regulator Reference

This simple reference circuit provides a stable voltage refer-ence almost totally free of supply voltage hash Typical

power supply rejection exceeds 100 dB

TLH679139

High Frequency Switch

The 2N4391 provides a low on-resistance of 30 ohms and ahigh off-impedance (k02 pF) when off With proper layout

and an ideal switch the performance stated above canbe readily achieved

13


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AN-32

FETCircuit

Applications

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National doesnot assumeany responsibilityfor useof anycircuitry described nocircuit patent licenses areimplied and National reserves the right at anytime without noticeto changesaid circuitryand specifications


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17/105Analog Dialogue 41-08, August (2007) 1

If All Else Fails, Read This Article

Avoid Common Problems WhenDesigning Amplifier Circuits

By Charles Kitchin [[email protected]]

INTRODUCTIONModern operational amplifiers (op amps) and instrumentation

amplifiers (in-amps) provide great benefits to the designer,

compared with assemblies of discrete semiconductors. A great

many clever, useful, and tempting circuit applications have been

published. But all too often, in ones haste to assemble a circuit,

some very basic issue is overlooked that leads to the circuit not

functioning as expectedor perhaps at all.

This article will discuss a few of the most common application

problems and suggest practical solutions.

Missing DC Bias Current Return Path When AC-CoupledOne of the most common application problems encountered is the

failure to provide a dc return path for bias current in ac-coupled

operational- or instrumentation-amplifier circuits. In Figure 1, a

capacitor is connected in series with the noninverting (+) input ofan op amp to ac couple it, an easy way to block dc voltages that are

associated with the input voltage (VIN). This should be especially

useful in high-gain applications, where even a small dc voltage

at an amplifiers input can limit the dynamic range, or even

result in output saturation. However, capacitively coupling into

a high-impedance input, without providing a dc path for current

flowing in the + input, will lead to trouble!

Figure 1. A malfunctional ac-coupled op-amp circuit.

What actually happens is that the input bias currents will flow

through the coupling capacitor, charging it, until the common-mode

voltage rating of the amplifiers input circuit is exceeded or the output

is driven into limits. Depending on the polarity of the input bias

current, the capacitor will charge up toward the positive supply

voltage or down toward the negative supply. The bias voltage is

amplified by the closed-loop dc gain of the amplifier.This process can take a long time. For example, an amplifier with

a field-effect-transistor (FET) input, having a 1-pA bias current,

coupled via a 0.1-F capacitor, will have a charging rate, I/C, of1012/107= 10 V/s, or 600 V per minute. If the gain is 100,the output will drift at 0.06 V per minute. Thus, a casual lab test

(using an ac-coupled scope) might not detect this problem, and the

circuit wil l not fail until hours later. Obviously, it is very important

to avoid this problem altogether.

http://www.analog.com/analogdialogue

Figure 2. Correct method for ac coupling an op-amp

input for dual-supply operation.

Figure 2 shows a simple solution to this very common problem

Here, a resistor is connected between the op-amp input and ground

to provide a path for the input bias current. To minimize offse

voltages caused by input bias currents, which track one anothe

when using bipolar op amps, R1 is usually set equal to the paralle

combination of R2 and R3.

Note, however, that this resistor will always introduce some noise

into the circuit, so there will be a trade-off between circuit inpuimpedance, the size of the input coupling capacitor needed, and

the Johnson noise added by the resistor. Typical resistor values are

generally in the range from about 100,000 to 1 M.

A similar problem can affect an instrumentation amplifier

circuit. Figure 3 shows in-amp circuits that are ac-coupled using

two capacitors, without providing an input-bias-current return

path. This problem is common with instrumentation amplifie

circuits using both dual- (Figure 3a) and single (Figure 3b)

power supplies.

Figure 3. Examples of nonfunctional ac-coupled

in-amp circuits.

The problem can also occur with transformer coupling, as

in Figure 4, if no dc return path to ground is provided in the

transformers secondary circuit.

Figure 4. A nonfunctional transformer-coupled in-amp circuit
mailto:[email protected]:[email protected]


18/105


19/105Analog Dialogue 41-08, August (2007) 3

For example, a popular in-amp design configuration uses three

op amps connected as above. The overall signal gain is

The gain for the reference input (if driven from low impedance)

is unity. However, in the case shown, the in-amp has its reference

pin tied directly to a simple voltage divider. This unbalances the

symmetry of the subtractor circuit and the division ratio of the

voltage divider. This would reduce the in-amps common-moderejection and its gain accuracy. However, if R4 is accessible, so

that its resistance value can be reduced by an amount equal to

the resistance looking back into the paralleled legs of the voltage

divider (50 k here), the circuit will behave as though a low-impedance voltage source equal to (in this example) one-half the

supply voltage were applied to the original value of R4, and the

subtractors accuracy would be maintained.

This approach cannot be used if the in-amp is provided as a

closed single package (an IC). Another consideration is that the

temperature coefficients of the resistors in the voltage divider

should track those of R4 and the other resistors in the subtractor.

Finally, the approach locks out the possibility of having the

reference be adjustable. If, on the other hand, one attempts to use

small resistor values in the voltage divider in an effort to make theadded resistance negligible, this will increase power supply current

consumption and increase the dissipation of the circuit. In any

case, such brute force is not a good design approach.

Figure 9 shows a better solution, using a low-power op-amp buffer

between the voltage divider and the in-amps reference input. This

eliminates the impedance-matching and temperature-tracking

problem and allows the reference to be easily adjustable.

Figure 9. Driving the reference pin of an in-amp from the

low-imdepance output of an op amp.

Preserving Power-Supply Rejection (PSR) When Amplifiers Are

Referenced from the Supply Rail Using Voltage DividersAn often overlooked consideration is that any noise, transients,

or drift of power-supply voltage, VS, fed in through the reference

input will add directly to the output, attenuated only by the

divider ratio. Practical solutions include bypassing and filtering,

and perhaps even generating the reference voltage with a precision

reference IC, such as the ADR121, instead of tapping off V S.

This consideration is important when designing circuits with both

in-amps and op amps. Power-supply rejection techniques are

used to isolate an amplifier from power supply hum, noise, and

any transient voltage variations present on the power rails. This i

important because many real-world circuits contain, connect to

or exist in environments that offer less-than-ideal supply voltage

Also, ac signals present on the supply lines can be fed back into

the circuit, amplified, and under the right conditions, stimulate

a parasitic oscillation.

Modern op amps and in-amps all provide substantial low-frequency

power-supply rejection as part of their design. This is somethingthat most engineers take for granted. Many modern op amps and

in-amps have PSR specs of 80 dB to over 100 dB, reducing the

effects of power-supply variations by a factor of 10,000 to 100,000

Even a fairly modest PSR spec of 40 dB isolates supply var iations

from the amplifier by a factor of 100. Nevertheless, high-frequency

bypass capacitors (such as those in Figure 1 through Figure 7) are

always desirable and often essential.

In addition, when designers use a simple resistance divider on the

supply rail and an op-amp buffer to supply a reference voltage

for an in-amp, any variations in power-supply voltage are passed

through this circuitry with little attenuation and add directly to

the in-amps output level. So, unless low-pass filtering is provided

the normally excellent PSR of the IC is lost.

In Figure 10, a large capacitor has been added to the voltage divide

to filter its output from power-supply variations and preserve PSR

The 3-dB pole of this filter is set by the parallel combination o

R1/R2 and capacitor C1. The pole should be set approximately

10 times lower than the lowest frequency of concern.

Figure 10. Decoupling the reference circuit to preserve PSR

The cookbook values shown provide a 3-dB pole frequency

of approximately 0.03 Hz. The small (0.01-F) capacitor acrossR3 minimizes resistor noise.

The filter will take time to charge up. Using the cookbook values

the rise time at the reference input is several time constants (where

T= R3Cf= 5 s), or about 10 to 15 seconds.


20/1054 Analog Dialogue 41-08, August (2007)

The circuit of Figure 11 offers a further refinement. Here, the

op-amp buffer is operated as an active filter, which allows the use

of much smaller capacitors for the same amount of power-supply

decoupling. In addition, the active filter can be designed to provide

a higher Q and thus give a quicker turn-on time.

Figure 11. An op-amp buffer connected as an active filter

drives the reference pin of an in-amp.

Test results: With the component values shown, and 12 V applied,

a 6-V filtered reference voltage was provided to the in-amp.

A 1-V p-p sine wave of varying frequency was used to modulate

the 12-V supply, with the in-amp gain set to unity. Under these

conditions, as frequency was decreased, no ac signal was visible

on an oscilloscope, at VREF or at the in-amp output, until

approximately 8 Hz. Measured supply range for this circuit was

4 V to greater than 25 V, with a low-level input signal applied to

the in-amp. Circuit turn-on time was approximately 2 seconds.

Decoupling Single-Supply Op-Amp CircuitsFinally, single-supply op-amp circuits require biasing of the input

common-mode level to handle the positive and negative swings

of ac signals. When this bias is provided from the power-supplyrail, using voltage dividers, adequate decoupling is required to

preserve PSR.

A common and incorrect practice is to use a 100-k/100-kresistor divider with a 0.1 F bypass capacitor to supply VS/2to the noninverting pin of the op amp. Using these values,

power-supply decoupling is often inadequate, as the pole

frequency is only 32 Hz. Circuit instability (motor-boating)

often occurs, especially when driving inductive loads.

Figure 12 (noninverting) and Figure 13 (inverting) show

circuits to accomplish VS/2 decoupled biasing for best results. In

both cases, bias is provided at the noninvert ing input, feedback

causes the inverting input to assume the same bias, and unity

dc gain also biases the output to the same voltage. Coupling

capacitor C1 rolls the low-frequency gain down toward unity

from BW3.

LOAD

Figure 12. A single-supply noninverting amplifier circuit,

showing correct power-supply decoupling. Midband

gain = 1 + R2/R1.

A good rule of thumb when using a 100 k/100 kvoltage divider,as shown, is to use a C2 value of at least 10F for a 0.3-Hz 3-dBroll-off. A value of 100 F (0.03-Hz pole) should be sufficient forpractically all circuits.

LOAD

Figure 13. Proper decoupling for a single-supply

inverting-amplifier circuit. Midband gain = R2/R1.


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Application Report

SLOA064 July 2001

1

A Differential Op-Amp Circuit Collection

Bruce Carter High Performance Linear Products

ABSTRACT

All op-amps are differential input devices. Designers are accustomed to working withthese inputs and connecting each to the proper potential. What happens when there aretwo outputs? How does a designer connect the second output? How are gain stagesand filters developed? This application note will answer these questions and give a

jumpstartto apprehensive designers.

1 INTRODUCTION

The idea of fully-differential op-ampsis not new. The first commercial op-amp, the K2-W,utilized two dual section tubes (4 active circuit elements) to implement an op-amp with

differential inputs and outputs. It required a 300 Vdc power supply, dissipating 4.5 W of power,

had a corner frequency of 1 Hz, and a gain bandwidth product of 1 MHz(1)

.

In an era of discrete tube or transistor op-amp modules, any potential advantage to be gainedfrom fully-differential circuitry was masked by primitive op-amp module performance. Fully-differential output op-amps were abandoned in favor of single ended op-amps. Fully-differentialop-amps were all but forgotten, even when IC technology was developed. The main reasonappears to be the simplicity of using single ended op-amps. The number of passive componentsrequired to support a fully-differential circuit is approximately double that of a single-endedcircuit. The thinking may have been Why double the number of passive components whenthere is nothing to be gained?

Almost 50 years later, IC processing has matured to the point that fully-differential op-amps arepossible that offer significant advantage over their single-ended cousins. The advantages ofdifferential logic have been exploited for 2 decades. More recently, advanced high-speed A/Dconverters have adopted differential inputs. Single-ended op-amps require a problematictransformer to interface to these differential input A/D converters. This is the application thatspurred the development of fully-differential op-amps. An op-amp with differential outputs,however, has far more uses than one application.

2 BASIC CIRCUITS

The easiest way to construct fully-differential circuits is to think of the inverting op-amp feedback

topology. In fully-differential op-amp circuits, there are two inverting feedback paths:

Inverting input to noninverting output

Noninverting input to inverting output

Both feedback paths must be closed in order for the fully-differential op-amp to operate properly.


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SLOA064

2 A Differential Op-Amp Circuit Collection

When a gain is specified in the following sections, it is a differential gain that is the gain atV

OUT+with a return of V

OUT-. Another way of thinking of differential outputs is that each signal is

the return path for the other.

2.1 A New Pin

Fully-differential op-amps have an extra input pin (VOCM

). The purpose of this pin is to provide aplace to input a potentially noisy signal that will appear simultaneously on both inputs i.e.common mode noise. The fully-differential op-amp can then reject the common mode noise.

The VOCM

pin can be connected to a data converter reference voltage pin to achieve tight trackingbetween the op-amp common mode voltage and the data converter common mode voltage. Inthis application, the data converter also provides a free dc level conversion for single supplycircuits. The common mode voltage of the data converter is also the dc operating point of thesingle-supply circuit. The designer should take care, however, that the dc operating point of thecircuit is within the common mode range of the op-amp + and inputs. This can most easily beachieved by summing a dc level into the inputs equal or close to the common mode voltage.

2.2 Gain

A gain stage is a basic op-amp circuit. Nothing has really changed from the single-endeddesign, except that two feedback pathways have been closed. The differential gain is still R

f/R

in

a familiar concept to analog designers.

-Vcc

+VccRf

Vout-

Ri n

Ri n

Gain = Rf/Rin

-

-

+C M

+

1

8

2

3

6

5

4Vin-

Vout+

Rf

Vocm

Vin+

Figure 1: Differential Gain Stage

This circuit can be converted to a single-ended input by connecting either of the signal inputs toground. The gain equation remains unchanged, because the gain is the differential gain.

2.3 Instrumentation

An instrumentation amplifier can be constructed from two single-ended amplifiers and a fully-differential amplifier as shown in Figure 2. Both polarities of the output signal are available, ofcourse, and there is no ground dependence.


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A Differential Op-Amp Circuit Collection 3

VocmGain = (R2/R1)*(1+2*R5/R6)

R1=R3

R2=R4

R5=R7

Vout-

-

-

+CM

+

1

8

2

3

6

5

4

+Vcc

-Vcc

Vout+

-

+

-Vcc

+Vcc

+Vcc

Vin+

-

+Vin-

-Vcc

Figure 2: Instrumentation Amplifier

3 FILTER CIRCUITS

Filtering is done to eliminate unwanted content in audio, among other things. Differential filtersthat do the same job to differential signals as their single-ended cousins do to single-endedsignals can be applied.

For differential filter implementations, the components are simply mirror imaged for each

feedback loop. The components in the top feedback loop are designated A, and those in thebottom feedback loop are designated B.

For clarity decoupling components are not shown in the following schematics. Proper operationof high-speed op-amps requires proper decoupling techniques. That does not mean a shotgun

approach of using inexpensive 0.1-F capacitors. Decoupling component selection should bebased on the frequencies that need to be rejected, and the characteristics of the capacitors usedat those frequencies.

3.1 Single Pole Filters

Single pole filters are the simplest filters to implement with single-ended op-amps, and the same

holds true with fully-differential amplifiers.A low pass filter can be formed by placing a capacitor in the feedback loop of a gain stage, in amanner similar to single-ended op-amps:


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SLOA064


R2B

R1A

C1B

C1A

Vocm

+Vcc

Vin-

R1B

Vout+-

-

+CM

+

1

8

2

3

6

5

4

Vin+

-Vcc

Vout-

fo=1/(2**R2*C1)gain=-R2/R1

R2A

Figure 3: Single Pole Differential Low Pass Filter

A high pass filter can be formed by placing a capacitor in series with an inverting gain stage asshown in Figure 4:

125

-Vcc

fo=1/(2**R1*C1)gain=-R2/R1

Vout+Vocm

Vout-

R1A

Vin-

C1A

C1B

R2B

R1B

R2A

Vin+

+Vcc

-

-

+CM

+

1

8

2

3

6

5

4

Figure 4: Single Pole Differential High Pass Filter

3.2 Double Pole Filters

Many double pole filter topologies incorporate positive and negative feedback, and thereforehave no differential implementation. Others employ only negative feedback, but use thenoninverting input for signal input, and also have no differential implementation. This limits thenumber of options for designers, because both feedback paths must return to an input.

The good news, however, is that there are topologies available to form differential low pass, highpass, bandpass, and notch filters. However, the designer might have to use an unfamiliartopology or more op-amps than would have been required for a single-ended circuit.


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3.2.1 Multiple Feedback Filters

MFB filter topology is the simplest topology that will support fully-differential filters.Unfortunately, the MFB topology is a bit hard to work with, but component ratios are shown forcommon unity gain filters.

Reference 5 describes the MFB topology in detail.

C1A

R1B

+VccR2A

Vout-

Vin-

Chebyshev 3 dB

Fo=1/(2

RC)

R1=0.644R

R2=0.456R

R3=0.267R

C1=12C

C2=C

Bessel

Fo=1/(2

RC)

R1=R2=0.625R

R3=0.36R

C1=C

C2=2.67C

R1A

Vocm

R2B

C2A

R3A

-

-

+CM

+

1

8

2

3

6

5

4

C2B

-Vcc

R3B

Vout+

Butterworth

Fo=1/(2

RC)

R1=R2=0.65R

R3=0.375R

C1=C

C2=4C

C1B

Vin+

Figure 5: Differential Low Pass Filter


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R1B

C1B

Vocm

Bessel

Fo=1/(2

RC)

R1=0.73R

R2=2.19R

C1=C2=C3=C

R2B

Butterworth

Fo=1/(2

RC)

R1=0.467R

R2=2.11R

C1=C2=C3=C

Chebyshev

Fo=1/(2

RC)

R1=3.3R

R2=0.215R

C1=C2=C3=C

R2A

Vout+

R1A -

-

+CM

+

1

8

2

3

6

54

Vin-

+Vcc

C2B

C2A

C3A

-Vcc

C1A

Vin+

C3B

Vout-

Figure 6: Differential High Pass Filter

There is no reason why the feedback paths have to be identical. A bandpass filter can beformed by using nonsymmetrical feedback pathways (one low pass and one high pass). Figure7 shows a bandpass filter that passes the range of human speech (300 Hz to 3 kHz).

88.7 k

R2

-VCC

Vout-

86.6 k

R5

1 nFC2

Vin+

19.1 kR4

270 pF

C1

Vout+

+VCC

Vin-

41.2 kR3

-

-

+CM

+

THS4121

U1

1

8

2

3

6

5

4

22 nFC4

Vcm

10 nFC3

22 nF

C5

100 k

R1

Figure 7: Differential Speech Filter


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SLOA064


Figure 8: Differential Speech Filter Response

3.2.2 Akerberg Mossberg Filter

Akerberg Mossberg filter topology is a double pole topology that is available in low pass, highpass, band pass, and notch. The single ended implementation of this filter topology has anadditional op-amp to invert the output of the first op-amp. That inversion in inherent in the fully-differential op-amp, and therefore is taken directly off the first stage. This reduces the totalnumber of op-amps required to 2:

ButterworthFo=1/(2RC)R2=R3=RR4=0.707RC1=C2=CGain: R/R1

-

-

+CM

+

U2

1

8

2

3

6

5

4

Vocm

R4B

+Vcc

Vin+

R2B

C1A

Vocm

-

-

+CM

+

U1

1

8

2

3

6

5

4

+Vcc

R4A

R1B

-Vcc-Vcc

Vout-

C2A

Vin-

C2B

BesselFo=1/(2RC)R2=R3=0.786RR4=0.453RC1=C2=CGain: R/R1

Vout+

R3B

R1A

ChebyshevFo=1/(2RC)R2=R3=1.19RR4=1.55RC1=C2=CGain: R/R1

R2A

R3A

C1B


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Figure 9: Akerberg Mossberg Low Pass Filter

R1A

Vout-

C1A

C3A

VocmVout+

Vin-

BesselFo=1/(2RC)R1=R2=1.27RR3=0.735RC2=C3=CGain: C1/C

ButterworthFo=1/(2RC)R1=R2=RR3=0.707RC2=C3=CGain: C1/C

+Vcc

R2B

Vin+

R3B

R3A

R2A

R1B

C2B-Vcc

C1B

-Vcc

C2A

ChebyshevFo=1/(2RC)R1=R2=0.84RR3=1.1RC2=C3=CGain: C1/C

-

-

+CM

+

U2

1

8

2

3

6

5

4

Vocm

-

-

+CM

+

U1

1

8

2

3

6

5

4

C3B

+Vcc

Figure 10: Akerberg Mossberg High Pass Filter

R1BVin+

Vocm

Vin-

C1A C2A

R1A

Vout-

Fo=1/(2RC)R2=R3=RR4=Q*R

Gain: -R4/R1C1=C2=C

-

-

+CM

+

U2

1

8

2

3

6

5

4

C2BC1B

Vocm

+Vcc

-Vcc-Vcc

+Vcc

R4A

Vout+

R4B

R3A

R3B

R2A

-

-

+CM

+

U1

1

8

2

3

6

5

4

R2B

Figure 11: Akerberg Mossberg Band Pass Filter


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SLOA064


+Vcc

C2B

R2A

Vout-

R3A

-Vcc

Vocm

C2A

R4B

C3A

-Vcc

-

-

+CM

+

U2

1

8

2

3

6

5

4

R1A

Fo=1/(2

RC)

R1=R2=R3=R

R4=Q*R

C1=C2=C3=C

Unity gain

C1B

Vin-

R1B

Vin+

R4A

C3B

VocmVout+

+Vcc

R3B

C1A

R2B

-

-

+CM

+

U1

1

8

2

3

6

5

4

Figure 12: Akerberg Mossberg Notch Filter

3.2.3 Biquad Filter

Biquad filter topology is a double pole topology that is available in low pass, high pass, bandpass, and notch. The highpass and notch versions, however, require additional op-amps, andtherefore this topology is not optimum for them. The single-ended implementation of this filtertopology has an additional op-amp to invert the output of the first op-amp. That inversion isinherent in the fully-differential op-amp, and therefore is taken directly off the first stage. Thisreduces the total number of op-amps required to 2:


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SLOA064


Butterworth

Fo=1/(2RC)

R3=RR2=0.707RGain: -R2/R1

C1=C2=C

R2A

Vocm

LPout-

-

-

+CM

+

U2

1

8

2

3

6

5

4-

-

+CM

+

U1

1

8

2

3

6

5

4

-Vcc

Chebyshev

Fo=1/(2RC)

R3=1.19RR2=1.55RGain: -R1/R2

C1=C2=C

R1B

R1A

R2B

C2A

Vin-

C2B

R3A

+Vcc

R4A

BPout+

R3B

R4B

C1B

LPout+

BPout-

C1A

LOWPASS

Vin+

BANDPASS

Fo=1/(2RC)R3=R

C1=C2=CGain= -R2/R1

R2=Q*R

+Vcc

Vocm

-Vcc

Bessel

Fo=1/(2RC)

R3=0.785RR2=0.45RGain: -R2/R1

C1=C2=C

Figure 13: Differential BiQuad Filter

4 Driving Differential Input Data Converters

Most high-resolution, high-accuracy data converters utilize differential inputs instead of single-ended inputs. There are a number of strategies for driving these converters from single-endedinputs.

A/D Common Mode Output

-

+

A/D -Input

Vi nA/D +Input

Figure 14: Traditional Method of Interfacing to Differential-Input A/D Converters

In Figure 14, one amplifier is used in a noninverting configuration to drive a transformer primary.The secondary of the transformer is center tapped to provide a common-mode connection pointfor the A/D converter V

refoutput.


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A/D +Input

-

+


A/D -Input

-

+

Vi n

Figure 15: Differential Gain Stage With Inverting Single-Ended Amplifiers

Gain can be added to the secondary side of the transformer. In Figure 15, two single-ended opamps have been configured as inverting gain stages to drive the A/D Inputs. The non-invertinginput inputs are connected to the transformer center tap and A/D V

refoutput.

-

+


A/D -InputVin-

A/D +Input-

+

Figure 16: Differential Gain Stage With Noninverting Single-Ended Amplifiers

Figure 16 shows how single-ended amplifiers can be used as noninverting buffers to drive theinput of an A/D. The advantage of this technique is that the unity gain buffers have exact gains,so the system will be balanced.

Transformer interfacing methods all have one major disadvantage:

The circuit does not include dc in the frequency response. By definition, the transformerisolates dc and limits the ac response of the circuit.

If the response of the system must include dc, even for calibration purposes, a transformer is aserious limitation.

A transformer is not strictly necessary. Two single-ended amplifiers can be used to drive an A/Dconverter without a transformer:


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-

+

-

+


A/D +InputVi n

A/D -Input

Figure 17: Differential Gain Stage With Noninverting Single-Ended Amplifiers

Although all of the methods can be employed, the most preferable method is the use a fully-differential op-amp:


-

-

+CM

+

1

8

2

3

6

5

4Vi n

+Vcc

-Vcc

A/D -Input

A/D +Input

Figure 18: Preferred Method of Interfacing to a Data Converter

A designer should be aware of the characteristics of the reference output from the A/Dconverter. It may have limited drive capability, and / or have relatively high output impedance. Ahigh-output impedance means that the common mode signal is susceptible to noise pickup. Inthese cases, it may be wise to filter and/or buffer the A/D reference output:

A/D Vref Output

-

+

Optional Buffer

Op Amp Vocm Input

Figure 19: Filter and Buffer for the A/D Reference Output


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SLOA064


Some A/D converters have two reference outputs instead of one. When this is the case, thedesigner must sum these outputs together to create a single signal as shown in Figure 20:

Op Amp Vocm Input

A/D Vref- Output

A/D Vref+ Output

-

+

Optional Buffer

Figure 20: Filter and Buffer for the A/D Reference Output

5 Audio Applications

5.1 Bridged Output Stages

The presence of simultaneous output polarities from a fully-differential amplifier solves a probleminherent in bridged audio circuits the time delay caused by taking a single-ended output andrunning it through a second inverting stage.

SPEAKER

-

+

Power Amp 1

-

+

Power Amp 2

INPUT

Figure 21: Traditional Bridge Implementation

The time delay is nonzero, and a degree of cancellation as one peak occurs slightly before theother when the two outputs are combined at the speaker. Worse yet, one output will contain oneamplifiers worth of distortion, while the other has two amplifiers worth of distortion. Assumingtraditional methods of adding random noise, that is a 41.4% noise increase in one output withrespect to the other, power output stages are usually somewhat noisy, so this noise increase willprobably be audible.

A fully-differential op-amp will not have completely symmetrical outputs. There will still be afinite delay, but the delay is orders of magnitude less than that of the traditional circuit.


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+

-

-

+C M

+

Differential Stage

1

8

2

3

6

5

4INPUT

SPEAKER

-

Figure 22: Improved Bridge Implementation

This technique increases component count and expense. Therefore, it will probably be more

appropriate in high end products. Most fully-differential op-amps are high-speed devices, andhave excellent noise response when used in the audio range.

5.2 Stereo Width Control

Fully-differential amplifiers can be used to create an amplitude cancellation circuit that willremove audio content that is present in both channels.


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R3100 k

+

C44.7 F

R1100 k

Rout

Lin

R4

100 k

-

+

U4

-

+

U2

+Vcc +Vcc

+

C14.7 F

R9100 k

R10

100 k

-

-

+CM

+

U1

1

8

2

3

6

5

4

-

-

+CM

+

U31

8

2

3

6

5

4

-Vcc

+

C24.7 F

R14

100 k

-Vcc

-Vcc

R11100 k

+Vcc

Rin

R8

100 k

+

C34.7 F

+Vcc

R5B

10 kPot

R13

100 k

R5A10 k Pot

R6

100 k

R15

100 k

R7

100 k

Lout

R2100 k

-Vcc

R12100 k

Figure 23: Stereo Width Control

The output mixers (U2 and U4) are presented with an inverted version of the input signal on oneinput (through R6 and R14), and a variable amount of out-of-phase signal from the otherchannel.

When the ganged pot (R5) is at the center position, equal amounts of inverted and noninvertedsignal cancel each other, for a net output of zero on the other input of the output mixers (throughR7 and R13).

At one extreme of the pot (top in this schematic), the output of each channel is the sum of theleft and right channel input audio, or monaural. At the other extreme, the output of each mixer isdevoid of any content from the other channel canceling anything common between them.

This application differs from previous implementations by utilizing fully-differential op-amps tosimultaneously generate inverted and noninverted versions of the input signal. The usualmethod of doing this is to generate an inverted version of the input signal from the output of abuffer amp. The inverted waveform, therefore, is subject to two op-amp delays as opposed toone delay for the non-inverted waveform. The inverted waveform, therefore, has some phasedelay which limits the ultimate width possible from the circuit. By utilizing a fully-differential op-amp, a near perfect inverted waveform is available for cancellation with the other channel.


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6 Summary

Fully-differential amplifiers are based on the technology of the original tube-based op-amps ofmore than 50 years ago. As such, they require design techniques that are new to mostdesigners. The performance increase afforded by fully differential op-amps more than outweighthe slight additional expense of more passive components. Driving of fully differential A/Dconverters, data filtering for DSL and other digital communication systems, and audioapplications are just a few ways that these devices can be used in a system to deliverperformance that is superior to single-ended design techniques.

References

1. Electrical Engineering Times, Design Classics, Unsung Hero Pioneered Op-Amp,http://www.eetimes.com/anniversary/designclassics/opamp.html

2. Fully-differential Amplifiers, Texas Instruments SLOA054A

3. A Single-supply Op-Amp Circuit Collection, Texas Instruments SLOA058

4. Stereo Width Controllers, Elliot Sound Products, http://www.sound.au.com/project21.htm5. Active Low-Pass Filter Design, Texas Instruments SLOA049A


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Copyright 2001, Texas Instruments Incorporated


38/105

TLH8501

Instrumenta

tionAmplifier

LB-1

National SemiconductorLinear Brief 1March 1969

Instrumentation Amplifier

The differential input single-ended output instrumentationamplifier is one of the most versatile signal processing am-

plifiers available It is used for precision amplification of dif-ferential dc or ac signals while rejecting large values of com-mon mode noise By using integrated circuits a high level of

performance is obtained at minimum cost

Figure 1 shows a basic instrumentation amplifier which pro-vides a 10 volt output for 100 mW input while rejecting

greater than g11V of common mode noise To obtain goodinput characteristics two voltage followers buffer the inputsignal The LM102 is specifically designed for voltage fol-

lower usage and has 10000 MXinput impedance with 3 nAinput currents This high of an input impedance provides two

benefits it allows the instrumentation amplifier to be usedwith high source resistances and still have low error and itallows the source resistances to be unbalanced by over

10000X with no degradation in common mode rejectionThe followers drive a balanced differential amplifier asshown in Figure 1 which provides gain and rejects the com-mon mode voltage The gain is set by the ratio of R

4to R

2and R5 to R3 With the values shown the gain for differen-

tial signals is 100

Figure 2shows an instrumentation amplifier where the gainis linearly adjustable from 1 to 300 with a single resistor An

LM101A connected as a fast inverter is used as an attenu-ator in the feedback loop By using an active attenuator a

very low impedance is always presented to the feedbackresistors and common mode rejection is unaffected by gain

changes The LM101A used as shown has a greater band-width than the LM107 and may be used in a feedback net-work without instability The gain is linearly dependent on R 6and is equal to 10b4 R6

To obtain good common mode rejection ratios it is neces-sary that the ratio of R4 to R2match the ratio of R5 to R3

For example if the resistors in circuit shown in Figure 1 hada total mismatch of 01% the common mode rejectionwould be 60 dB times the closed loop gain or 100 dB The

circuit shown in Figure 2 would have constant commonmode rejection of 60 dB independent of gain In either cir-

cuit it is possible to trim any one of the resistors to obtaincommon mode rejection ratios in excess of 100 dB

For optimum performance several items should be consid-ered during construction R1 is used for zeroing the output

It should be a high resolution mechanically stable potenti-ometer to avoid a zero shift from occurring with mechanical

disturbances Since there are several ICs operating in closeproximity the power supplies should be bypassed with001 mF disc capacitors to insure stability The resistors

should be of the same type to have the same temperaturecoefficient

A few applications for a differential instrumentation amplifier

are differential voltage measurements bridge outputsstrain gauge outputs or low level voltage measurement

TLH85011

FIGURE 1 Differential-Input Instrumentation Amplifier

C1995 National Semiconductor Corporation RRD-B30M115Printed in U S A


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LB-1

Instrumentat

ionAmplifier

GAIN ADJUST

Av e 10b4 R6

TLH85012

FIGURE 2 Variable Gain Differential-Input Instrumentation Amplifier

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1

AN1298.2

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.

Copyright Intersil Americas Inc. 2007, 2009. All R ights ReservedAll other trademarks mentioned are the property of their respective owners.

Instrumentation Amplifier Application Note

Table of Contents

Introduction to the Instrumentation Amplifier................................................................................................................................... 2

Review of Standard Instrumentation Amplifier Design Techniques ................................................................................................ 2

Monolithic Instrumentation Amplifier Architecture........................................................................................................................... 4

Introduction to Instrumentation Amplifier Product Family................................................................................................................ 4

Instrumentation Amplifier Specifications ......................................................................................................................................... 4

Instrumentation Amplifier Product Family Theory of Operation....................................................................................................... 6

Features of Instrumentation Amplifier Product Family .................................................................................................................... 7

Care and Feeding of Instrumentation Amplifiers............................................................................................................................. 10

Application Circuits.......................................................................................................................................................................... 20

Pressure Sensor Interface Circuit ................................................................................................................................................... 21

Thermocouple Input with A/D Converter Output ............................................................................................................................. 22

Thermocouple Input with 4mA to 20mA Output Current ................................................................................................................. 23

RTD Input with A/D Converter Output............................................................................................................................................. 24

Low Voltage High Side Current Sense............................................................................................................................................ 27

Multiplexed Low Voltage Current Sense ......................................................................................................................................... 30

Bi-Directional Current Sense........................................................................................................................................................... 32

Applicat ion Note May 27, 2009


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2 AN1298.2May 27, 2009

Introduction to the InstrumentationAmplif ier

This Application Note describes the Intersil bipolar and MOS

input (see Table 1). Instrumentation Amplifiers, theory of

operation, advantages, and typical application circuits.

These devices are micropower Instrumentation Amplifiers

which deliver rail-to-rail input amplification and rail-to-rail

output swing on a single 2.4V to 5V supply. TheseInstrumentation Amplifiers deliver excellent DC and AC

specifications while consuming only 60A typical supply

current. Because they provide an independent pair of

feedback terminals to set the gain and to adjust output level,

these Instrumentation Amplifiers achieve high

common-mode rejection ratios regardless of the tolerance of

the gain setting resistors. The ISL28271 and ISL28272 have

an ENABLE pin to reduce power consumption, typically less

than 5.0A, while the Instrumentation Amplifier is disabled.

An Instrumentation Amplifier is a confused animal

confused by its cousin, the op amp.

Its symbol looks like an op amp (see Figure 1)

It has many of the same basic properties andspecifications as an op amp Offset Voltage, Input BiasCurrent, CMRR, PSRR, etc.

You can make an Instrumentation Amplifier from a simpleop amp circuit.

But the behavior of an Instrumentation Amplifier is

profoundly different than an op amp! And it is very difficult to

make a precision Instrumentation Amplifier from a simple op

amp circuit many have tried, but most have failed.

An Instrumentation Amplifier provides a voltage subtraction

block followed by a fixed gain block; i.e.

Often, there is an optional output reference input which

allows the output voltage to be shifted by a fixed voltage:

In contrast, an op amp by definition only provides extremely

high gain with provisions to apply negative feedback to

establish a fixed gain or unique transfer function, H(s), such

as an integrator or filter.

Review of Standard InstrumentationAmpl if ier Design Techniques

Difference Amplifier

In its most basic topology, an Instrumentation Amplifier can

be configured from a single op amp and four resistors as

shown in Figure 4; this is often referred to as a Difference

Amplifier.

TABLE 1.

PARTINPUTSTAGE

# OFAMPLIFIERS

MINIMUM

CLOSEDLOOP GAIN

BW(kHz) ENABLE?

EL8170 Bipolar 1 100 192 No

EL8171 PMOS 1 10 450 No

EL8172 PMOS 1 100 170 No

EL8173 Bipolar 1 10 396 No

ISL28270 Bipolar 2 100 240 No

ISL28271 PMOS 2 10 180 Yes

ISL28272 PMOS 2 100 100 Yes



OP AMP

+VCC

-VCC

VOUT

+

-

+VCC

-VCC

VOUT

+

-

INSTRUMENTATION AMPLIFIER

FIGURE 1.

R1

R2R1R2

IN-

IN+

VOUT= (IN+ - IN-) * (1 + R2/R1)+

-

+

-

FIGURE 2. TWO OP AMP INSTRUMENTATION AMPLIFIER

VOU T IN+ IN-( ) Gain= (EQ. 1)

VOU T IN+ IN-( ) Gain VRE F+= (EQ. 2)

IN+

IN-VOUT

VREF

GAIN

+

-

FIGURE 3.

Appl ication Note 1298
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3 AN1298.2May 27, 2009

In this configuration, the gain is set by resistors R1and R2:

For the ability to reject a voltage that appears on both IN-

and IN+ (i.e., common mode voltage), resistor values must

match such that R1= R3and R2= R4. The common mode

rejection ratio (CMRR) is set by the matching ratio of R1:R3

and R2:R4. High common mode rejection ratio requires a

very high degree of ratio matching.

It can be shown that the CMRR is:

Worse case CMRR occurs when the tolerance of R4and R1

are at their maximum, and R2and R3are at their minimum

value. The following table shows the relationship between

resistor tolerance and CMRR for gains of 1, 10, and 100.

The Difference Amplifier has the advantage of simplicity and

the ability to operate with high common mode voltage on its

inputs, IN+ and IN-. However, the input resistance is set by

the resistor values R3and R4, and does not provide high

input resistance as is common in most InstrumentationAmplifier circuits.

Additionally, the REF input must be driven by a very low

source impedance since the CMRR will be degraded by any

source resistance that contributes to the value of R4and

causes increased mismatch between R2and R4.

Also note that the common mode voltage will bias internal

nodes at a voltage that is set by the ratio of R3and R4, or the

gain of the circuit. For example, in Figure 5, for a gain of 100

and a common voltage of 10V, the inputs to the op amp will

be sitting at a voltage of 9.9V. This circuit would not be

possible if the op amp was operated with VCCof +5V since

the op amp inputs voltage would exceed the supply voltage.

Two Amplifier Instrumentation Amplifier

To provide a high input impedance, a two amplifier

Instrumentation Amplifier can be used as in Figure 6.

In this configuration, the gain is set by resistors R3and R4:

The ability to reject a voltage that appears on both IN- and

IN+ (i.e., common mode voltage), depends on matched

resistor values such that, R1= R3and R2= R4. The common

mode rejection ratio (CMRR) is set by the matching ratio of

R1:R3and R2:R4, and, high CMRR requires a very high

degree of ratio matching. For example, with 10V of common

mode voltage, resistor tolerances must be at least 0.01%

to achieve 12-bit accuracy (72dB).

Classic Three Amplifier Instrumentation Amplifier

By adding a third op amp, the Classic Three Amplifier

Instrumentation Amplifier can be configured as shown in

Figure 7.

TABLE 2.

RESISTOR CMRR

TOLERANCE GAIN =1 GAIN = 10 GAIN =100

5% -20.4dB -15.6dB -14.8dB

1% -34.1dB -28.9dB -28.1dB

0.1% -54.0dB -48.8dB -48.0dB

0.01% -74.0dB -68.8dB -68.0dB

R1

R3

R2

R4

IN+

IN-

VREF

VOUT+

-

FIGURE 4.

Gain R2 R1= (EQ. 3)


CMRR 20 10(x)log= (EQ. 5)

Where x R4 R3 R4+( ) R1 R2+( ) R1 R2 R1= (EQ. 6)

1k

VREF

VOUT+

-

VCC

Vcm= 10V

R1

1k

R3

100k

R4

100k

R2

FIGURE 5.

R3

R4R1R2

IN-

IN+

VOUT+

-

+

-

FIGURE 6. TWO AMPLIFIER INSTRUMENTATION AMPLIFIER

Gain 1 R4 R3+=(EQ. 7)

VOU T IN+ IN-( ) Gain= (EQ. 8)



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4 AN1298.2May 27, 2009

Usually, resistors R1through R6are equal value resistors of

R and the gain:

With this circuit, the Gain can be set with a single resistor,

RGAINand the input impedance is very high. However, the

common mode rejection ratio, CMRR, just like the Difference

Amplifier topology, is still set by the resistor matching

between R1, R2, R3, and R4. Extremely low tolerance

resistors or precision resistor trimming is required to achieve

high CMRR. The equations and Table shown for the

Difference Amplifier apply directly to the Classic Three

Amplifier Instrumentation Amplifier configuration.

Monolithic Instrumentation AmplifierArchitecture

Each of the three basic Instrumentation Amplifier architectures

that have been already discussed have been implemented in

standard integrated circuit packages. To achieve a high CMRR,

extensive resistor trimming is required with lasers or other

suitable techniques. While each of these devices provide

adequate specifications for a precision Instrumentation

Amplifier, each device has its own compromise based on

operating voltage range, supply current, common mode

operating range, input impedance, etc. These instrumentation

amplifiers use one external resistor to set the gain; while this

may seem to be an advantage, there are considerations which

make the single resistor configuration undesirable from a

design viewpoint. The temperature coefficient (TC) of theexternal resistor will be a direct gain drift. Also, an external filter

can not be applied to the feedback network because it is

internal to the device.

Introduction to Instrumentation AmplifierProduct Family

This Application Note describes the Intersil Instrumentation

Amplifier Product Family, which includes the following

features:

1. Bipolar transistor inputs for low voltage noise

2. PMOS transistor inputs for low input bias current

3. Micropower operation requiring only 60A supply current

4. Rail-to-rail inputs and rail-to-rail output swing

5. Single supply operation from 2.4V to 5V supply

6. An independent pair of feedback terminals to set the gainand to adjust output level allow these Instrumentation

Amplifier to achieve high CMRR (>104dB) regardless ofthe tolerance of the gain setting resistors.

7. Internal loop compensation to provide optimumbandwidth trade-off as shown in Table 1

8. The ISL28271 and ISL28272 have an ENABLE pin toreduce the supply current to a typical of less than 5A andtri-state the output stage to a high impedance state.

Instrumentation Amplifier Specifications

Many of the Instrumentation Amplifier specifications are very

similar to the standard specifications for operational

amplifiers. However, the unique architecture of the Intersil

Instrumentation Amplifiers make some of these

specifications differ slightly. Table 3summarizes the

Specifications and Features of the Instrumentation Amplifier

Product Family.

FIGURE 7. CLASSIC THREE AMPLIFIER INSTRUMENTATION

AMPLIFIER

R1

R3

R5

R6

Rg

R2

R4

VOUT

VREF

IN-

IN+

+

-

+

-

+

-

Gain 1 2 R Rgain+( )= (EQ. 9)


IN+

IN-

FB+

FB-

VOUT

VIN V1

V2

V3

V4

Rf

Rg

VOUT


http://-/?-http://-/?-http://-/?-http://-/?-


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5 AN1298.2May 27, 2009

TABLE 3.

PARAMETERS EL8170 ISL28270 ISL28470 EL8173 ISL28273 EL8171 ISL28271 EL8172 ISL28272 UNITS

Input Stage Bipolar Bipolar PMOS PMOS

Minimum Gain 100 10 10 100

Gain Set 2 Ext R 2 Ext R 2 Ext R 2 Ext R

Supply Current: Enabled

per Channel

65 65 65 60 65 60 A

Supply Current: Shutdown - - - - - 4 - 4 A

Minimum VCC 2.4 2.4 2.4 2.4 VDC

Maximum VCC 5.5 5.5 5.5 5.5 VDC

Input Offset Voltage 200 150 150 1000 600 1500 600 300 500 V

Offset Drift 0.24 0.7 0.7 2.5 0.7 1.5 0.7 0.14 0.7 V/C

Input Bias Current, Maximum 3000 2000 2500 2000 2500 50 30 50 30 pA

Input Offset Current, Maximum 2000 2000 25 30 25 30 pA

Input Bias Current Cancellation Yes Yes - -

Bandwidth (-3dB) at AV= 10 - 396 265 450 180 - kHz

Bandwidth (-3dB) at AV= 100 192 240 240 - - 170 100 kHz

Slew Rate (Typ) 0.55 0.5 0.5 0.55 0.6 0.55 0.5 0.55 0.5 V/s

Rail-to-Rail Input Yes Yes Yes Yes

Rail-to-Rail Output Yes Yes Yes Yes

Output Current Limit, V+ = 5V 26 29 29 26 29 26 26 mA

Output in Shutdown Mode - - - HiZ - HiZ

Gain Accuracy 0.35 0.5 0.5 0.1 0.12 0.15 0.08 0.2 -0.19 %

CMRR (Typ) 114 110 110 106 110 100 100 dB

PSRR (Typ) 106 110 110 90 95 90 100 100 dB

eN at 1kHz 58 60 60 220 210 220 240 80 78 nv/HzeN 0.1Hz to 10Hz 3.5 3.6 3.5 14 10 10 6 VP-P

Input Protection - Diodes to

Rails

Yes Yes Yes Yes

Input Protection - Diodes

across Inputs

Yes No No No

Max Input Diode Current 5 5 5 5 mA

Package SO8 SO8 SO8 SO8

Operating Temp. Range -40 to +85 -40 to +85 -40 to +85 -40 to +85 C

RoHS Compliant Yes Yes Yes Yes



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Instrumentation Amplifier Product FamilyTheory of Operation

Each of the features specifications of the Intersil

Instrumentation Amplifier Product Family will be discussed in

more detail in a future section of this Application Note, but

first, lets study the internal operation of this unique

Instrumentation Amplifier Product Family.

A simplified schematic is shown in Figure 9.

Assuming high transistors:

Similarly for Q3and Q4:

Summing currents:

where A is the gain of the output stage

Assume Ry/Re= 1 (i.e., Reand Ryare equal value).

Since A is very large:

Let VIN= V2 V1, and V3= FB+, V4= FB-

or

As you can see from Equation 31, negative feedback is

applied around the amplifier so that the voltage applied to

the feedback terminals (FB+ - FB-) must be equal to the

voltage applied to the input terminals (IN+ - IN-).

FIGURE 9. SIMPLIFIED SCHEMATIC

Q1 Q2

Re

Q3 Q4

Re

I I I

V1V2 V3

V4

IN- IN+ FB-FB+

Ry Ry

VOUT

GAIN = A

I1 I2 I3 I4

Ix1 Ix2

+Ven

Va Vb

I5 I6

V5

V6

I

Ix1

V2 Vbe2+( ) V1 Vbe1+( )

Re---------------------------------------------------------------------- and since,=

(EQ. 11)

Vbe1 Vbe 2=

Ix1

V2 V1

Re-------------------- =

I1 I Ix1 I V2 V1( ) Re+=+= (EQ. 12)

I2 I Ix1 I V2 V1( ) Re== (EQ. 13)

I3 I Ix2 I V4 V3( ) Re+=+= (EQ. 14)

I4 I Ix2 I V4 V3( ) Re== (EQ. 15)

I5 I2 I3 I V2 V1( ) Re I V4 V3( ) Re+ +=+= (EQ. 16)

I5 2 I V1 V2( ) Re V4 V3( ) Re++= (EQ. 17)

I6 I1 I4 I V2 V1( ) Re I V4 V3( ) Re++=+= (EQ. 18)

I6 2 I V2 V1( ) Re V3 V4( ) Re++= (EQ. 19)

V5 I5 Ry 2 Ry I V1 V2( ) Ry Re V4 V3( ) Ry Re++==

(EQ. 20)

V6 I6 Ry 2 Ry I V2 V1( ) Ry Re V3 V4( ) Ry Re++==(EQ. 21)

VOU T A V5 V6( )= (EQ. 22)

VOU T A 2 Ry I V1 V2( ) V4 V3( )+ +2 Ry I V2 V1( ) V3 V4( )+ +[ ]

[]

=

(EQ. 23)

VOU T A V1 V2( ) V4 V3( ) V1 V2( ) V4 V3( )+ + +[ ]=

(EQ. 24)

VOU T 2 A V1 V2( ) V4 V3( )+[ ]= (EQ. 25)

VOU T 2 A( ) V1 V2( ) V4 V3( )+[ ]= (EQ. 26)

VOU T 2 A( ) 0 (EQ. 27)

0 V1 V2( ) V4 V3( )+= (EQ. 28)

0 -V IN FB- FB+( )+= (EQ. 29)

VIN FB- FB++ (EQ. 30)

IN+ IN- FB- FB+= (EQ. 31)



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7 AN1298.2May 27, 2009

For the standard data sheet connection:

Features of Instrumentation AmplifierProduct Family

A simplified schematic and block diagram is shown in

Figure 11to illustrate the rail-to-rail operation for both the

input stage and the output stage. The same schematic

applies to the PMOS input devices when the PNP transistors

(Q1 to Q4) are replaced with P-Channel MOSFETs for

ultra-low input bias current.

The input terminals (IN+ and IN-) and feedback terminals (FB+

and FB-) are single differential pair devices aided by an Input

Range Enhancement Circuit to increase the headroom of

operation of the common-mode input voltage. As a result, the

input common-mode voltage range for all these Instrumentation

Amplifiers is rail-to-rail. The parts are able to handle input

voltages that are at or slightly beyond the supply and ground

making these in-amps well suited for single 5V or 3.3V low

voltage supply systems. There is no need then to move the

common-mode input voltage of the these Instrumentation

Amplifiers to achieve symmetrical input voltage.

The use of a bipolar transistor input stage vs. the MOSFET

input stage allows the user to choose low bias current, high

input resistance.

Rail-to-rail operation for both the inputs and outputs is an

important and unique feature. The rail-to-rail inputs allow the

input voltages to be slightly below the VS- rail (typically

Ground) to slightly above the VS+ rail.

The conventional technique to achieve a rail-to-rail input

stage is to use two separate input stages, as shown in

Figure 12. One input stage (Q1and Q2) provides common

mode input range to the top rail (VS+), and the other input

stage (Q3and Q4) provides common mode input range to

the bottom rail.

IN+

IN-

FB+

FB-

VOUT

VIN V1

V2

V3

V4

Rf

Rg

VOUT


FB+ 0V=

FB- VOU T Rg Rg Rf+( )=

VIN F

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