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A high efficient integrated planar transformer for
primary-parallel isolated boostconverters
Sen, Gökhan; Ouyang, Ziwei; Thomsen, Ole Cornelius; Andersen,
Michael A. E.; Møller, Lars
Published in:Energy Conversion Congress and Exposition
Link to article, DOI:10.1109/ECCE.2010.5618405
Publication date:2010
Document VersionPublisher's PDF, also known as Version of
record
Link back to DTU Orbit
Citation (APA):Sen, G., Ouyang, Z., Thomsen, O. C., Andersen, M.
A. E., & Møller, L. (2010). A high efficient integrated
planartransformer for primary-parallel isolated boost converters.
In Energy Conversion Congress and Exposition (pp.4605-4610). IEEE.
https://doi.org/10.1109/ECCE.2010.5618405
https://doi.org/10.1109/ECCE.2010.5618405https://orbit.dtu.dk/en/publications/f6043cba-4b96-48fd-83f0-fb46f5e9374fhttps://doi.org/10.1109/ECCE.2010.5618405
-
A High Efficient Integrated Planar Transformer for
Primary-Parallel Isolated Boost Converters
Gokhan Sen1, Ziwei Ouyang1, Ole C. Thomsen1, Michael A. E.
Andersen1, and Lars Møller2
1. Department of Electrical Engineering, 2. H2 Logic A/S
Technical University of Denmark, Herning, DK-7400, Denmark Kgs.
Lyngby, DK-2800, Denmark, [email protected]
Abstract -- A simple, easy to manufacture and high efficient
integrated planar transformer design approach for primary parallel
isolated boost converters is presented. Utilizing the same phase
flux flow, transformers are integrated, reducing the total ferrite
volume and core loss for the same peak flux density. Number of
turns is minimized for easy manufacturing by cascade placement of
planar cores increasing the effective cross-sectional area. AC
losses in the windings as well as the leakage inductance of the
transformer are kept low by extensive interleaving of the primary
and secondary turns. The idea of transformer integration is further
extended to multiple primary power stages using modular geometry of
the planar core, further reducing the core loss and allowing a
higher power density. To verify the validity of the design
approach, a 4-kW prototype converter with two primary power stages
is implemented for a fuel cell fed battery charger application with
50-110 V input and 65-105 V output. Input inductors are coupled for
current sharing, eliminating the use of current sharing
transformers. An efficiency of 94% is achieved during nominal
operating condition where the input is 70-V and the output is
84-V.
Index Terms—planar integrated magnetics, coupled inductor,
isolated boost converter, fuel cell.
I. INTRODUCTION
Traction drive systems based on fuel cells and batteries require
the dc-dc converter components to be selected for a wide range of
input and output voltages depending on the fuel cell current and
battery state-of-charge. This requirement limits the efficiency
compared to a converter optimized around an operating point since
the components have to be selected to cover the limits. So the
dc-dc converter should be designed carefully to compensate for this
situation [1].
Transformer design is a critical stage in high performance dc-dc
converter design. In isolated boost converter applications, leakage
inductance of the transformer should be minimized as well as the ac
resistance since it causes spikes on the primary switch voltages
increasing the inductive clamp losses [2]. Planar transformers have
unique advantages in terms of increased power density, better
cooling capability, modularity and manufacturing simplicity which
make them attractive for high current dc-dc converter applications
[3], [10]. In [4] and [5] integrated planar transformers for dc-dc
converters have been studied. Improvements of AC resistance in
transformer design are presented in [7], [8] and [9]. Also optimal
design of transformers in high power, high frequency applications
is mentioned in [11], [12] and [15].
Recently, significant achievements have been obtained in simple,
low cost and high performance paralleling of
converters for handling high currents in fuel cell applications
which are presented in [13] and [14].
In this paper a new approach in transformer design for
primary-parallel secondary-series isolated boost converters is
presented. The idea is verified by simulation and experimental
results.
II. PRIMARY PARALLEL ISOLATED BOOST CONVERTER
Boost derived topologies are preferred in fuel cell applications
due to their low input current ripple. Fig.1 shows a primary
parallel isolated boost derived topology suitable for handling high
input currents for fuel cell applications. The dc current is forced
to be equal in both primary stages by the series secondary
connection of the two transformers. In order to compensate for the
non-zero differential voltages in the two power stages that may
occur due to gate signaling delays or parameter mismatches, a
direct coupled input inductor is used, instead of a current sharing
transformer. This coupled inductor divides the input current into
two parts reducing the conduction losses and integrating two
windings into a single core with a greater equivalent
inductance.
In this topology, primary power stages share the same control
signals with same phase switching sequence for the corresponding
switches which allows a simpler control. Output rectification unit
as well as input and output filters are common to both of the
primary stages. Details of this topology and basic waveforms are
presented in [2] and [13].
Fig. 1. Primary parallel isolated boost converter with coupled
inductors for current sharing
978-1-4244-5287-3/10/$26.00 ©2010 IEEE 4605
-
Transformers T1 and T2 have the same number of turns with the
same core size and material. Since the ac current sharing is
ensured by the input coupled inductor, the same flux flows in both
transformer cores. If it would be possible to wind both of the
windings into the same core, they would produce the same peak flux
density which would correspond to half the core loss since the
total volume of ferrite is halved. But this method is not practical
for standard E-cores due to the limited winding space; however it
is possible to be achieved using planar E- cores by utilizing their
modularity.
III. TRANSFORMER DESIGN
This section presents the new integrated planar transformer
designed for the primary parallel isolated boost converter in Fig.
1, based on the arguments in the previous section. The new design
provides higher power density, lower cost and higher efficiency
compared to two separate transformers with the same core type.
Planar structure also allows PCB windings to be used which make the
manufacturing simpler and cheaper considering volume production.
Also it allows easier realization for the interleaved winding
arrangement.
A. Winding arrangement
Winding losses in transformers increase dramatically
with increasing frequency due to skin and proximity effects.
Based on Dowell’s assumptions and the general field solutions for
the distribution of current density in a single layer of an
infinitely long foil conductor, the expression for AC resistance of
a certain layer can be derived as in [6] and [9],
2, ,sinh sin sinh sin(2 1)
2 cosh cos cosh cosac m dc mR R mξ ξ ξ ξ ξ
ξ ξ ξ ξ⎡ ⎤+ −= ⋅ + − ⋅⎢ ⎥− +⎣ ⎦
(1)
where ξ is defined as
hξδ
= (2)
and δ is the skin depth for a specific frequency, which is given
by
1f
δπ μ σ
=⋅ ⋅ ⋅
(3)
m is defined as
( )( ) (0)
F hmF h F
=−
(4)
where (0)F and ( )F h are Magneto Motive Forces (MMF) at the
limits of a layer as shown in Fig. 2. The first term in equation
(1) describes the skin effect and the second term represents the
proximity effect. The proximity effect loss in a multilayer
winding, may strongly dominate the skin effect loss if the value of
m increases which is related to the winding arrangement.
Interleaving transformer windings can reduce the proximity loss
significantly (decrease m) when the primary and secondary currents
are in phase. In this research
3:1 turns ratio is used for each transformer. There are 3 turns
in series on the primary side; and a single turn with four layers
in parallel on the secondary side to sustain high current. Fig. 2
shows the MMF distributions along the vertical direction for the
interleaved arrangement of the designed transformer. Fig. 3 shows
the AC resistance relative to the DC resistance in a layer having a
thickness equal to the skin depth. The layer thickness can be
selected for minimizing the AC losses according to the value of m.
In this case switching frequency is 50-kHz and 0.14-mm PCB layer
thickness has been chosen for each layer.
B. Cascaded structure
Proposed integrated planar transformer has been built
using a cascaded core structure to minimize the number of turns.
Fig. 4 shows the construction of each transformer with two planar
E-cores placed side-by-side which adds up to four
Fig. 3. AC resistance relative to dc resistance in a layer with
different values of thickness over skin depth.
Fig.2. MMF distributions for interleaved arrangement for the
proposed transformer.
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-
E-cores and two I-cores for the whole integrated structure.
Increased cross-section by cascading avoids saturation and reduces
core loss when the same number of turns is used in a non-cascaded
(single) core construction. According to Faraday’s law peak-to-peak
flux density is,
e
V tBN A
⋅ ΔΔ =⋅
(5)
where N, Ae are the number of turns and the cross-section of
centre leg of the core respectively. V·Δt is volt-seconds applied
to the transformers. Core loss of a transformer depends on the
material that the core is made from, the switching frequency, the
flux density, and the volume of the core. The Steinmetz Equation
can be used to compute the core loss given in equation (6),
( / 2) eP K f B Vα β= ⋅ ⋅ Δ ⋅ (6)
K, α and β are constants that are provided by the manufacturer
or can be calculated from the curves of a specific core loss data.
Combining equation (5) with equation (6) and having the ratio of
the two different cases,
12cc coP Pβ−= ⋅ (7)
where Pcc and Pco represent the core losses of 2 cores cascaded
transformer and a single core transformer with the same number of
turns, respectively. Generally β is more than 1, therefore the
cascaded transformer has lower core loss according to eqn. (7). It
has to be noted that winding loss will be sacrificed due to longer
turn lengths. In this work 3F3 core material is used which has β
approximately equal to 2.6.
C. Magnetics Integration
In order to reduce the transformer footprint area and increase
the converter power density, transformers T1 and T2 in Fig. 1 are
integrated utilizing the modular flexibility of the planar cores.
The new structure is shown in Fig. 4. Two integrated transformers
are located on top of each other. Two cascaded I-cores are shared
in the centre part. The symmetrical windings of the two
transformers will allow the flux in the I-cores to be cancelled
which reduce the core loss. 3D FEA simulation result in Fig.7 shows
the flux in I-core (centre part) to be almost zero. More
transformers can be
integrated based on this principle which will further reduce the
overall core loss since the flux will only flow in the outer core
sections (Fig. 5).
D. Magnetic reluctance model
Fig.6 shows the magnetic reluctance model of the integrated
structure where R1 and R2 are the reluctances of outer legs of
E-core and half I-core respectively. Similarly Rc represents the
reluctance of the center leg of E-core.
Based on the equivalent magnetic model in Fig.6, major flux
directions can be determined. In order to calculate the magnetizing
inductance of the integrated module referred to the secondary side,
the equations (8) and (9) can be obtained assuming the primary
windings to be shorted,
1 1 1 1 2 1 2 20.5 ( ) 0.5 ( )s ms cN i R R R Rφ φ φ φ⋅ = ⋅ + ⋅
⋅ + + ⋅ − ⋅ (8) 2 2 2 1 2 2 1 20.5 ( ) 0.5 ( )s ms cN i R R R Rφ φ
φ φ⋅ = ⋅ + ⋅ ⋅ + + ⋅ − ⋅ (9)
where ims is the magnetizing current referred to the secondary
side. Since the two transformers, T1 and T2 are identical,
Fig.4. Two transformers in E-I-E integration.
X X X
. . .X X X
. . .
Fig.5. Integration of n number of transformers with main flux
lines (green) and cancelled flux lines (blue).
Fig.6. Reluctance model of the integrated transformer.
4607
-
Ø1=Ø2 and Ns1=Ns2 can be used. Therefore, the magnetizing
inductance of each transformer in the integrated module referred to
the secondary side can be derived as,
2
'
1 2
22
sm
c
NLR R R
=+ +
(10)
The magnetizing inductance of a single separated transformer
would have the following expression,
2
1 2
22 2
sm
c
NLR R R
=+ +
(11)
Comparing magnetizing inductances of the integrated and
separated transformers using (10) and (11), L’m > Lm can be
observed which means the integrated transformers have higher
magnetizing inductance compared to the separated case. This is
because of the fact that the flux cancellation occurs in the shared
I-core effectively reducing the length of flux path. Higher
magnetizing inductance reduces the magnetizing current which helps
in the current stress over the components. E. Measurement
results
Fig.8 and Fig.9 show the leakage inductance and AC resistance
measurement results of the designed single transformer and the
integrated transformers respectively using PSM1735 impedance
analyzer. For the single transformer, the leakage inductance is
25.76-nH and the AC resistance is 2.50-mΩ referred to secondary
side when the frequency is 50-kHz. This measurement has been taken
by shorting the primary side and connecting the secondary terminals
to the impedance analyzer. For the integrated transformers, the
total leakage inductance referred to the secondary side is 40.25-nH
and the total AC resistance is 5.53-mΩ. Fig.10 shows the
measurement results of magnetizing inductances. Results confirm the
previous analysis that the total magnetizing inductance of the two
integrated transformers is greater than that of the two series
connected separated transformers.
Fig.7. 3D FEA simulation for integrated transformers
Fig.8. Measurement results of ac resistance and leakage
inductance of a single transformer referred to the secondary
side
Fig.9. Measurement results of ac resistance and leakage
inductance referred to the secondary side for the deigned
integrated transformer
Fig.10. Measurement results of magnetizing inductances of both
single and integrated transformers referred to secondary side
Table I summarizes the design parameters for the integrated
transformer. Each PCB winding has 8 layers where 4 layers used in
parallel for the single secondary turn and the remaining 4 layers
are used for three primary turns with one of the turns used two
layers in parallel.
4608
-
Table І
Design parameters for the integrated transformers
Parameters Values
Number of turns in primary of T1 (Np1) 3
Number of turns in primary of T2 (Np2) 3
Turns ratio of each transformer 3:1
Planar Core type EILP 64
Core material 3F3
Number of layers of PCB winding 8
Copper thickness for each layer of PCB winding
4 OZ (140um)
AC resistance referring to the secondary side 5.53mΩ Leakage
inductance referring to the
secondary side 40.25 nH
Magnetizing inductance referring to the secondary side 31.3
uH
IV. EXPERIMENTAL RESULTS
A 4-kW prototype converter has been built to verify the new
integrated transformer design approach. Input voltage is between
50-110 V and output voltage is 65-105 V. Primary switches are
IRF4668, 200-V, 8-mΩ power MOSFETs from International Rectifier.
Output rectification is handled by 80CPQ150 schottky diodes with
0.82-V forward voltage drop.
The two integrated transformers are implemented using four E-64
and two I-64 planar cores with 3F3 material. The primary and
secondary windings are implemented using PCB windings with 0.14-mm
layer thickness.
The coupled input inductor is wound on a KoolMμ 6527E040 core
with 0.4-mm copper foil having 12 turns in each winding. In order
to reduce the AC conduction losses of the inductor, 0.1 mm copper
foil is used for a parallel inner winding with the same number of
turns giving a lower AC resistance. This two parallel winding
approach helps in providing a low AC resistance and leakage
inductance path for the inductor current ripple.
IRS2110 high and low side gate drivers are used in the gate
driver circuit together with ISO722C capacitive digital isolators
for control signal protection. Open loop control signals are
produced by TMS28027 DSP from Texas Instruments. Fig. 11 shows the
implemented prototype with all the switches mounted to a heat sink
beneath the converter. Output is filtered by six 20-uF and two
820-nF film capacitors placed very close to the rectifiers for
minimizing the ac loop.
Converter waveforms are presented in Figs. 12 and 13. Coupled
inductor current waveforms, iL1 and iL2, are observed to be very
close in average value confirming the current sharing function of
the coupled inductor. An efficiency of 94% has been observed with
70-V input voltage, 84-V output voltage and 3-kW input power.
Fig. 11. Experimental prototype of integrated transformer
primary parallel isolated boost converter
Fig. 12. Converter waveforms (Vin: 70V, Vo: 84V). Ch4: iL2
(10A/div), Ch2: iL1 (10A/div), Ch1: iTP1 (25A/div), Ch3: VT1
(200V/div).
Fig. 13. Converter waveforms (Vin: 70V, Vo: 84V). Ch2: iO
(20A/div), Ch1: iin ripple (25A/div), Ch3: VDS of S2 (100V/div),
Ch4: VGS of 2S (20V/div).
4609
-
93,2%
94,6%
90,0091,00
92,0093,00
94,0095,0096,00
97,0098,00
99,00100,00
1000 2000 3000 4000 5000
Effi
cien
cy (%
)
Input power (W)
Fig. 14. Converter efficiency(Vin: 70V, Vo: 84V).
V. CONCLUSION
A new method is proposed for transformer integration in primary
parallel isolated boost converters. Utilizing the symmetry and
in-phase switching in each power stage, cores of the transformers
have been integrated sharing the same flux. Any small difference in
flux generated by each integrated transformer is compensated by the
“I” cores placed in between. Number of turns is minimized by
cascade placement of E cores per transformer, decreasing the
effective cross sectional area. This simplifies manufacturing and
decreases the winding losses. Extensive interleaving has been used
for minimizing ac resistance and leakage inductance. The new
integration method has been tested in a 4-kW prototype.
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