A 32 GHz Low-Power Low-Phase-Noise VCO … in SiGe BiCMOS Technology . ... the low-phase-noise oscillator core make it ... compensating the losses of the inductor represented by resistor

A 32 GHz Low-Power Low-Phase-Noise VCO

Implemented in SiGe BiCMOS Technology

Yuhua Qi, Rulong He, and Zhao Shen Electronic Engineering College, Naval University of Engineering, PLA, Wuhan 430033, China

Email: [email protected]; [email protected]; [email protected]

Abstract—A low-phase-noise, low-power Ka-band Voltage-

Controlled Oscillator (VCO) using cross-coupled pair

configuration is presented. The Ka-band VCO circuit uses 0.18

μm SiGe BiCMOS technology. The VCO has low phase noise

of -114.6 dBc/Hz at 1 MHz offset from 32.12 GHz carrier

frequency and can be tuned from 30.17 to 33.48 GHz. The

figure of merit is -202.1 dBc/Hz. The power consumption of the

VCO with 0.27 mm2 chip area is 1.8 mW from a 1.5 V power

supply.

Index Terms—VCO, low power, low phase noise, SiGe

BiCMOS

I. INTRODUCTION

The recent advances in wireless communications

indicate a trend toward integrating multiple

communication standards into a single device. Such

devices could be implemented with multiple dedicated

front-ends integrated in parallel. However, that solution

requires larger silicon area and possibly more power

consumption. Reconfigurable radio transceivers, on the

other hand, optimize the functionality versus power and

area tradeoffs by programming a wideband front-end to

the desired standard. In such multistandard reconfigurable

radio transceivers, one of the most challenging building

blocks is the frequency synthesizer, which needs to

generate the local-oscillator signal with both a wide

tuning range and sufficiently high spectrum purity [1],

Phase-locked loops are widely used in wireless

communication systems. For broadband or multiband

systems, such as TV broadcasting systems and Global

Navigation Satellite System [1]. Voltage-controlled

oscillators (VCOs) are crucial building blocks in radio-

frequency phase-locked loops (RF PLLs), which are

usually used as frequency synthesizers to provide stable

and reliable frequency sources in wireless communication

systems [2], [3].

VCOs are important blocks in any communication or

test and measurement system. They are widely used in

wireless and wired receiving, data communication and

clock recovery systems [1]. Hence, extensive efforts have

Manuscript received December 19, 2016; revised February 28, 2017.

This work was supported by the Advance Research project of China under Grant No. 51307xxxx09, and the Advance Research project of

PLA in China under Grant No. KJ2013××××. Corresponding author email: [email protected].

doi: 10.12720/jcm.12.2.98-104

been made to improve the performance of VCOs. Several

parameters to evaluate the performance of a VCO are

available, including oscillation frequency, oscillation

amplitude, frequency tuning range, phase noise and

power consumption. Frequency tuning range and phase

noise are the most critical two parameters among all these

specifications [2], [3]. In this paper, we focus on the

improvement of power and the reduction of phase noise.

Traditionally, such VCO circuits have been the

exclusive domain of III-V compound semiconductors

such as GaAs or InP [4], [5]. However, over the past

several years, silicon and silicon germanium (SiGe) have

emerged as low cost alternatives that deliver the

performance of III-V compound semiconductors. A

number of RF VCOs have been reported using SiGe

BiCMOS [6]-[8], and CMOS technology [9]-[11]. With

rapid developments in silicon technologies, these systems

become feasible to be implemented using BiCMOS or

CMOS to achieve low cost, low power, and high

integration. In summary, the VCOs which are highly

integrated, having low phase noise, low power and wide

band are the trend for the future. So this paper presents

the design of an integrated millimeter wave VCO which

has a low power consumption.

The phase noise of VCO is determined by the variable

device used in the resonating tank, for instance, varactor

diodes commonly control the phase noise in a typical LC

VCO. Therefore, in an LC VCO varactor size should be

carefully chosen to decrease the phase noise. At

millimeter-wave frequencies, two topologies (cross-

[9]) are widely

used. However, it is well known that the maximum

oscillation frequency of a Colpitts VCO is higher than

that of a cross-coupled VCO, meaning that a Colpitts

VCO can achieve relatively lower phase noise. One way

to further extend the output frequency range is to use

push-push oscillators. However, as the frequency

increases, the output power from a push-push oscillator is

usually low, as reported in. The output power can be

improved through a tuning network at the output, but

only for a narrow bandwidth [11]. It can be concluded

that both topologies are capable of very good phase noise.

However, it is well known that the maximum oscillation

frequency of a Colpitts VCO is higher than that of a

cross-coupled VCO.

Much attention has been devoted to the power

consumption performance of oscillators in the past, while

Journal of Communications Vol. 12, No. 2, February 2017

98©2017 Journal of Communications

[2].

[6], coupled VCO [7] and Colpitts VCO

phase noise was not concerned with much. Since power

consumption is precious and more stages lead to more

parasitic problems and risks in circuit design at high

frequency, the low-phase-noise oscillator core make it

predominant in applications, especially where cost, power

consumption and reliability are important. In recent years,

some researches on oscillator phase-noise were carried

out. By introducing transform method into oscillator

design, low phase noise was achieved but power rose at

the same time. In this paper, a Ka-band cross-coupled

VCO was made with low phase noise and low power for

direct use without any additional stage. For this purpose,

the analysis for low power consumption has been done in

detail and the inductors of the resonating tank were

constructed by microstrip lines to obtain high quality

factor (Q). Design optimizations of the resonating tank

were made carefully, and an excellent result was achieved.

The cross-coupled VCO achieves an oscillation

frequency range from 30.17 to 33.48 GHz. The phase

noise is -114.6 dBc/Hz at 1 MHz offset from the

oscillation frequency 32.12 GHz, which is very excellent

result for low-power Ka-band oscillators.

II. CIRCUIT DESIGN

A. Design for Low Power

A general LC-tuned VCO can be symbolized as in Fig.

1. The oscillator consists of an inductor L and a capacitor

C, building a parallel resonance tank, and an active

element -R, compensating the losses of the inductor

represented by resistor RL in Fig. 1, and the losses of the

capacitor represented by the resistor RC. This circuit

results in an oscillator with angular center frequency:

C

1

LC (1)

The general LC-tuned VCO of Fig. 1 can be redrawn

in Fig. 2, neglecting the capacitor losses, as the series

resistance of the integrated inductors largely dominates

the tank losses. Using the energy conservation theorem,

the maximal energy in the inductor must equal the

maximal energy stored in the capacitor:

2 2

peak peak

2 2

CV LI (2)

where Vpeak is the peak amplitude voltage of the sine

wave voltage across the capacitor and Ipeak is the peak

amplitude current of the sine wave current through the

inductor. This current is flowing to the resistor RS, so

the effective loss in the tank can be calculated as:

2

peak2S Sloss peak

2 2

CVR RP I

L (3)

or with (4):

2 2 2S Sloss peak peak2 22 2

C

C

R RP C V V

L

(4)

L

C RC

RL

-R

Fig. 1. Basic LC-VCO.

L

C

RS

Fig. 2. Basic LC-resonator tank.

This loss must be compensated by the active part of the

VCO to sustain the oscillation. Ploss in the above equation

is the fundamental minimum for the power consumption

of a LC-VCO. The equations lead to some significant

conclusions for the design of low power consumption of

any LC-VCO:

1) Power consumption decreases linearly with the

series resistances in the resonance;

2) Power consumption decreases quadratically when

the tank inductance is increased so that the oscillator

frequency is assumed to be a constant;

3) Power consumption decreases quadratically when

the tank capacitance is decreased so that the oscillator

frequency is assumed fixed;

4) Power consumption decreases quadratically with the

oscillator frequency when the tank inductance is a

constant. Power consumption increases quadratically with

the oscillation frequency when the tank capacitance is a

constant.

B. Design for Low Phase N oise

The phase noise (PN) for a cross-coupled VCO is

given by [2]:

2

o

sig t ank

2PN 10log

2

kT

P Q

(5)

where K is Boltzmann constant, T is the absolute

temperature, ωo is the oscillation frequency, Qtank is the

quality factor of the LC resonating tank, Δω is the

frequency offset from the carrier, and Psig is the average



signal power consumption in the resistive part of the tank,

which is given by

tank

2

sig

P2

VP

R (6)

where Vtank is the voltage swing of the LC tank and RP is

the parallel equivalent resistance.

From (5) and (6), the phase noise can be deduced as

2

oP

2

t ank t ank

4PN 10log

2

kTR

V Q

(7)

Equation (7) indicates that the phase noise at a given

frequency offset depends on the voltage swing of the tank

Vtank and the quality factor of the LC resonating tank Qtank.

To improve the phase noise, one can increase either the

Vtank or the Qtank.

However, the Vtank is limited by the bias current of the

oscillator and the breakdown voltage of the active devices.

In order to obtain high Vtank, the size of the transistors

should be selected large enough to apply high bias current,

but elevating the bias current leads to high power

consumption. Therefore, there is always a compromise

between the high Vtank of an LC VCO and its power

dissipation.

The Qtank is another one of the most important

parameters in LC VCOs. It affects most VCO properties,

including the overall tank loss, and the phase noise. Qtank

is also a key factor for power calculation and estimation.

The better Qtank is, the lower power achieves. In the low-

frequency bands (<10 GHz), the tank is dominated by the

Q of the inductor (QL). However, as the frequency

increases to higher than 10 GHz, the Q of the varactors

(Qvar) decreases significantly [12]. Thus, the Qvar

determines the overall tank at the millimeter-wave

frequencies because Qvar is far less than QL.

tan k C L var

1 1 1 1

Q Q Q Q (8)

At the same time, the tuning range is also determined

by the varactor size. Thus, it is critical to obtain high-Q

and large-ratio (Cmax/Cmin) varactors to achieve better

VCO performance. However, there is a tradeoff between

the Qvar and the capacitance ratio. If the large capacitance

but the Qvar is minimized, in turn to degrade the phase

noise. In addition, the varactor should be carefully laid

out to reduce parasitic.

Although an oscillator is a strongly nonlinear circuit

which operates under large-signal conditions, an efficient

method to gain insight into the limitations of the available

transistor technology is to start off with a small signal (S-

parameter) analysis. Compared with transient or

harmonic balance simulations, small-signal simulations

are fast, do not suffer from nonconvergence problems,

and do not require initial conditions. This is particularly

important when simulating bipolar VCOs at millimeter-

wave frequencies, the transistor model must capture self-

heating and avalanche multiplication, which are two

positive feedback phenomena that cause significant

nonconvergence problems at large voltage swings.

During the transient simulations, the impedance of the S-

parameter signal source is set to a small value (i.e.,10–20),

to clearly reflect the expected effective resistance of the

tank inductance at the oscillation frequency, and thus to

allow for the onset of oscillations to occur.

As illustrated in Fig. 3, by plotting the real and

imaginary parts of the impedance looking into the base of

the HBT, Rin and Xin, respectively, one can quickly assess

whether the circuit produces an adequate negative

resistance at the desired oscillation frequency, fosc,

(approximately the frequency at which Xin=0 and Rin<0 ),

and one can estimate the tuning range of the VCO. From

Fig. 3, it can be concluded that fosc is about 32 GHz and

36.5 GHz when Vtune is equal to be 0 V and 1.5 V,

respectively. Fig. 4 shows the simulated oscillation

frequency of the VCO by the means of harmonic balance

simulation, it can be seen that the oscillation frequency

range is from 31.2 GHz to 35.89 GHz, which is in

accordance with that concluded with S-parameter analysis.

25 30 35 40-1.5x10

-2

-1.0x10-2

-5.0x10-3

0.0

5.0x10-3

1.0x10-2

1.5x10-2

2.0x10-2

Imp

eda

nce

/

Frequency /GHz

Rin @V

tune=0 V

Xin @V

tune=0 V

Rin @V

tune=1.5 V

Xin @V

tune=1.5 V

Fig. 3. Simulated input impedance as the tune voltage Vtune is changed

from 0 V to 5 V.

0.0 0.5 1.0 1.529

30

31

32

33

34

35

36

37

Fre

qu

ency

/G

Hz

Vtune

/V

Fig. 4. Simulated oscillation frequencies of the VCO.

III. IMPLEMENTATION AND MEASUREMENT RESULTS

The schematic of cross-coupled VCO designed in this

paper is shown in Fig. 5, in which the inductors (Ltank)

and the MOS varactors (Ctank) form the LC resonating

tank. A cross-coupled differential transistor pair (Q1 and

Q2) with positive capacitive feedback (C) provide the

necessary negative resistance to compensate the tank



(6).

ratio is chosen, the Tuning Range (TR) can be maximized,

losses for sustaining oscillation. A simple current source

(Io) provides a constant current for the VCO.

Ltank

Ctank

Vtune

VDD

Q2

Io

Ltank

Ctank

Q1

Vout- Vout+

Vbias

Rb Rb

C C

Fig. 5. Schematic of cross-coupled VCO designed in this paper.

The VCO has been designed and fabricated in a 0.18

μm SiGe BiCMOS technology. The die microphotograph

of the chip is shown in Fig. 6 and the whole chip area

including all testing pads is 0.47 × 0.58 mm2. The voltage

and current source HP4142B was used to supply the dc

voltages, meanwhile the output was connected through a

Ground-Signal-Ground (GSG) probe to the spectrum

analyzer Agilent N9030A with a phase-noise

measurement utility.

of the VCO

Fig. 7 shows the measured tuning characteristic of the

VCO versus Vtune. At the supply voltage of 1.5 V, the tail

current of the VCO core is set to be 1.2 mA current. The

tuning range is 3.31 GHz or 10.4% from 30.17 to 33.48

GHz when the Vtune changes from 0 to 1.5 V. The linear

range is 2 GHz from 31.02 to 33.02 GHz while the Vtune

varies from 0.45 to 1.05 V, thereby the linear gain of

VCO is calculated to be 3.33 GHz/V. The measured

phase noise at 32.12 GHz is shown in Fig. 8. At 100 kHz

and 1 MHz offset frequency from the carrier, the phase

noise is -78.8 dBc/Hz and -114.6 dBc/Hz, respectively.

In order to prove the feasibility of the design in the

paper, the comparison of simulated results and measured

results is shown in Table I.

TABLE I: COMPARISON OF SIMULATED RESULTS AND MEASURED

RESULTS

fosc (GHz) PN (dBc/Hz) △f

(MHz) PVCO (mW)

simulated 31.20～35.89 -116.8 1 2.0

measured 30.17～33.48 -114.6 1 1.8

0.0 0.5 1.0 1.529

30

31

32

33

34

35

Fre

qu

ency

/G

Hz

Vtune

/V

of the VCO

102

103

104

105

106

-140

-120

-100

-80

-60

-40

-20

0

Frequency offset (Hz)

Ph

ase

no

ise

(dB

c/H

z)

of the VCO

We compare the performance of VCO circuit at

different frequencies with previous reports, and the

measured characteristics are normalised to the figure of

merit (FOM), defined as:

osc VCOFOM PN 20log 10log1mW

f P

f

(9)

where PN is the measured phase noise at the offset

frequency Δf from the carrier frequency fosc. PVCO is the

VCO power consumption in mW. A larger |FOM| denotes

a better oscillator. The FOM of this 32 GHz VCO circuit

is -202.1 dBc/Hz at 1 MHz offset. Table II summarises

the measured performance of this VCO and includes

other reported performances for comparison. The VCO



Fig. 7. Measured oscillator frequency .

Fig. 6. Die microphotograph .

Fig. 8. Measured phase noise .

reported in this paper achieves a low power and low

phase noise, which compares well with other published

results [2], [6], [10]-[20].

The important parameters of a VCO are the center

oscillation frequency fosc, the phase noise PN measured at

an offset of Δf from fosc, the output power Pout, the power

dissipation PVCO, the tuning range TR achieved by

varying the tuning voltage Vtune, and the variation of

tuning voltage ΔVtune. To fairly compare the VCOs

reported in the available literatures, several figure-of-

merits (FOMs) have been presented.

in is the most widely used, but it

excludes the TR which is one of the most important

properties of a VCO. Even though some systems do not

require a wide range or need only operate at a single

frequency, a wide TR is nevertheless required because of

variations caused by the parasitics at millimeter-wave

frequencies. There is a direct tradeoff between the TR

and other properties. Therefore, the TR should be

included, as described in FOMT depicted in 10).

oscT

VCO

FOM PN 20log10%

10log1mW

f TR

f

P

(10)

However, FOMT does not take Pout into account, and as

a result, output power will not affect the FOMT even if a

VCO generates a very little output power. It may be

supposed that the output power is already considered in

the phase noise, but the phase noise is determined by the

tank swing, which should be distinguished from the

output power. As the tank swing is large, the output

power can be low if the output buffer is improperly

designed. In addition, if the output power is low, then

more buffers and more power dissipation are required to

satisfy the system specifications because mixers, for

example, require a large amount of LO power. Therefore,

the output power should be included, as described in

FOMTP depicted in 11).

oscTP

VCOout

FOM PN 20log10%

10log1mW

f TR

f

PP

(11)

Lastly, the ΔVtune is another important parameter of a

VCO, because there is a direct tradeoff between the

ΔVtune and the TR. The bigger the ΔVtune is, the wider the

TR achieves. The importance of ΔVtune is described in

FOMTV depicted in 12) [11]. However, FOMTP and

FOMTV do not simultaneously account for ΔVtune and Pout

to be beneficial to VCO comparison. A new ΔVtune-added

and Pout-added FOM factor, called FOMTVP, is hence

proposed as 13).

oscTV

tune

VCO

1FOM PN 20log

10%

10log1mW

f TR

f V

P

(12)

oscTVP

tune

VCOout

1FOM PN 20log

10%

10log1mW

f TR

f V

PP

(13)

Table II shows the comparison of the performance of

the designed VCO in this paper with those of previously

reported VCOs in the Ku and X band. As can be seen

from Table I, The VCO reported in this work has an

excellent FOMTVP comparing with the other oscillators

processed in SiGe BiCMOS, GaAs PHEMT or CMOS

technology.

TABLE II: COMPARISON OF KU AND X VCOS

Ref. fosc

(GHz)

PN

(dBc/Hz)

△f

(MHz)

PVCO

(mW)

FOMTVP

(dB/Hz) Technology

[2] 22.7 -114 1 18 -150.9 SiGe HBT

[6] 32.55 -97 1 20 -140.6 SiGe

BiCMOS

[10] 24 -119.4 1 3.86 -165.3 CMOS

[12] 19 -112 1 200 -153.9 SiGe

BiCMOS

[13] 13.6 -116.2 1 3.78 -153.8 SiGe

BiCMOS

[14] 23.1 -94 1 2.5 -157.3 SiGe

BiCMOS

[15] 20.89 -97.2 1 40 -167.6 SiGe

BiCMOS

[16] 24.27 -100.3 1 7.8 -152.2 CMOS

[17] 19 -112 1 200 -140.5 SiGe

BiCMOS

[18] 31.6 -94 1 410 -130.3 GaAs

PHEMT

[19] 24 -119.4 1 3.86 -160.1 CMOS

[20] 24.3 -106.05 1 6.2 -158.3 CMOS

This

work 31.8 -114.6 1 1.8 -162.1

SiGe

BiCMOS

IV. CONCLUSION

A fully integrated 0.18 μm SiGe BiCMOS Ka-band

VCO, with cross-coupled pair configuration shows good

circuit performance, in terms of phase noise and low

power dissipation. This Ka-band VCO exhibits a -114.6

dBc/Hz phase noise at a 1 MHz offset and 3.31 GHz

tuning range. The FOM achieves -202.1 dBc/Hz.

ACKNOWLEDGMENT

This work was supported by the Advance Research

project of China (Grant No. 51307xxxx09), and the

Advance Research project of PLA in China

(KJ2013××××).



(

(9) FOM depicted

(

(

(

BAND

REFERENCES

[1] B. Hui, “Delay lock loop assisted phase lock loop for gnss

signal tracking,” Journal of Communications, vol. 11, no. 5,

pp. 471-477, May 2016.

[2] F. Padovan, M. Tiebout, K. L. R. Mertens, A. Bevilacqua,

and A. Neviani, “Design of low-noise K-band SiGe bipolar

VCOs: Theory and implementation,” IEEE Transactions

on Circuits and Systems-I: Regular Papers, vol. 62, no. 2,

pp. 607-615, 2015.

[3] H. Chang, C. C. Chan, I. Y. Shen, Y. L. Yeh, and S. Huang,

“Design and analysis of CMOS low-phase-noise low-jitter

subharmonically injection-locked VCO with FLL self-

alignment technique,” IEEE Transactions on Microwave

Theory and Techniques, vol. 64, no. 12, pp. 4632-4645,

Dec. 2016.

[4] D. Yoon, J. Yun, and J. S. Rieh, “A 310–340-GHz

coupled-line voltage-controlled oscillator based on 0.25-

μm InP HBT technology,” IEEE Transactions on

Terahertz Science and Technology, vol. 5, no. 4, pp. 652-

654, July 2015.

[5] J. Yun, N. Kim, D. Yoon, H. Kim, S. Jeon, et al., “A 248

262 GHz InP HBT VCO with interesting tuning behavior,”

IEEE Microwave and Wireless Components Letters, vol.

24, no. 8, pp. 560-562, Aug. 2014.

[6] M. Kucharski, F. Herzel, H. J. Ng, and D. Kissinger, “A

Ka-band BiCMOS LC-VCO with wide tuning range and

low phase noise using switched coupled inductors,”

presented at 2016 11th European Microwave Integrated

Circuits Conference (EuMIC), Germany, Oct. 3-4, 2016.

[7] S. Shopov, A. Balteanu, J. Hasch, P. Chevalier, A. Cathelin,

et al., “A 234–261-GHz 55-nm SiGe BiCMOS signal

source with 5.4–7.2 dBm output power, 1.3% DC-to-RF

efficiency, and 1-GHz divided-down output,” IEEE

Journal of Solid-State Circuits, vol. 51, no. 9, pp. 2054-

2065, Sep. 2016.

[8] S. Shopov, J. Hasch, P. Chevalier, A. Cathelin, and S. P.

Voinigescu, “A 240 GHz synthesizer in 55nm SiGe

BiCMOS,” presented at 2015 IEEE Compound

Semiconductor Integrated Circuit Symposium (CSICS),

Toronto, Oct. 11-14, 2015.

[9] T. Xi, S. Guo, P. Gui, D. Huang, Y. Fan, et al., “Low-

phase-noise 54-GHz transformer-coupled quadrature VCO

and 76-/90-GHz VCOs in 65-nm CMOS,” IEEE

Transactions on Microwave Theory and Techniques, vol.

64, no. 7, pp. 2091-2103, July 2016.

[10] M. Jalalifar and G. S. Byun, “A current-reused back-gate

coupling QVCO using transformer feedback structure,”


26, no. 7, pp. 534-536, July 2016.

[11] H. Koo, C. Y. Kim, and S. Hong, “A 254 GHz CMOS

transmitter with VCO-Q modulation,” IEEE Microwave

and Wireless Components Letters, vol. 26, no. 6, pp. 458-

460, June 2016.

[12] S. W. Kang, J. C. Chien, and A. M. Niknejad, “A W-band

low-noise PLL with a fundamental VCO in SiGe for

millimeter-wave applications,” IEEE Transactions on

Microwave Theory and Techniques, vol. 62, no. 10, pp.

2390-2404, Oct. 2014.

[13] X. M. Li, Z. Kang, and L. Jia, “A ku-band low-phase-noise

wide-tuning-range VCO in SiGe BiCMOS technology,”

Journal of Communications, vol. 11, no. 12, pp. 1102-1105,

Dec. 2016.

[14] Y. Huang, D. Wu, L. Zhou, F. Jiang, J. Wu, et al., “A 23

GHz low power VCO in SiGe BiCMOS technology,”

Journal of Semiconductors, vol. 34, no. 4, pp. 045003,

2013.

[15] J. He, J. Li, D. Hou, Y. Z. Xiong, and D. L. Yan, “A 20-

GHz VCO for PLL synthesizer in 0.13-μm BiCMOS,”

presented at the IEEE International Symposium on Radio-

Frequency Integration Technology, Singapore, Nov. 21-23,

2012.

[16] J. Yang, C. Y. Kim, D. W. Kim, and S. Hong, “Design of a

24-GHz CMOS VCO with an asymmetric-width

transformer,” IEEE Transactions on Circuits and

Systems—II: Express Briefs, vol. 57, no. 3, pp. 173-177,

March 2010.

[17] W. Wang, Y. Takeda, Y. S. Yeh, and B. Floyd, “A 20 GHz

VCO and frequency doubler for W-band FMCW radar

applications,” presented at the IEEE 14th Topical Meeting

on Silicon Monolithic Integrated Circuits in RF Systems

(SiRF), Kanagawa, Japan, Jan. 19-23, 2014.

[18] W. Lee, S. Lee, J. Choi, J. So, and Y. Kwon, “Ka-band

VCO with parasitic capacitance cancelling technique,”

Electronics Letters, vol. 53, no. 1, pp. 38-40, Jan. 2017.

[19] M. Jalalifar and G. S. Byun, “A current-reused back-gate

coupling QVCO using transformer feedback structure,”


26, no. 7, pp. 534-536, July 2016.

[20] P.

Y.

Wang,

Y.

C.

Chang,

K.

H.

Chuang,

D.

C.

Chang,

and

S.

S.

H.

Hsu, “A

low

phase-noise

24

GHz

CMOS

quadrature-VCO

using

PMOS

source-follower

coupling

technique,” presented

at

Proc.

Euro.

Microw.

Conf.,

Rome,

Italy, Oct.

6-9,

2014.

Yuhua Qi was born in Shishou, Hubei

province, China, in

1977. He received

the bachelor degree in Naval University

of Engineering, Wuhan, China, in

2000.

He received the master degree in Naval

University of Engineering, Wuhan,

China, in

2006. His research is focused

on information network technology,

modeling of HBTs and radio-frequency circuit design.

Rulong He was born in Hubei Province,

China, in 1977. He received the B.S.

degree from Naval University of

Engineering, PLA(NUEPLA), Wuhan,

in 2000 and the M.S. degree in 2006,

both in

communication engineering. His

research interests include submarine

cable communication, signal processing,

and information network theory.



http://ieeexplore.ieee.org/search/searchresult.jsp?searchWithin=%22Authors%22:.QT.Namhyung%20Kim.QT.&newsearch=true

http://ieeexplore.ieee.org/search/searchresult.jsp?searchWithin=%22Authors%22:.QT.Daekeun%20Yoon.QT.&newsearch=true

http://ieeexplore.ieee.org/search/searchresult.jsp?searchWithin=%22Authors%22:.QT.Hyunchul%20Kim.QT.&newsearch=true

http://ieeexplore.ieee.org/search/searchresult.jsp?searchWithin=%22Authors%22:.QT.Sanggeun%20Jeon.QT.&newsearch=true

http://ieeexplore.ieee.org/document/6824863/

http://ieeexplore.ieee.org/document/6824863/

http://ieeexplore.ieee.org/search/searchresult.jsp?searchWithin=%22Authors%22:.QT.Juergen%20Hasch.QT.&newsearch=true

http://ieeexplore.ieee.org/search/searchresult.jsp?searchWithin=%22Authors%22:.QT.Juergen%20Hasch.QT.&newsearch=true

http://ieeexplore.ieee.org/search/searchresult.jsp?searchWithin=%22Authors%22:.QT.Andreia%20Cathelin.QT.&newsearch=true

Zhao Shen was born in Hubei Province,

China, in 1980. He received the B.S.

degree from Naval University of

Engineering, PLA(NUEPLA), Wuhan,

in 2002 and the M.S. degree in 2005,

both in communication engineering. He

is currently pursuing the Ph.D. degree

with the School of Electronic

Information and Communications, Huazhong University of

Science and Technology(HUST), China. His research interests

include wireless communication, signal processing, and

information theory.



A 32 GHz Low-Power Low-Phase-Noise VCO … in SiGe BiCMOS Technology . ... the low-phase-noise oscillator core make it ... compensating the losses of the inductor represented by resistor

Documents