2012_60_02_P1

FEBRUARY 2012

VOLUME 60

NUMBER 2

IETPAK

(ISSN 0018-926X)

PART I OF TWO PARTS

SPECIAL ISSUE ON MULTIPLE-INPUT MULTIPLE-OUTPUT (MIMO) TECHNOLOGY

Guest Editorial . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. W. Wallace, J. B. Andersen, B. K. Lau, B. Daneshrad, and J. Takada

434

Antenna Design, Modeling, and Analysis Design of a MIMO Dielectric Resonator Antenna for LTE Femtocell Base Stations . . . . . . . . J.-B. Yan and J. T. Bernhard A Compact Eighteen-Port Antenna Cube for MIMO Systems . . . . . . . . . . . . . . . . . . . J. Zheng, X. Gao, Z. Zhang, and Z. Feng Printed MIMO-Antenna System Using Neutralization-Line Technique for Wireless USB-Dongle Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S.-W. Su, C.-T. Lee, and F.-S. Chang Simple and Efcient Decoupling of Compact Arrays With Parasitic Scatterers . . . . . . . . . . . . . B. K. Lau and J. B. Andersen Reducing Mutual Coupling of MIMO Antennas With Parasitic Elements for Mobile Terminals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Z. Li, Z. Du, M. Takahashi, K. Saito, and K. Ito A Compact Wideband MIMO Antenna With Two Novel Bent Slits . . . . . . . . .. . . . . . . . J.-F. Li, Q.-X. Chu, and T.-G. Huang Characteristic Mode Based Tradeoff Analysis of Antenna-Chassis Interactions for Multiple Antenna Terminals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . H. Li, Y. Tan, B. K. Lau, Z. Ying, and S. He Multiple Antenna Systems With Inherently Decoupled Radiators . . . . . . . . . M. Pelosi, M. B. Knudsen, and G. F. Pedersen A Pattern Recongurable U-Slot Antenna and Its Applications in MIMO Systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P.-Y. Qin, Y. J. Guo, A. R. Weily, and C.-H. Liang Multiple Element Antenna Efciency and its Impact on Diversity and Capacity . . . . . . . . . . . . . J. X. Yun and R. G. Vaughan On the Accuracy of Equivalent Circuit Models for Multi-Antenna Systems . . . . . . . . . . . . . . J. W. Wallace and R. Mehmood Channel Sounding and Modeling A Low-Cost MIMO Channel Sounder Architecture Without Phase Synchronization . .. . D. Pinchera and M. D. Migliore Impact of Incomplete and Inaccurate Data Models on High Resolution Parameter Estimation in Multidimensional Channel Sounding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Landmann, M. Kske, and R. S. Thom A General Coupling-Based Model Framework for Wideband MIMO Channels . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Y. Zhang, O. Edfors, P. Hammarberg, T. Hult, X. Chen, S. Zhou, L. Xiao, and J. Wang

438 445 456 464 473 482 490 503 516 529 540 548 557 574

(Contents Continued on p. 433)

(Contents Continued from Front Cover) Multi-Link MIMO Channel Modeling Using Geometry-Based Approach . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. Poutanen, F. Tufvesson, K. Haneda, V.-M. Kolmonen, and P. Vainikainen Land Mobile Satellite Dual Polarized MIMO Channel Along Roadside Trees: Modeling and Performance Evaluation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Cheffena, F. P. Fontn, F. Lacoste, E. Corbel, H.-J. Mametsa, and G. Carrie Empirical-Stochastic LMS-MIMO Channel Model Implementation and Validation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P. R. King, T. W. C. Brown, A. Kyrgiazos, and B. G. Evans System Performance Evaluation Effectiveness of Relay MIMO Transmission by Measured Outdoor Channel State Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . K. Nishimori, N. Honma, T. Murakami, and T. Hiraguri Single and Multi-User Cooperative MIMO in a Measured Urban Macrocellular Environment . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . B. K. Lau, M. A. Jensen, J. Medbo, and J. Furuskog User Inuence on MIMO Channel Capacity for Handsets in Data Mode Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. . Nielsen, B. Yanakiev, I. B. Bonev, M. Christensen, and G. F. Pedersen Exposure Compliance Methodologies for Multiple Input Multiple Output (MIMO) Enabled Networks and Terminals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . N. Perentos, S. Iskra, A. Faraone, R. J. McKenzie, G. Bit-Babik, and V. Anderson MIMO Transmission Using a Single RF Source: Theory and Antenna Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . O. N. Alrabadi, J. Perruisseau-Carrier, and A. Kalis MIMO Capacity Enhancement Using Parasitic Recongurable Aperture Antennas (RECAPs) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . R. Mehmood and J. W. Wallace Eigen-Coherence and Link Performance of Closed-Loop 4G Wireless in Measured Outdoor MIMO Channels . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . M. Webb, M. Hunukumbure, and M. Beach Multipath Simulator Measurements of Handset Dual Antenna Performance With Limited Number of Signal Paths . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P. Hallbjrner, J. D. Snchez-Heredia, P. Lindberg, A. M. Martnez-Gonzlez, and T. Bolin On Small Terminal Antenna Correlation and Impact on MIMO Channel Capacity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . B. Yanakiev, J. . Nielsen, M. Christensen, and G. F. Pedersen Compensating for Non-Linear Ampliers in MIMO Communications Systems . . . . . . . . . S. A. Banani and R. G. Vaughan

587 597 606

615 624 633 644 654 665 674 682 689 700

CALL FOR PAPERS

Call for Papers: Special Issue on Antennas and Propagation at Millimeter and Sub-millimeter Waves . . . . . . . . . .. . . . . . . . . .

715

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Digital Object Identier 10.1109/TAP.2012.2186030

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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 2, FEBRUARY 2012

Guest Editorial for the Special Issue on Multiple-Input Multiple-Output (MIMO)

W

E are pleased to present this special issue on multiple-input multiple-output (MIMO), which represents a breakthrough in the use of antenna arrays in wireless transmission. Unlike traditional phased array or diversity techniques that enhance one signal of interest, MIMO systems employ antenna arrays jointly at the transmitter and receiver to spatially multiplex signals, providing tremendous capacity gains. Although there has already been intense research in MIMO wireless communications, and many obstacles in signal processing, modulation, and coding for MIMO systems have been overcome, outstanding questions in the areas of antennas and propagation remain, making MIMO a timely topic for our community. The need for research in this area becomes even more apparent as new standards such as IEEE 802.11n, LTE Advanced, and WiMAX that include MIMO operation are implemented, revealing that physical devices, antennas, and channels can no longer be oversimplied or neglected. This special issue is organized into three main sections: 1) antenna design, modeling, and analysis, 2) channel sounding and modeling, and 3) system performance evaluation. A. Antenna Design, Modeling, and Analysis

Although signal processing treatments of MIMO may treat antennas as isotropic elements that are not affected by nearby antennas or scatterers, real antennas exhibit non-isotropic patterns and inter-element coupling. This section contains papers that consider the challenges of designing compact MIMO antennas with good performance, as well as novel and rigorous ways to model and analyze such antenna systems. Exploiting multiple polarizations is a possible method of achieving a compact MIMO design with low coupling. Yan and Bernhard present a clever design allowing two orthogonal resonant modes of a compact dielectric resonator antenna (DRA) for LTE700 femtocell applications, achieving polarization and angle diversities and 30 dB isolation. A low prole tri-polarized antenna consisting of a dual-polarized ring patch and a disk-loaded monopole is explored in Zheng et al. to build an 18-port antenna cube, exhibiting lower mutual coupling and simpler feeding than a dipole MIMO cube. Several papers address the challenge of MIMO antenna design for compact user terminals exhibiting higher mutual coupling and correlation. Su et al. implement a printed neutralization line along one ground plane edge to decouple a twomonopole array for a USB dongle application at 2.4 GHz, requiring little modication of the ground plane. The use of parasitic structures for coupling mitigation is explored in several contributions. Lau and Bach Andersen introduce the theory of parasitic decoupling, whereby two arbitrary antennas of a given antenna spacing can be perfectly decoupled with a reactively loaded parasitic element acting as a reector. Experiments reveal that decoupling is achieved with only a small penalty in total efciency. Z. Li et al. introduce a complementary perspective that the parasitic elements create a second path for couDigital Object Identier 10.1109/TAP.2012.2183909

pling cancellation, demonstrating the principle by decoupling two closely-coupled slot antennas using two monopoles as parasitic elements. J. Li et al. design an efcient wideband MIMO antenna by combining a parasitic decoupling strip with right-angled slits in the ground plane to obtain 2.4 GHz6.55 GHz operation and 18 dB isolation. The ground plane of compact user terminals can play a major role in the radiation of MIMO antennas at low frequency where the chassis is excited. The theory of characteristic mode is explored by H. Li et al. in the context of designing efcient MIMO antennas by placing the elements to avoid simultaneous excitation of the chassis by more than one antenna element. A tradeoff analysis shows that MIMO performance is signicantly improved by the increased isolation. Pelosi et al. carry out a comprehensive study on the performance of small narrowband antennas with and without a user in either MIMO mode or transceiver separation mode (TSM). This approach can relax the duplex lter requirement in TSM, although user effects may largely inuence the antenna performance. In order to further improve MIMO antenna performance in a time-varying propagation channel, recongurable antenna elements may be employed to optimize the antenna-channel interaction. Qin et al. show that two pattern recongurable U-slot antenna elements can provide capacity gain in measured line-ofsight (LOS) and non-LOS channels, relative to two omnidirectional reference antennas. Metrics and models for MIMO antennas are considered by two contributions. Yun and Vaughan isolate the role of antenna efciencies from correlation in the diversity and capacity performance of a given MIMO antenna. Thereafter, the MIMO antenna can be represented with an equivalent number of ideal antenna branches that are called diversity order and capacity order, respectively. The question of the validity and accuracy of equivalent circuit models for MIMO arrays is addressed by Wallace and Mehmood, where a method-of-moments analysis based on rst principles reveals that such models are exact under normal circumstances, and that transmit and receive modes can be analyzed with a single unied model. B. Channel Sounding and Modeling This section focuses on accurate channel characterization through sounding and modeling, which is vital to correctly assess the benets of MIMO transmission, allowing critical tradeoffs and design decisions to be made. Two papers in this section directly consider the topic of channel sounding. Pinchera and Migliore present an interesting measurement approach using a parasitic array instead of a switched array. Using low cost switched parasitic elements instead of a large multiport microwave switch dramatically reduces the cost of MIMO channel sounding with only modest reduction in accuracy. The impact of an imperfect underlying model on the accuracy of high-resolution double-directional MIMO channel estimation is studied by Landmann et al. It is shown that modeling this uncertainty allows multipath to be correctly classied as discrete or diffuse, and that imperfect calibration can lead to large error in multipath estimates.

0018-926X/$31.00 2012 IEEE


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Sounder-based channel modeling is considered in two papers. Zhang et al. extend tensor-based MIMO modeling approaches to the case of wide bandwidth, which is required for todays wireless standards. The model is assessed using measured indoor channels, indicating a tradeoff between complexity and accuracy when generating synthetic MIMO channel data. Poutanen et al. propose a method for extending geometry-based stochastic channel models to the case of multiple links, which is important to analyze MIMO systems using coordinated transmission or relays. This model is accomplished by having certain clusters that are shared by the links, creating dependence in the statistics of the MIMO channels. Finally, this section includes two papers that present measurement of land mobile satellite (LMS) channels. Cheffena et al. consider the effect of signal shadowing by trees in MIMO-LMS links, proposing a multipath model for trees based on multiple scattering theory. The model is compared with direct FDTD simulation, indicating that good accuracy can be obtained with modest complexity. King et al. investigate the use of multiple antennas to increase the capacity of LMS networks, where a Markov chain is employed to characterize the time-variant nature of shadowing and depolarization effects. The utility of the proposed technique is illustrated through direct measurements with an articial LMS platform. C. System Performance Evaluation The nal section deals principally with system-level aspects, indicating how detailed characteristics of the propagation channel, antennas, and devices affect the performance of the overall MIMO system or network. Two papers consider the emerging topic of relays and coordinated MIMO transmission. Nishimori et al. evaluate the capacity of relay-enhanced multi-antenna transmission in a cellular environment through direct propagation measurements taken in Yokkaichi City, Japan. This study shows that characterizing path-loss differences is critical and that relay-enhanced MIMO can provide a 50% improvement in capacity. Lau et al. analyze urban propagation measurements involving three coherent base stations and a mobile unit equipped with four antennas. Capacity for cooperative transmission from the base stations is analyzed, revealing dramatic sum-rate capacity gains compared to non-cooperative methods. User inuence and exposure limits are considered in two contributed papers. Nielsen et al. provide a detailed study of user inuence on the outage capacity for mobile devices in the data mode operation. Six different handsets at two bands are characterized for twelve different users, showing that handset design and hand position critically impact body loss, mean effective gain, and outage capacity. Perentos et al. consider compliance and exposure testing of MIMO devices, which is important as multi-antenna technology is increasingly incorporated into advanced devices. The developed methodologies allow such testing to be performed with scalar eld probes, avoiding expensive upgrades of existing test equipment. The use of parasitic arrays for MIMO transmission are considered in two papers, providing reduced complexity or capacity enhancement compared to classical MIMO systems. Alrabadi et al. develop the methodology of using a switched parasitic array (SPA) with only a single active RF source to replace a full MIMO transmitter with reduced cost and complexity. The generalized method for forming the required orthogonal bases is demonstrated through simulation and direct measurement with

a prototype SPA. Mehmood and Wallace propose exible recongurable aperture (RECAP) antennas to increase MIMO capacity in interference-limited scenarios. Multi-user simulations with a detailed noise model suggest that high recongurability can lead to many-fold capacity increase. Finally, four papers are included that extend or verify assumptions made in existing modeling approaches for MIMO systems. Webb et al. consider the coherence time and bandwidth of channel state information in measured time-varying urban channels, indicating how sensitive feedback methods are to time and frequency offsets. The study shows that controlling the feedback rate can lead to signicant improvements in mobile MIMO systems. Hallbjrner et al. explore the impact of sparse multipath on antenna correlation and diversity, in contrast to classical treatments where innite and uniform arrivals are assumed. Multipath channels are simulated using antenna arrays in an anechoic chamber, showing that sparse multipath can lead to high variability or spread of channel statistics like correlation. Yanakiev et al. study the use of correlation as a metric in the design stage to predict handset performance in terms of MIMO capacity in real scenarios. The surprising result is that correlation may have little bearing on capacity, indicating correlation may be a misleading gure of merit. Finally, Banani and Vaughan investigate the effect of non-linear ampliers in practical MIMO systems and how to compensate the resulting degradations to channel capacity. A model for non-linear MIMO systems is introduced, and a blind channel-estimation technique is developed to estimate and track the channel in the presence of non-linearities. To conclude this guest editorial, we would like to thank the former Editor-in-Chief Dr. Trevor S. Bird and his successor Prof. Michael A. Jensen, for providing us with the opportunity to coordinate and organize this special issue and for their continued support throughout the process. We are also grateful to the many anonymous reviewers who helped make the special issue possible. We believe that the issue provides a true snapshot of the state-of-the-art in antennas and propagation research in MIMO systems, serving as interesting reading as well as a useful reference for years to come. JON W. WALLACE, Guest Editor School of Engineering and Science Jacobs University Bremen, Germany JRGEN BACH ANDERSEN, Guest Editor Department of Electronic Systems Aalborg University Aalborg, Denmark BUON KIONG LAU, Guest Editor Department of Electrical and Information Technology Lund University Lund, Sweden BABAK DANESHRAD, Guest Editor Department of Electrical Engineering University of California, Los Angeles Los Angeles, CA JUN-ICHI TAKADA, Guest Editor Graduate School of Engineering Tokyo Institute of Technology Tokyo, Japan

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ON

ANTENNAS AND

Jon W. Wallace (S99M03) received the B.S. (summa cum laude) and Ph.D. degrees in electrical engineering from Brigham Young University (BYU), Provo, UT, in 1997 and 2002, respectively. From 1995 to 1997, he worked as an Associate of Novell, Inc., Provo, and during 1997 he was a Member of Technical Staff for Lucent Technologies, Denver, CO. He received the National Science Foundation Graduate Fellowship in 1998 and worked as a Graduate Research Assistant at BYU until 2002. From 2002 to 2003, he was with the Mobile Communications Group, Vienna University of Technology, Vienna, Austria. From 2003 to 2006, he was a Research Associate with the BYU Wireless Communications Laboratory. Since 2006, he has been Assistant Professor of electrical engineering at Jacobs University, Bremen, Germany. His current research interests include wireless channel sounding and modeling, physical-layer security, MIMO communications, cognitive radio, and ultrawideband (UWB) systems. Dr. Wallace currently serves as an Associate Editor of the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION. He was awarded the H. A. Wheeler paper award in the IEEE TRANSACTIONS PROPAGATION in 2002.

Jrgen Bach Andersen (M68SM78F92LF02) received the M.Sc. and Dr.Techn. degrees from the Technical University of Denmark (DTU), Lyngby, Denmark, in 1961 and 1971, respectively. In 2003 he was awarded an honorary degree from Lund University, Sweden. From 1961 to 1973, he was with the Electromagnetics Institute, DTU and since 1973 he has been with Aalborg University, Aalborg, Denmark, where he is now a Professor Emeritus and Consultant. He was head of a research center, Center for Personal Communications, CPK, from 19932003. He has been a Visiting Professor in Tucson, AZ; Christchurch, New Zealand; Vienna, Austria; and Lund, Sweden. He has published widely on antennas, radio wave propagation, and communications, and has also worked on biological effects of electromagnetic systems. He has coauthored a book, Channels, Propagation and Antennas for Mobile Communications (IEE, 2003). He was on the management committee for COST 231 and 259, a collaborative European program on mobile communications. Prof. Andersen is a former Vice President of the International Union of Radio Science (URSI) from which he was awarded the John Howard Dellinger Gold Medal in 2005.

Babak Daneshrad received the B.Eng. and M.Eng. degrees with emphasis in communications from McGill University, Montreal, Quebec, Canada, in 1986 and 1988, respectively, and the Ph.D. degree with emphasis in integrated circuits and systems from the University of California, Los Angeles (UCLA), in 1993. In January 2001, he co-founded Innovics Wireless, a company focused on developing 3G cellular mobile terminal antenna diversity solutions and in 2004 he co-founded Silvus Communications. From 1993 to 1996, he was a member of technical staff with the Wireless Communications Systems Research Department, AT&T Bell Laboratories, where he was involved in the design and implementation of systems for high-speed wireless packet communications. Currently, he is a Professor with the Electrical Engineering Department, UCLA. His research interests are in the areas of wireless communication system design, experimental wireless systems, and VLSI for communications. His current research interests are cross disciplinary in nature and deal with addressing practical issues associated with the realization of advanced wireless systems. The work is focused on low power MIMO wireless systems, as well as cognitive radio communications. Prof. Daneshrad is the recipient of the 2005 Okawa Foundation award, a coauthor of the best paper award at PADS 2004, and was awarded rst prize in the DAC 2003 design contest. He is the beneciary of the endowment for UCLA-Industry Partnership for Wireless Communications and Integrated Systems.


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Buon Kiong Lau (S00M03SM07) received the B.E. degree (with honors) from the University of Western Australia, Perth, Australia and the Ph.D. degree from Curtin University of Technology, Perth, in 1998 and 2003, respectively, both in electrical engineering. During 2000 to 2001, he worked as a Research Engineer with Ericsson Research, Kista, Sweden. From 2003 to 2004, he was a Guest Research Fellow at the Department of Signal Processing, Blekinge Institute of Technology, Sweden. Since 2004, he has been at the Department of Electrical and Information Technology, Lund University, where he is now an Associate Professor. He has been a Visiting Researcher at the Department of Applied Mathematics, Hong Kong Polytechnic University, China, Laboratory for Information and Decision Systems, Massachusetts Institute of Technology, and Takada Laboratory, Tokyo Institute of Technology, Japan. His primary research interests are in various aspects of multiple antenna systems, particularly the interplay between antennas, propagation channels and signal processing. Dr. Lau is an Associate Editor for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION. From 2007 to 2010, he was a Co-Chair of Subworking Group 2.2 on Compact Antenna Systems for Terminals (CAST) within EU COST Action 2100. Since 2011, he is a Swedish national delegate and the Chair of Subworking Group 1.1 on Antenna System Aspects within COST IC1004.

Jun-ichi Takada (SM11) received B.E. and D.E. degrees from Tokyo Institute of Technology (Tokyo Tech), Japan, in 1987 and 1992, respectively. He was a Research Associate at Chiba University from 1992 to 1994, and an Associate Professor at Tokyo Tech from 1994 to 2006 where he has been a Professor since 2006. From 2003 to 2007, he was also a Researcher at the National Institute of Information and Communications Technology (NICT), Japan. His current interests include the radiowave propagation and channel modeling for various wireless systems, and regulatory issues of spectrum sharing.

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Design of a MIMO Dielectric Resonator Antenna for LTE Femtocell Base StationsJie-Bang Yan, Member, IEEE, and Jennifer T. Bernhard, Fellow, IEEEAbstractA novel multiple-input multiple-output (MIMO) dielectric resonator antenna (DRA) for long term evolution (LTE) femtocell base stations is described. The proposed antenna is able to transmit and receive information independently using TE and HE modes in the LTE bands 12 (698716 MHz, 728746 MHz) and 17 (704716 MHz, 734746 MHz). A systematic design method based on perturbation theory is proposed to induce mode degeneration for MIMO operation. Through perturbing the boundary of the DRA, the amount of energy stored by a specic mode is changed as well as the resonant frequency of that mode. Hence, by introducing an adequate boundary perturbation, the TE and HE modes of the DRA will resonate at the same frequency and share a common impedance bandwidth. The simulated mutual coupling between the modes was as low as . It was estimated that in a rich scattering environment with an Signal-to-Noise Ratio (SNR) of 20 dB per receiver branch, the proposed MIMO DRA was able to achieve a channel capacity of 11.1 b/s/Hz (as compared to theoretical maximum 2 2 capacity of 13.4 b/s/Hz). Our experimental measurements successfully demonstrated the design methodology proposed in this work. Index TermsDielectric resonator antenna (DRA), long term evolution (LTE), multiple-input multiple-output (MIMO) antenna, mutual coupling, perturbation method.

I. INTRODUCTION

T

HE Federal Communications Commission (FCC) recently released the 700 MHz spectrum which was previously used for analog television broadcasting [1]. A new nationwide wireless broadband network based on long term evolution (LTE) technology has been proposed to operate in the 700 MHz spectrum [2], [3]. In the LTE Evolved UMTS terrestrial radio access (E-UTRA) air interface, multiple-input multiple-output (MIMO) technology plays an important role in increasing the systems spectral efciency [4], [5]. Given the lower operating frequency of the LTE system, as compared to the WiFi and cellular standards, the antenna in handheld devices such as a smartphone or a netbook must be electrically

Manuscript received May 27, 2010; revised December 14, 2010; accepted February 05, 2011. Date of publication October 28, 2011; date of current version February 03, 2012. This work was supported by the Motorola Center for Communications at the University of Illinois at Urbana-Champaign and a Croucher Foundation Scholarship. J.-B. Yan was with the Electromagnetics Laboratory, Department of Electrical and Computer Engineering, University of Illinois at Urbana-Champaign, Urbana, IL 61801 USA. He is now with the Center for Remote Sensing of Ice Sheets (CReSIS), University of Kansas, Lawrence, KS 66045 USA. J. T. Bernhard is with the Electromagnetics Laboratory, Department of Electrical and Computer Engineering, University of Illinois at Urbana-Champaign, Urbana, IL 61801 USA (e-mail: [email protected]). Color versions of one or more of the gures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identier 10.1109/TAP.2011.2174021

small. This implies the mobile antennas are likely to be inefcient and the coverage of the system is therefore limited. This is especially true if MIMO operation is needed at both mobile and base station since the antenna efciency would be further reduced due to strong mutual coupling between closely-packed mobile antennas. In view of this, LTE architecture includes a femtocell solution for coverage extension [6]. Femtocells can be considered as low-power access points serving indoor areas. To exploit the richness in multipath propagation in indoor scenarios, it is desired to employ MIMO antennas with a very low mutual coupling as the base station antenna in a femtocell. One possible solution would be the orthogonally polarized MIMO antennas proposed in [7]. However the problem is that such antennas would be oversized when scaled to operate at 700 MHz. Hence, a new MIMO antenna solution for LTEs femtocell base station is necessary. In this work, a 700 MHz dual-mode MIMO dielectric resonator antenna (DRA) that is suitable for the new wireless system is proposed. Although the cost of DRAs may be high as compared to traditional PIFAs or microstrip antennas, they have the advantages of compact size, high radiation efciency, and wide impedance bandwidth [8]. Another important feature of DRAs is that the three dimensional structure offers more degrees of freedom in exciting various orthogonal resonant modes, and each mode can be utilized to transmit and receive information independently. This makes the DRA an ideal candidate for application in MIMO communication systems. Indeed, a multi-mode usage of a single dielectric resonator has been suggested in [9], but the emphasis is not on MIMO applications. The concept of a MIMO DRA was rst described and demonstrated by Ishimiya et al. in [10], [11]. It was experimentally shown that a cubic MIMO DRA is able to achieve a diversity gain of about 10 dB and has comparable performance to traditional MIMO dipole arrays in practical IEEE 802.11n systems. Nevertheless, in Ishimiyas papers, no explicit design method has been described. The major difculty of applying DRAs in MIMO systems is to make various modes to resonate at the same frequency while maintaining low coupling between the modes. Here, we introduce a systematic design method for MIMO DRAs. The key in MIMO DRA design is to induce degenerate modes (i.e., modes that have the same resonant frequency). Conventionally, only DRAs that exhibit symmetry can support degenerate modes [12] and this limits any further size reduction of MIMO DRAs. Hence, a novel mode degeneration method based on boundary perturbation is proposed and demonstrated in this work. Section II describes the base design for the proposed MIMO DRA, then, Section III introduces the boundary perturbation for

0018-926X/$26.00 2011 IEEE

YAN AND BERNHARD: DESIGN OF A MIMO DIELECTRIC RESONATOR ANTENNA FOR LTE FEMTOCELL BASE STATIONS

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Fig. 1. Perspective view of the split-cylindrical DRA ( , , , and

, ).

mode degeneration. In Section IV, we evaluate the performance of the perturbed antenna structure. Simulated results including those for MIMO capacity are provided. Following that, some experimental results are given in Section V as a validation to the developed design methodology. Finally a conclusion and a discussion of future work are given in Section VI. II. BASE DESIGN Consider a split-cylindrical DRA , with a radius of 44 mm and a length of 80 mm residing on a ground plane with dimensions as shown in Fig. 1. The mode and the mode can be excited simultaneously using appropriate excitation methods, such as probe feeds, aperture coupling or microstrip feeds. The value of the subscript ranges between zero and one, depending on the method of feeding [12]. Here, a 50 microstrip-fed rectangular slot and a probe feed were chosen to excite the and modes, respectively (see Fig. 1). FR-4 epoxy board with thickness of 1.6 mm is used as the substrate of the microstrip line. The dimensions of the slot are 50 mm 4 mm and the probe that excites the mode has a length of 27 mm. Fig. 2 shows the plots of the theoretical magnetic eld distributions for the two modes inside the DRA computed using Wolfram Research Mathematica [13]. It can be seen that mode behaves as a magnetic dipole on the -axis while mode radiates as a short magnetic dipole oriented along the -axis. The two modes are therefore orthogonal to each other and should exhibit low mutual coupling. The resonant frequencies of the mode and mode can be derived from the separation equation [8] and are found to be 653 MHz and 520 MHz, respectively,Fig. 2. Theoretical magnetic eld distributions for the (a) mode. (b) mode, and

Fig. 3. Simulated -parameters of the unperturbed cylindrical DRA.

(1) (2) where is the speed of light in free space, and and are the rst zeros of the zero-order Bessel function and the derivative of the rst-order Bessel function, respectively. A full-wave

simulation was performed using Ansys HFSS [14] and the simulated -parameters of the antenna are shown in Fig. 3. The theoretically predicted and simulated operating frequencies of the modes agree very well with each other. It can also be seen that the coupling between the two modes is very low as expected. III. DESIGN OF MIMO DRA A. Boundary Perturbation In order to work in a MIMO system, the two modes should have the same resonant frequency and have a shared impedance bandwidth. To accomplish this, we propose a mode degeneration method based on boundary perturbation. For an arbitrarily

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shaped dielectric resonator, the change in resonant frequency due to a change of the cavity wall can be determined using perturbation theory [15], and is given by,

(3)

Fig. 4. Boundary perturbation from the base of the split-cylindrical DRA (Cross-sectional ( -plane) view).

where and are the permittivity and the permeability of the dielectric resonator respectively, and are the resonant radian frequencies of the perturbed and unperturbed resonator, respectively, and are the volume perturbed and the original volume of the resonator, and and are the unperturbed elds. Equation (3) indicates that the change in resonant frequency is equal to the electric and magnetic energies removed by the perturbation divided by the total energy stored [15], i.e.,

(4) where and are time-averaged electric and magnetic energies originally contained in the volume perturbed and is the total energy stored in the unperturbed cavity. Now consider a boundary perturbation from the base of the split-cylindrical DRA as depicted in Fig. 4. The changes in resonant frequencies of the mode and mode can be computed using (3), and the result is shown in Fig. 5. It can be observed that as the electric boundary is moved up, the resonant frequency of the mode increases more rapidly than that of the mode. Hence, at a certain perturbation value, the two resonant frequencies should overlap and thus fulll the primary requirement for MIMO antenna design. According to (4), the difference in the rate of change of resonant frequency can be explained by the difference in the energy stored by the two modes in the perturbation volume . To verify the boundary perturbation method, an HFSS simulation was carried out and the result is also shown in Fig. 5. It can be seen that the result predicted by the boundary perturbation method starts to deviate from the result obtained from HFSS when the perturbation, , increases. This is due to the substitution of the original elds into the perturbed elds during the derivation of (3). The difference between the original elds and the perturbed elds would be intolerable when the perturbation is too large. Thus, the deviation at large perturbations is inherent in the perturbation analysis. Nonetheless, the boundary perturbation method gives a good initial guess on how much perturbation is required to make the two modes resonate at the same frequency. According to the HFSS simulation result, the two modes both resonate at 700 MHz when the perturbation, , is 13 mm. In (3), there is no specic constraint on the geometry of the cavity, hence, the proposed boundary perturbation method can be applied to DRAs of any other shapes with arbitrary perturbations. However, the difculty of analysis of such structures might be the evaluation of the integrals in (3).

Fig. 5. Plot of change of resonant frequency eter .

against the perturbation param-

Fig. 6. Elliptical approximation of the perturbed cylindrical boundary (Crosssectional ( -plane) view).

Fig. 7. Simulated -parameters of the perturbed cylindrical DRA.


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Fig. 8. Comparison of the simulated and measured radiation patterns of Port 1 (HE mode).

Fig. 9. Comparison of the simulated and measured radiation patterns of Port 2 (TE mode).

B. Boundary Approximation While the above described boundary perturbation method obtains the solutions by approximating the elds, boundary approximation estimates the change in resonant frequency by approximating the structure of the resonator. According to Fig. 6, a perturbed cylindrical boundary can be modeled by a half ellipse. The accuracy of this method depends on how well the perturbed circular arc is approximated by an elliptic arc. The resonant frequencies of various modes in an elliptical DRA can be found by expanding the elds inside the cavity in Mathieu functions and applying the technique of separation of variables. A detailed analysis of an elliptical DRA can be found in [16]. The resonant frequencies of a series of split-elliptical DRAs of various minor axes, which corresponded to the previously described set of perturbed cylindrical DRAs, are computed, and the change in resonant frequency estimated by this boundary approximation method is plotted in Fig. 5. The results obtained agree very well with those calculated by both the boundary perturbation method and full-wave simulations. IV. SIMULATED ANTENNA PERFORMANCE A. Antenna Characteristics mode and the mode From Section III-A, the of a split-cylindrical DRA will both resonate at 700 MHz when

the perturbation, , is 13 mm. The dimensions of the perturbed DRA are 80 mm 84 mm 31 mm. Given the same resonant frequency and a half-wavelength antenna separation, the dimensions of a two-element MIMO PIFA would be 107 mm 214 mm 5 mm. Since coupling between antenna ports is another important parameter to characterize MIMO antennas, the proposed antenna structure was simulated in HFSS. The simulated -parameters of the antenna were obtained with 50 terminations at both ports and are given in Fig. 7. It can be observed that the mutual coupling between the two modes is insensitive to the perturbation (see Fig. 3) and is less than . This is signicantly lower than the mutual coupling in conventional MIMO antennas that are based on dipole antennas, patch antennas or PIFAs [17][20]. The impedance bandwidths (dened as ) of the mode and the mode are 10 MHz and 35 MHz respectively. The mode has a relatively narrow bandwidth and limits the overall bandwidth of the antenna. Nevertheless, the bandwidth of the mode can be improved using well known bandwidth broadening techniques, such as inserting an air gap between the ground plane and the DRA [21], [22], or adding a matching stub at the end of the microstrip line [23]. The simulated gains of the and modes are 3.96 and 3.19 dBi, respectively. The simulated radiation patterns of the two modes, which are orthogonal to each other, are given in Figs. 8 and 9. Hence, the antenna is

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Fig. 10. The oor plan (16 m 19 m) of the ofce environment that was used to estimate the MIMO channel capacity (notice that there is no line-of-sight (LOS) path between the transmitter and any of the receivers). Fig. 11. Estimated channel capacity of the proposed MIMO DRA.

able to exploit polarization diversity and the pattern orthogonality leads to low mutual coupling between the ports. B. MIMO Performance Evaluation The channel capacity gain by using the proposed antenna was evaluated with the aid of Remcom Wireless Insite [24]. In the simulation setup, a single transmitter and 1000 identical receivers were placed in an ofce environment as shown in Fig. 10. The ofce environment was constructed to resemble a rich scattering environment (i.e., the channel statistics are approximately Rayleigh distributed). In order to resemble a timevarying MIMO channel, the receivers were randomly spread across the designated area of the ofce such that a 1000 nonline-of-sight (NLOS) communication links were established. In all the simulations, there were 80 paths for each channel realization. The simulated complex radiation patterns (including both polarizations, and ) of the proposed antennas were used at the transmitter and all the receivers. 1000 samples of the unnormalized channel matrix were then obtained from the simulation [25], the transmit antennas, the at-fading MIMO channel capacity of the -th link, , can be calculated by [25][27]

(7) where and denote the number of transmit and receiver antennas, respectively; is an identity matrix with dimension ; is the mean signal-to-noise ratio (SNR) per receive branch; represents a complex conjugate transpose; and is the -th normalized channel matrix

(8)

(5) communication links, and where there are is the unnormalized channel matrix of the -th link. Here, represents the -th sample of the complex channel gain between port of the transmitter and port of the receiver, where subscripts , 2 and , 2:

(6) Here is the number of path in the -th link; is the received power contributed by the -th path in the -th link; is the phase of the -th path in the -th link. From the simulated channel data, it was found that the coherence bandwidth of the wireless channel was much larger than the bandwidth of the proposed antenna. Hence, for equal power distributed among

where denotes a Frobenius norm. The mean capacity and the maximum achievable capacity obtained by using the proposed antenna are plotted in Fig. 11. Fig. 11 also gives the theoretical channel capacities for single-input single-output (SISO), 2 2 and 3 3 channels with zero mean, unity variance, i.i.d. complex Gaussian distributed channel elements for comparison. The results indicate that the estimated mean channel capacity is 11.1 b/s/Hz at an SNR of 20 dB per receiver branch. The maximum achievable capacity is very close to the theoretical maximum 2 2 MIMO capacity of 13.4 b/s/Hz. The small discrepancy between the theoretical and simulated capacities may be due to the non-ideal scattering environment and nite mutual coupling between the modes. Nevertheless, the simulation results reect the utility of the antenna design, and a prototype antenna is presented in Section V. V. MEASUREMENT RESULTS The perturbed cylindrical DRA was built and tested in the Electromagnetics Laboratory at University of Illinois at Urbana-Champaign. The dielectric material ( and ) was supplied by Countis Laboratories [28]. The dielectric block was bonded onto the ground plane


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REFERENCES[1] Federal Communications Commission, 700 MHz band, Auction 73 Feb. 2009. [2] News Archives AT&T Inc., 2008 [Online]. Available: http://www.att. com/ [3] News Archives Verizon Wireless, 2009 [Online]. Available: http://news.vzw.com/ [4] 3GPP TS36.300, Evolved Universal Terrestrial Radio Access (E-UTRA) and Evolved Universal Terrestrial Radio Access Network (E-URRAN): Overall Description,. [5] D. Astely, E. Dahlman, A. Furuskar, Y. Jading, M. Lindstrom, and S. Parkvall, LTE: The evolution of mobile broadband[LTE part II: 3GPP release 8], IEEE Commun. Mag., vol. 47, no. 4, pp. 4451, Apr. 2009. [6] V. Chandrasekhar and J. Andrews, Femtocell networks: A survey, IEEE Commun. Mag., vol. 46, no. 9, pp. 5967, Sep. 2008. [7] C.-Y. Chiu, J.-B. Yan, and R. D. Murch, Compact three-port orthogonally polarized MIMO antennas, IEEE Antennas Wireless Propag. Lett., vol. 6, pp. 619622, 2007. [8] K.-M. Luk and K.-W. Leung, Dielectric Resonator Antennas. Hertfordshire, England: Research Studies Press Ltd., 2003. [9] L. K. Hady, D. Kajfez, and A. A. Kishk, Triple mode use of a single dielectric resonator, IEEE Trans. Antennas Propag., vol. 57, no. 5, pp. 13281335, May 2009. [10] K. Ishimiya, J. Langbacka, Z. Ying, and J.-I. Takada, A compact MIMO DRA antenna, in Proc. IEEE Int. Workshop on Antenna Technology: Small Antennas and Novel Metamaterials (IWAT 08), Chiba, Japan, Mar. 2008, pp. 286289. [11] K. Ishimiya, Z. Ying, and J.-I. Takada, A compact MIMO DRA for 802.11n application, presented at the IEEE Antennas and Propagation Society Int. Symp., San Diego, CA, Jul. 2008. [12] A. Petosa, Dielectric Resonator Antenna Handbook. Norwood, MA: Artech House, 2007. [13] Mathematica, Wolfram Research Inc., 2010. [14] HFSS Ansys, Inc., 2010. [15] R. F. Harrington, Time-Harmonic Electromagnetic Fields. New York: IEEE Press, 2001. [16] A. Tadjalli and A. Sebak, Resonance frequencies and far eld patterns of elliptical dielectric resonator antenna: Analytical approach, in Progress in Electromagnetic Research, PIER 64, 2006, pp. 8198. [17] C.-C. Hsu, K. H. Lin, H.-L. Su, H.-H. Lin, and C.-Y. Wu, Design of MIMO antennas with strong isolation for portable applications, presented at the IEEE Antennas and Propagation Society Int. Symp., Charleston, SC, Jun. 2009. [18] H. Zhang, Z. Wang, J. Yu, and J. Huang, A compact MIMO antenna for wireless communication, IEEE Antennas Propag. Mag., vol. 50, no. 6, pp. 104107, Dec. 2008. [19] K.-S. Min, D.-J. Kim, and M.-S. Kim, Multi-channel MIMO antenna design for WiBro/PCS band, in Proc. IEEE Antennas and Propagation Society Int. Symp., Hawaii, Jun. 2007, pp. 12251228. [20] K. Chung and J. H. Yoon, Integrated MIMO antenna with high isolation characteristics, Electron. Lett., vol. 43, no. 4, pp. 199201, Feb. 2007. [21] M. Cooper, Investigation of Current and Novel Rectangular Dielectric Resonator Antennas for Broadband Applications at L-band Frequencies, M.Sc. thesis, Carleton University, Ottawa, ON, Canada, 1997. [22] S.-M. Deng, C.-L. Tsai, S.-F. Chang, and S.-S. Bor, A CPW-fed suspended, low prole rectangular dielectric resonator antenna for wideband operation, in Proc. IEEE Antennas and Propagation Society Int. Symp., Washington, D.C., Jul. 2005, vol. 4B, pp. 242245. [23] P. V. Bijumon, S. K. Menon, M. N. Suma, M. T. Sebastian, and P. Mohanan, Broadband cylindrical dielectric resonator antenna excited by modied microstrip line, Electron. Lett., vol. 41, no. 7, pp. 385387, Mar. 2005. [24] Wireless Insite, Remcom Inc., 2006. [25] J. D. Boerman and J. T. Bernhard, Performance study of pattern recongurable antennas in MIMO communication systems, IEEE Trans. Antennas Propag., vol. 56, no. 1, pp. 231236, Jan. 2008. [26] G. J. Foschini and M. J. Gans, On limits of wireless communications in a fading environment when using multiple antennas, in Wireless Personal Commun.. New York: Kluwer Academic Press, 1998, pp. 311335. [27] Z. Tang and A. S. Mohan, Experimental investigation of indoor MIMO Ricean channel capacity, IEEE Antennas Wireless Propag. Lett., vol. 4, pp. 5558, 2005. [28] Countis Laboratories [Online]. Available: http://www.countis.com/

Fig. 12. Measured -parameters of the perturbed cylindrical DRA.

with silver epoxy so as to prevent any air gaps between the dielectric and ground plane. This is important because for DRAs with high permittivities, air gaps of less than 0.05 mm can be enough to signicantly alter the expected input impedance [12]. The -parameters of the perturbed cylindrical DRA were measured using Agilents two-port Network Analyzer E8363B (with 50 reference impedance). The measured results are given in Fig. 12, which are very close to the simulated results given in Fig. 7. Both modes are well matched at 717 MHz. The coupling between the ports is less than at the operating frequency. The measured impedance bandwidths of the mode and the mode are 13.5 MHz and 35 MHz, respectively. The measured radiation patterns along the three principal cuts are given in Figs. 8 and 9. Despite a small distortion of the pattern at some angles, the measured patterns agreed reasonably well with the simulated ones. The complementary nature of the two orthogonal modes can still be observed clearly. VI. CONCLUSION A 2-port MIMO antenna based on a split-cylindrical DRA is described in this work. A mode degeneration method derived from perturbation theory is proposed to make the TE and HE modes of the split-cylindrical DRA resonate at the same frequency. The proposed method has been veried by both fullwave simulations and the boundary (elliptical) approximation method, and can be applied to DRAs of any shape. The fabricated MIMO DRA was tested and the experimental results show very good agreement with the simulated results. Indeed, given that the same operating frequency and the same dielectric material, the antenna described in this paper is smaller in volume, has lower prole, has a smaller ground plane and has much lower mutual coupling as compared to the work in [10], [11]. The proposed antenna is potentially suitable as the femtocell base station antenna in the forthcoming nationwide mobile broadband system based on LTE technology. Future work related to this paper will be a frequency recongurable MIMO DRA which can easily be adapted to other LTE bands and other wireless standards.

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Jie-Bang Yan (S09M11) received the B.Eng. degree (rst class honors) in electronic and communications engineering from the University of Hong Kong, in 2006, the M.Phil. degree in electronic and computer engineering from the Hong Kong University of Science and Technology, in 2008, and the Ph.D. degree in electrical and computer engineering from the University of Illinois at Urbana-Champaign, in 2011. He was a Croucher Scholar from 2009 to 2011 while he did his Ph.D. at the University of Illinois at Urbana-Champaign. Upon graduation, he joined the Center for Remote Sensing of Ice Sheets (CReSIS), University of Kansas, where he is currently an Assistant Research Professor. His research interests include design and analysis of MIMO and recongurable antennas, RF propagation, radar antenna designs, and fabrication of on-chip antennas. He holds two U.S. patents and a U.S. patent application related to novel antenna technologies. Dr. Yan was the recipient of the Best Paper Award at the 2007 IEEE (HK) AP/MTT Postgraduate Conference and the 2011 Raj Mittra Outstanding Research Award at Illinois. He serves as a Reviewer for several journals and conferences on antennas and electromagnetics.

Jennifer T. Bernhard (S89M95SM01F10) was born on May 1, 1966, in New Hartford, NY. She received the B.S.E.E. degree from Cornell University, Ithaca, NY, in 1988 and the M.S. and Ph.D. degrees in electrical engineering from Duke University, Durham, NC, in 1990 and 1994, respectively, with support from a National Science Foundation Graduate Fellowship. While at Cornell, she was a McMullen Deans Scholar and participated in the Engineering Co-op Program, working at IBM Federal Systems Division in Owego, New York. During the 199495 academic year she held the position

of Postdoctoral Research Associate with the Departments of Radiation Oncology and Electrical Engineering at Duke University, where she developed RF and microwave circuitry for simultaneous hyperthermia (treatment of cancer with microwaves) and MRI (magnetic resonance imaging) thermometry. From 19951999, she was an Assistant Professor in the Department of Electrical and Computer Engineering, University of New Hampshire, where she held the Class of 1944 Professorship. Since 1999, she has been with the Electromagnetics Laboratory, Department of Electrical and Computer Engineering, University of Illinois at Urbana-Champaign, where she is now a Professor. Her industrial experience includes work as a Research Engineer with Avnet Development Labs and, more recently, as a private consultant for members of the wireless communication and sensors community. Her research interests include recongurable and wideband microwave antennas and circuits, wireless sensors and sensor networks, high speed wireless data communication, electromagnetic compatibility, and electromagnetics for industrial, agricultural, and medical applications, and has four patents on technology in these areas. Prof. Bernhard is a member of URSI Commissions B and D, Tau Beta Pi, Eta Kappa Nu, Sigma Xi, and ASEE. She is a Fellow of the IEEE. She was an organizing member of the Women in Science and Engineering (WISE) Project at Duke, a graduate student-run organization designed to improve the climate for graduate women in engineering and the sciences. In 1999 and 2000, she was a NASA-ASEE Summer Faculty Fellow at the NASA Glenn Research Center, Cleveland, OH. She received the NSF CAREER Award in 2000. She is also an Illinois College of Engineering Willett Faculty Scholar and a Research Professor in Illinois Coordinated Science Laboratory, and the Information Trust Institute. She and her students received the 2004 H. A. Wheeler Applications Prize Paper Award from the IEEE Antenna and Propagation Society for their paper published in the March 2003 issue of the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION. She served as an Associate Editor for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION from 20012007 and for IEEE Antennas and Wireless Propagation Letters from 20012005. She is also a member of the editorial board of Smart Structures and Systems. She served as an elected member of the IEEE Antennas and Propagation Societys Administrative Committee from 20042006. She was President of the IEEE Antennas and Propagation Society in 2008.


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A Compact Eighteen-Port Antenna Cube for MIMO SystemsJianfeng Zheng, Xu Gao, Zhijun Zhang, Senior Member, IEEE, and Zhenghe Feng, Senior Member, IEEEAbstractAn 18-port compact antenna cube is proposed in this , paper. The cube, which has a volume of 0.76 0.76 0.76 provides 18 individual channels and is ideal for multiple-input multiple-output (MIMO) wireless communications. On each of the total six faces of the cube, a three-port tri-polarization antenna is installed. All antennas adopt a metal backing conguration, so the ground of all antennas forms a well shield Faraday cage, in which other functional circuits can be installed. Experimental measurements were carried out to evaluate the performance of the antenna cube in different MIMO scenarios. The results show that MIMO systems with the proposed compact antenna cube outperform those with dipole antennas which occupy the same number of RF channels but with much larger space. When a vertical 3-dipole array, a horizontal 3-dipole array and a dual polarization antenna are used in the user end (UE), respectively, the capacity of the global selected MIMO systems with antenna cube is about 2.7, 4.6, and 2.9 bits/s/Hz more than the full MIMO systems with a vertical 3-dipole array as the access point (AP) antennas. It is 1.9, 3.9, and 2.0 bits/s/Hz more than the full MIMO systems with a vertical 5-dipole array as AP antennas. The performance differences between the MIMO systems using global and simplied selection circuits are small. Index TermsAntenna cube, antenna selection, multiple-input multiple-output (MIMO), polarization.

I. INTRODUCTION PPLYING multiple-input multiple-output (MIMO) technology especially with antenna selection in access points (AP) can improve the overall system capacity. However, to construct enough antennas within a small volume is always a challenge. In previous works, a number of compact MIMO antennas have been proposed consisting of up to four ports, compact antenna designs with more than 10 ports are less common and mainly consist of a at panel approach and are used in large size base station. Recently an interesting approach, the antenna cube, emerges. An antenna cube takes advantage of spatial and polarization orthogonality to implement a large amount of antennasManuscript received December 20, 2010; revised March 28, 2011; accepted August 15, 2011. Date of publication October 25, 2011; date of current version February 03, 2012. This work was supported in part by the National Basic Research Program of China under Contract 2007CB310605, in part by the National Science and Technology Major Project of the Ministry of Science and Technology of China 2010ZX03007-001-01, in part by Qualcomm Inc., and in part by the Chuanxin Foundation of Tsinghua University. The authors are with the State Key Lab of Microwave and Communications, Tsinghua National Laboratory for Information Science and Technology, Tsinghua University, Beijing 100084, China (e-mail: [email protected]). Color versions of one or more of the gures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identier 10.1109/TAP.2011.2173449

A

within a constrained volume. In [1][3], MIMO cube accommodates up to 12 electrical dipole antennas on all its 12 edges. The 24-port and 36-port antenna cubes suitable for MIMO wireless communications are presented in [4]. However, existing cubes [1][4] demand a completely dedicated space for antennas. As the antenna elements in those cubes are omni-directional, the inner space must be kept empty to avoid performance degradation, i.e., other circuits cannot be installed in the space. To resolve the problem, a compact 18-port planar tri-polarization antenna cube for MIMO systems is proposed in this paper. A tri-polarization antenna makes full use of the promising polarization domain, which is considered an important resource for constructing compact antenna arrays and enhancing system performance [6][8]. The antenna cube employs tri-polarization antennas [9] as the basic elements. To form a compact antenna cube, six tri-polarization antennas are distributed on separate faces of a cube. This arrangement achieves low mutual coupling and wide coverage within a small volume mm with an operating frequency band of 2.402.48 GHz. In a real communication system, it is difcult to implement a large amount of RF channels even at AP. Thus some sorts of antenna switching must be involved for antenna-abundant MIMO systems [10][12]. Accompanying with the antenna cube, two simplied antenna switching schemes are proposed in this paper. Measurement results demonstrate that in an indoor environment, performance achieved by simplied switching schemes is almost as good as that of a fully switching system. Antenna design, measurement results and experimental verications of the proposed compact planar tri-polarization antenna cube are described in Sections IIV. Specically, the tripolarization antenna is briey introduced in Section I. Measurement results of the 18-port antenna cube are presented and discussed in Section II. Measurement procedure and analysis framework are explained in Section III. Experimental results of MIMO systems between the antenna cube and various terminal antennas are discussed in Section IV. Conclusion is drawn in Section V. II. ANTENNA CUBE DESIGN The conformal and low-prole tri-polarization antenna which was proposed in [9] is a fundamental building block in the planar MIMO cube and is briey introduced here. The conguration of the tri-polarization antenna is shown in Fig. 1. A ring patch, which functions as two independent orthogonal polarized antennas, and a disk-loaded monopole compose the tri-polarization antenna, and the operating frequency band is chosen to be 2.42.48 GHz.

0018-926X/$26.00 2011 IEEE

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Fig. 1. Geometry of the tri-polarization antenna: (a) top view and (b) side view.

Different from antennas used in other MIMO cubes proposed in the literature, the planar tri-polarization antenna has a very low prole, and the total height of the tri-polarization antenna is 5.8 mm. Furthermore, the full 74 74 mm sized ground plane of the tri-polarization antenna makes it particularly suitable for antenna mounted on the equipments. The advantage of low prole together with the easiness of its conformal integration on a cube surface makes the tri-polarization antenna a good candidate to construct the MIMO cube. Apart from the above advantage of the tri-polarization antenna, most importantly, the patch antenna mode and the monopole antenna mode of the tri-polarization have the orthogonal polarization property to each other. With three ports of this antenna working independently, the far eld of this antenna has three orthogonal linear polarizations. Specically, when the tri-polarization antenna is placed as in Fig. 1, i.e., the monopole is along the -axis while the feed lines P1 and P2 are along the - and -axes in horizontal plane, respectively, the E-eld radiated by the ring patch is parallel to the ground plane and can provide two orthogonal polarizations excited through P1 and P2, while the monopole provides the vertical polarization component and has an omni-directional radiation in the azimuth plane. Fig. 2 shows the measured radiation patterns of the tri-polarization antenna at 2.42 GHz. As shown, the radiation pattern of monopole mode (port M3) and patterns of the patch mode (port P1 and port P2) have orthogonal polarizations to each other. The gains of the directional slot-fed antennas at 2.42 GHz are 7.5 dBi for P1 and P2, while the gain of the omni-directional coaxial-fed disk-loaded monopole fed by M3 is 2.5 dBi. The main reason for the lower gain of M3 compared with the gains of other two ports is the different radiation properties between monopole and patch antennas.

The omni-directional radiation property gives the monopole mode lower gain compared to the directional patch mode. In real communication applications, the position of mobile terminals may rotate due to different communication scenarios and the arbitrariness of users behavior. For the fact that the three ports of this antenna radiate three polarized elds that are orthogonal to one another, this antenna could receive electromagnetic wave with any kind of polarization by switching among the ports of the antenna cube, thus avoid situations of the polarization mismatch. The tri-polarization antenna has a low planar prole and the complete common ground, thus it is easy to construct the planar tri-polarization antenna cube by embedding one tri-polarization antenna on each face of a cube. The structure of the planar tri-polarization antenna cube is shown in Fig. 3. As shown, the six planar tri-polarization antennas are xed on the six faces of the cube. Each antenna has 3 ports, and the antenna cube has 18 ports, which can provide up to 18 individual communication channels. The antenna cube operates at 2.42.48 GHz, and the volume is 94 94 94 mm , about 0.76 0.76 0.76 where is the wavelength in vacuum. For convenience of description, the faces of the cube are numbered as shown in Fig. 3, the up face is #1, the front, right, back and left faces are numbered as #2, #3, #4, and #5 respectively, and the bottom face is #6. The three ports in a face are noted as P1, P2, and M3. Each port in the antenna cube is denoted with the numbers of faces and ports, for example, F#1-P1 represents the P1 port of the tri-polarization antenna in the #1 face of the cube. For the three ports of each tri-polarization antenna in the face have three orthogonal polarizations, it is easy to obtain the full radiation coverage in the whole sphere. Therefore, the MIMO cube can provide good convergence for user terminals with any rotation and position. An important aspect to construct the antenna cube is to maintain relative low mutual coupling between any individual ports, as mutual coupling will deteriorate the performance of MIMO wireless communication systems. For the compact tri-polarization antenna cube, relatively low mutual coupling between antennas of the proposed MIMO cube is mainly due to the choices of antenna types, positions and orientations. As the three antennas in a tri-polarization antenna employ orthogonal polarizations, the mutual coupling between each port is relatively low. The tri-polarization antenna has a ground backing, so the tri-polarization antennas in different faces radiate toward different directions and inherently have low mutual coupling. To verify the performance of the planar tri-polarization antenna cube, a prototype antenna cube was fabricated, and the photo of the cube is shown in Fig. 3. Due to the symmetric characteristic of the antenna cube, only the tri-polarization antenna #1 is measured. The measured reection and transmission coefcients are shown in Fig. 4. The results are pretty much identical to the results reported in [9]. Between any two tri-polarization antennas in adjacent faces, there are nine sets of transmission coefcients. As shown in Fig. 5, there are three most signicant results between antennas in face #1 and #3. The isolations at 2.42.48 GHz band are all better than 20 dB. The isolations between ports in opposite

ZHENG et al.: COMPACT EIGHTEEN-PORT ANTENNA CUBE FOR MIMO SYSTEMS

447

Fig. 2. Measured electrical patterns of the tri-polarization antenna: (a) plane; (f) , plane.

,

plane; (b)

,

plane; (c)

,

plane; (d)

, plane; (e)

,

Fig. 3. Structure and photograph of the planar tri-polarization antenna cube.

antennas are all better than 25 dB, which is not illustrated here for the reason of concision. Overall, these results show that the proposed planar tri-polarization antenna cube has good isolation among the individual ports, which satises the requirement of MIMO systems. III. MIMO SYSTEMS WITH THE ANTENNA CUBE In prior works, the antennas presented for MIMO systems were often validated by examining the channel capacity of the full MIMO systems between antenna cubes in a narrow frequency band. However, the full MIMO systems which support more than 10 individual channels are too expensive and complicated to use in personal wireless communication systems nowadays, such as WLAN equipments, and the communication systems mostly are wideband. To overcome these shortcomings, the performance of the MIMO systems employing the antenna cube is examined in typical indoor scenarios with antenna selection among the whole WLAN frequency bands. The measurements were carried out in Room 1010 on the 10th oor of Weiqing Building in Tsinghua University, which is a

Fig. 4. (a) Measured return loss of the tri-polarization antenna #1 in the cube. (b) Measured isolation between ports in antenna #1.

typical laboratory room as schemed in Fig. 6. The framework of the room is reinforced concrete, the walls are mainly built

448


Fig. 7. Schematic of test-bench for MIMO system.

Fig. 5. Measured isolation between ports in adjacent tri-polarization antennas.

Fig. 8. Antennas used in measurement besides antenna cube: (a) vertical 5-dipole array; (b) vertical 3-dipole array; (c) horizontal 3-dipole array; (d) dual-polarization antenna.

Fig. 6. Structure of the measured ofce.

by brick and plaster, and the ceiling is made with plaster plates with aluminium alloy framework. The scheme of the test-bench is shown in Fig. 7. The measurement system consists of an Agilent E5071B network analyzer, AP antennas, user equipment (UE) antennas, RF switches, a computer, an auxiliary amplier, and RF cables. The AP and the UE are connected to a 16-to-1 RF switch and a 4-to-1 RF switch respectively, and the switches are then connected to the network analyzer. The auxiliary amplier is between the transmit antenna and the network analyzer to amplify the transmit signal. The computer controls the measurement procedure and records the data. In the measurement, the transmit power of network analyzer is set to 10 dBm, IFBW is 10 kHz, and sweep averaging is set on with sweep averaging factor as 16, the noise oor of the network analyzer is below than 90 dB when measuring S21. The loss of the cable is less than 15 dB, the insertion loss of the switch is about 4 dB and the power gain of the amplier is about 10

dB. With the SNR limitation of 15 dB, the dynamic range of the measurement system is above 66 dB. For the conveniences of measurement and installation, the tri-polarization antenna in the bottom face of cube was removed, thus only 15 ports of the cube were used. The congurations of the measurements are listed in Table I. The measurement campaign was carried out for twelve representative MIMO systems, and the measured channel responses are noted as , here is the type number of AP antennas and is the one of UE antennas. On the AP side, four different arrays were used respectively. They are a vertical three-dipole array, a vertical ve-dipole array, and an antenna cube with three/ve selected branches. The separation between adjacent antennas of the dipole array is one wavelength, so the three/ve-dipole arrays size is two/four wavelengths. On the UE side, three different arrays were used alternatively. They are a vertical threedipole array, a horizontal three-dipole array and a compact dual polarization antenna [13]. The size of three-dipole array is two wavelengths and the dual polarization antenna is 0.8 wavelength in size. The schemes of all dipole arrays and the dual-polarization antenna are shown in Fig. 8. In each measurement, the AP antenna was xed in the center of the ofce room with a height of 1.2 m, and UE antennas were placed in the 10 locales around the room sequentially with a height of about 0.8 m. The locales UE2, 5, 8, 10 had Line-ofsight (LOS) paths and UE1, 3, 4, 6, 7, 9 only had non-line-ofsight (NLOS) paths, as illustrated in Fig. 6, where the locales UE5, 10 and the locales UE2, 8 were in the broad-sight and

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TABLE I MEASUREMENT CONFIGURATIONS

A. Transmitter Power Constraints As the received power and richness of scattering are quite different at different UE locations, the measured channel response matrices must be appropriately normalized. For the rich scattering required to achieve low correlations for MIMO communications often produces low SNR, which in turn decreases the channel capacity [14], we adopt the MIMO system with vertical 5-dipole at AP and vertical 3-dipole at UE as reference to normalize the channel responses with average transmitted power as discussed in [15]. This normalization considers not only richness of the multipath but also the power gain. Obviously, the can be expressed as in (1)

TABLE II CONFIGURATIONS OF THE STUDIED MIMO SYSTEMS

where , which is different with the average SNR of the measurements, is the assumed average SNR of the referenced MIMO systems with channel response , is the number of the locales and is the number of the measured points in each locale, denotes the number of the channel bands, and are the numbers of transmit and receive antennas of channel responses, respectively, and is the number of the measured frequency bins in the th band. means the average received noise per frequency bin, and is the trace operation. In the following, the assumed average SNR is set to 15 dB in analyzing the channel capacity. B. Channel Capacity of MIMO Systems With Antenna Selections Over Wide Bands

end-re directions of the referenced dipole array at AP. In each place, channel matrices at 4 points separated by 6.5 cm, i.e., half-wavelength, were measured in order to obtain independent fading, and denoted as , where represents the serial number of the locale and is the serial number of measurement point in the locale place. For each , the responses over the whole WLAN band were measured. As we measured the channel responses after midnight and before dawn, the channels were supposed to be static, so the elements in the channel matrix were measured in sequence and the switching of the was completed by using RF switches. In the measurements of referenced MIMO system with a vertical 5-dipole at AP and a 3-dipole array at UE, the maximum measured S21 is 45 dB, and the average measured S21 is 52 dB. That is, the average SNR of the measurement is about 52 + 90 dB 38 dB with referenced linear dipole arrays, here 90 dB is the S21 noise oor of the proposed measurement system. The measured channel responses are assorted to construct the wideband channel of referenced dipole-array MIMO systems and MIMO systems with antenna cube. The frequency bands are divided following the IEEE 802.11 specications as shown in Table II. That means, when studying the channel capacity of any MIMO system with specied antennas and places, 14 wideband channels are adopted based on the frequency partition of IEEE 802.11 specications.

Though prior proposed cubes were demonstrated in full MIMO systems, the transceivers of a full MIMO system using antenna cubes might be too expensive to accommodate in todays personal wireless communication systems. Antenna selection is a good approach to reduce a systems cost while maintain its performance. The bulk selection [16], [17] method is adopted as a reference. Bulk selection method is a global optimization method, which assumes there is a direct path between any input port and output port. The channel capacity with equal power emission strategy [18] is adopted to evaluate performance of measured MIMO systems covering the th band. Then channel capacity of wideband systems with bulk antenna selection is (2) denotes the different combination of AP antenna elwhere ements and is the set of the selectable antenna combinations. Assuming there are 15 ports in the AP, each time the total number of combinations is 455 and 3003 when 3 and 5 branches are selected respectively. On the UE side, all available antennas are always used. is the channel capacity of the MIMO systems with selected antennas combination over the th band, and expressed as (3)

450


Fig. 9. Congurations of the switching circuits: (a) 15-to-3 global selection circuit; (b) pattern selection circuit; (c) 15-to-5 global selection circuit; (d) polarization selection circuit.

formance of the MIMO systems will depend not only on the number of the antennas used in the base station but also on the radiation pattern, polarization and array structure. The whole spherical coverage characteristic and capability of receiving any polarized impinging wave make the compact antenna cube particularly suitable for indoor communications. We examined the performance of the MIMO systems with the antenna cube and various terminal antennas in different locales and postures as followed. The performance of the selection MIMO systems with antenna cube in AP is compared with that of the referenced full MIMO systems with the often used uniformly spaced vertical dipole arrays. In the following studies, the measured data of all locations illustrated in Fig. 6 is adopted. The number of locations is 10. Four spots are measured at each location. Each measurement includes 14 channels. The total size of the channel samples for all following gures is and in each channel sample, 23 frequency bins are measured with a frequency step of 1 MHz to cover the 22 MHz channel bandwidth. A. Average Normalized Receive Power In wireless communication systems, the performance is affected by the signal to noise ratio. Thus, the capability of collecting more power is quite important to AP antennas. The average normalized receive power of the compact antenna cube and the referenced dipole arrays with various antenna at UE are listed in Table III, which is normalized according to the average receive power on each port of referenced vertical 5-dipole array at AP with vertical 3-dipole array at UE as

C. Simplied Pattern and Polarization Selection Methods Although antenna selection is capable of reducing the cost of the RF channels while maintaining the performance of the MIMO systems, determinations of the forms of the antenna array and implementations of the RF selection circuits are not trivial [12], [19][21]. The existing research activities on antenna selection little involve designs of antenna arrays and selection circuits. The often used or assumed global selection circuits require many RF switches and complicated RF circuits, which are difcult to realize and may introduce inevitable high insertion loss. Two simplied selection circuits with low complexity and cost are presented to reduce the complexity of global selection circuit and maintain a comparable performance, which are pattern and polarization selection circuits. As shown in Fig. 9,

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