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Analog Integrated Circuits

Feb 19, 2018



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    Analog Integrated Circuits

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    Bode plot 1

    Current mirror 11

    Differential amplifier 19

    Operational amplifier 25


    Article Sources and Contributors 43

    Image Sources, Licenses and Contributors 44

    Article Licenses

    License 45

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    Bode plot 1

    Bode plot

    Figure 1(a): The Bode plot for a first-order (one-pole) highpass filter; the straight-line

    approximations are labeled "Bode pole"; phase varies from 90 at low frequencies (due to

    the contribution of the numerator, which is 90 at all frequencies) to 0 at high

    frequencies (where the phase contribution of the denominator is 90 and cancels the

    contribution of the numerator).

    Figure 1(b): The Bode plot for a first-order (one-pole) lowpass filter; the straight-line

    approximations are labeled "Bode pole"; phase is 90 lower than for Figure 1(a) because

    the phase contribution of the numerator is 0 at all frequencies.

    A Bode plot is a graph of the transfer

    function of a linear, time-invariant

    system versus frequency, plotted with

    a log-frequency axis, to show the

    system's frequency response. It is

    usually a combination of a Bode

    magnitude plot, expressing the

    magnitude of the frequency response

    gain, and a Bode phase plot,

    expressing the frequency response

    phase shift.


    Among his several important

    contributions to circuit theory and

    control theory, engineer Hendrik Wade

    Bode (19051982), while working at

    Bell Labs in the United States in the

    1930s, devised a simple but accurate

    method for graphing gain and

    phase-shift plots. These bear his name,

    Bode gain plot and Bode phase plot

    (pronounced Boh-dee in English,

    Bow-duh in Dutch).[1]

    The magnitude axis of the Bode plot is

    usually expressed as decibels of power,

    that is by the 20 log rule: 20 times the

    common (base 10) logarithm of the

    amplitude gain. With the magnitude

    gain being logarithmic, Bode plots

    make multiplication of magnitudes a

    simple matter of adding distances on

    the graph (in decibels), since

    A Bode phase plot is a graph of phase

    versus frequency, also plotted on a

    log-frequency axis, usually used in

    conjunction with the magnitude plot, to

    evaluate how much a signal will be

    phase-shifted. For example a signal
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    Bode plot 2

    described by:Asin(t) may be attenuated but also phase-shifted. If the system attenuates it by a factor x and phase

    shifts it by the signal out of the system will be (A/x) sin(t). The phase shift is generally a function of


    Phase can also be added directly from the graphical values, a fact that is mathematically clear when phase is seen as

    the imaginary part of the complex logarithm of a complex gain.

    In Figure 1(a), the Bode plots are shown for the one-pole highpass filter function:

    where f is the frequency in Hz, and f1

    is the pole position in Hz, f1= 100 Hz in the figure. Using the rules for

    complex numbers, the magnitude of this function is

    while the phase is:

    Care must be taken that the inverse tangent is set up to return degrees, not radians. On the Bode magnitude plot,

    decibels are used, and the plotted magnitude is:

    In Figure 1(b), the Bode plots are shown for the one-pole lowpass filter function:

    Also shown in Figure 1(a) and 1(b) are the straight-line approximations to the Bode plots that are used in hand

    analysis, and described later.

    The magnitude and phase Bode plots can seldom be changed independently of each other changing the amplitude

    response of the system will most likely change the phase characteristics and vice versa. For minimum-phase systems

    the phase and amplitude characteristics can be obtained from each other with the use of the Hilbert transform.

    If the transfer function is a rational function with real poles and zeros, then the Bode plot can be approximated with

    straight lines. These asymptotic approximations are called straight line Bode plots or uncorrected Bode plots and

    are useful because they can be drawn by hand following a few simple rules. Simple plots can even be predicted

    without drawing them.

    The approximation can be taken further by correcting the value at each cutoff frequency. The plot is then called a

    corrected Bode plot.
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    Bode plot 3

    Rules for hand-made Bode plot

    The premise of a Bode plot is that one can consider the log of a function in the form:

    as a sum of the logs of its poles and zeros:

    This idea is used explicitly in the method for drawing phase diagrams. The method for drawing amplitude plots

    implicitly uses this idea, but since the log of the amplitude of each pole or zero always starts at zero and only has one

    asymptote change (the straight lines), the method can be simplified.

    Straight-line amplitude plot

    Amplitude decibels is usually done using the version. Given a transfer function in the form

    where and are constants, , , andH is the transfer function:

    at every value of s where (a zero), increase the slope of the line by per decade.

    at every value of s where (a pole), decrease the slope of the line by per decade.

    The initial value of the graph depends on the boundaries. The initial point is found by putting the initial angular

    frequency into the function and finding |H(j)|.

    The initial slope of the function at the initial value depends on the number and order of zeros and poles that are at

    values below the initial value, and are found using the first two rules.

    To handle irreducible 2nd order polynomials, can, in many cases, be approximated as


    Note that zeros and poles happen when is equal to a certain or . This is because the function in question isthe magnitude of H(j), and since it is a complex function, . Thus at any place where there

    is a zero or pole involving the term , the magnitude of that term is


    Corrected amplitude plot

    To correct a straight-line amplitude plot:

    at every zero, put a point above the line,

    at every pole, put a point below the line,

    draw a smooth curve through those points using the straight lines as asymptotes (lines which the curveapproaches).

    Note that this correction method does not incorporate how to handle complex values of or . In the case of an

    irreducible polynomial, the best way to correct the plot is to actually calculate the magnitude of the transfer function

    at the pole or zero corresponding to the irreducible polynomial, and put that dot over or under the line at that pole or

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    Bode plot 4

    Straight-line phase plot

    Given a transfer function in the same form as above:

    the idea is to draw separate plots for each pole and zero, then add them up. The actual phase curve is given by


    To draw the phase plot, for each pole and zero:

    if A is positive, start line (with zero slope) at 0 degrees

    if A is negative, start line (with zero slope) at 180 degrees

    at every (for stable zeros ), increase the slope by degrees per decade,

    beginning one decade before (E.g.: )

    at every (for stable poles ), decrease the slope by degrees per decade,

    beginning one decade before (E.g.: )

    "unstable" (right half plane) poles and zeros ( ) have opposite behavior flatten the slope again when the phase has changed by degrees (for a zero) or degrees (for a


    After plotting one line for each pole or zero, add the lines together to obtain the final phase plot; that is, the final

    phase plot is the superposition of each earlier phase plot.


    A passive (unity pass band gain) lowpass RC filter, for instance has the following transfer function expressed in the

    frequency domain:

    From the transfer function it can be determined that the cutoff frequency pointfc

    (in hertz) is at the frequency

    or (equivalently) at

    where is the angular cutoff frequency in radians per second.

    The transfer function in terms of the angular frequencies becomes:

    The above equation is the normalized form of the transfer function. The Bode plot is shown in Figure 1(b) above,

    and construction of the straight-line approximation is discussed next.
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    Bode plot 5

    Magnitude plot

    The magnitude (in decibels) of the transfer function above, (normalized and converted to angular frequency form),

    given by the decibel gain expression :

    when plotted versus input frequency on a logarithmic scale, can be approximated by two lines and it forms the

    asymptotic (approximate) magnitude Bode plot of the transfer function:

    for angular frequencies below it is a horizontal line at 0 dB since at low frequencies the term is small and

    can be neglected, making the decibel gain equation above equal to zero,

    for angular frequencies above it is a line with a slope of 20 dB per decade since at high frequencies the

    term dominates and the decibel gain expression above simplifies to which is a straight line with a

    slope of 20 dB per decade.

    These two lines meet at the corner frequency. From the plot, it can be seen that for frequencies well below the corner

    frequency, the circuit has an attenuation of 0 dB, corresponding to a unity pass band gain, i.e. the amplitude of the

    filter output equals the amplitude of the input. Frequencies above the corner frequency are attenuated the higher

    the frequency, the higher the attenuation.

    Phase plot

    The phase Bode plot is obtained by plotting the phase angle of the transfer function given by

    versus , where and are the input and cutoff angular frequencies respectively. For input frequencies much

    lower than corner, the ratio is small and therefore the phase angle is close to zero. As the ratio increases the

    absolute value of the phase increases and becomes 45 degrees when . As the ratio increases for input

    frequencies much greater than the corner frequency, the phase angle asymptotically approaches 90 degrees. The

    frequency scale for the phase plot is logarithmic.

    Normalized plot

    The horizontal frequency axis, in both the magnitude and phase plots, can be replaced by the normalized

    (nondimensional) frequency ratio . In such a case the plot is said to be normalized and units of the frequencies

    are no longer used since all input frequencies are now expressed as multiples of the cutoff frequency .

    An example with pole and zero

    Figures 2-5 further illustrate construction of Bode plots. This example with both a pole and a zero shows how to use

    superposition. To begin, the components are presented separately.

    Figure 2 shows the Bode magnitude plot for a zero and a low-pass pole, and compares the two with the Bode straight

    line plots. The straight-line plots are horizontal up to the pole (zero) location and then drop (rise) at 20 dB/decade.

    The second Figure 3 does the same for the phase. The phase plots are horizontal up to a frequency factor of ten

    below the pole (zero) location and then drop (rise) at 45/decade until the frequency is ten times higher than the pole
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    Bode plot 6

    (zero) location. The plots then are again horizontal at higher frequencies at a final, total phase change of 90.

    Figure 4 and Figure 5 show how superposition (simple addition) of a pole and zero plot is done. The Bode straight

    line plots again are compared with the exact plots. The zero has been moved to higher frequency than the pole to

    make a more interesting example. Notice in Figure 4 that the 20 dB/decade drop of the pole is arrested by the 20

    dB/decade rise of the zero resulting in a horizontal magnitude plot for frequencies above the zero location. Notice in

    Figure 5 in the phase plot that the straight-line approximation is pretty approximate in the region where both poleand zero affect the phase. Notice also in Figure 5 that the range of frequencies where the phase changes in the

    straight line plot is limited to frequencies a factor of ten above and below the pole (zero) location. Where the phase

    of the pole and the zero both are present, the straight-line phase plot is horizontal because the 45/decade drop of the

    pole is arrested by the overlapping 45/decade rise of the zero in the limited range of frequencies where both are

    active contributors to the phase.

    Example with pole and zero

    Figure 2: Bode magnitude plot for zero and low-pass pole; curves

    labeled "Bode" are the straight-line Bode plots

    Figure 3: Bode phase plot for zero and low-pass pole; curves labeled

    "Bode" are the straight-line Bode plots

    Figure 4: Bode magnitude plot for pole-zero combination; the location

    of the zero is ten times higher than in Figures 2&3; curves labeled

    "Bode" are the straight-line Bode plots

    Figure 5: Bode phase plot for pole-zero combination; the location of

    the zero is ten times higher than in Figures 2&3; curves labeled "Bode"

    are the straight-line Bode plots

    Gain margin and phase margin

    Bode plots are used to assess the stability of negative feedback amplifiers by finding the gain and phase margins of

    an amplifier. The notion of gain and phase margin is based upon the gain expression for a negative feedback

    amplifier given by

    where AFB

    is the gain of the amplifier with feedback (the closed-loop gain), is the feedback factor andAOL

    is the

    gain without feedback (the open-loop gain). The gainAOL

    is a complex function of frequency, with both magnitude

    and phase.[2]

    Examination of this relation shows the possibility of infinite gain (interpreted as instability) if theproduct A

    OL= 1. (That is, the magnitude of A

    OLis unity and its phase is 180, the so-called Barkhausen
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    Bode plot 7

    stability criterion). Bode plots are used to determine just how close an amplifier comes to satisfying this condition.

    Key to this determination are two frequencies. The first, labeled here as f180

    , is the frequency where the open-loop

    gain flips sign. The second, labeled here f0dB

    , is the frequency where the magnitude of the product | AOL

    | = 1 (in

    dB, magnitude 1 is 0 dB). That is, frequencyf180

    is determined by the condition:

    where vertical bars denote the magnitude of a complex number (for example, | a+jb | = [ a2 + b2]1/2 ), and


    is determined by the condition:

    One measure of proximity to instability is the gain margin. The Bode phase plot locates the frequency where the

    phase of AOL

    reaches 180, denoted here as frequencyf180

    . Using this frequency, the Bode magnitude plot finds

    the magnitude of AOL

    . If |AOL


    = 1, the amplifier is unstable, as mentioned. If |AOL


    < 1, instability does not

    occur, and the separation in dB of the magnitude of |AOL


    from |AOL

    | = 1 is called the gain margin. Because a

    magnitude of one is 0 dB, the gain margin is simply one of the equivalent forms: 20 log10

    ( |AOL


    ) = 20 log10




    ) 20 log10

    ( 1 / ).

    Another equivalent measure of proximity to instability is the phase margin. The Bode magnitude plot locates thefrequency where the magnitude of |A

    OL| reaches unity, denoted here as frequency f

    0dB. Using this frequency, the

    Bode phase plot finds the phase of AOL

    . If the phase of AOL


    ) > 180, the instability condition cannot be met

    at any frequency (because its magnitude is going to be < 1 whenf = f180

    ), and the distance of the phase atf0dB


    degrees above 180 is called thephase margin.

    If a simpleyes or no on the stability issue is all that is needed, the amplifier is stable if f0dB

    < f180

    . This criterion is

    sufficient to predict stability only for amplifiers satisfying some restrictions on their pole and zero positions

    (minimum phase systems). Although these restrictions usually are met, if they are not another method must be used,

    such as the Nyquist plot.[3][4]

    Examples using Bode plots

    Figures 6 and 7 illustrate the gain behavior and terminology. For a three-pole amplifier, Figure 6 compares the Bode

    plot for the gain without feedback (the open-loop gain)AOL

    with the gain with feedbackAFB

    (the closed-loop gain).

    See negative feedback amplifier for more detail.

    In this example,AOL

    = 100 dB at low frequencies, and 1 / = 58 dB. At low frequencies,AFB

    58 dB as well.

    Because the open-loop gain AOL

    is plotted and not the product AOL

    , the conditionAOL

    = 1 / decides f0dB

    . The

    feedback gain at low frequencies and for large AOL


    1 / (look at the formula for the feedback gain at the

    beginning of this section for the case of large gain AOL

    ), so an equivalent way to find f0dB

    is to look where the

    feedback gain intersects the open-loop gain. (Frequencyf0dB

    is needed later to find the phase margin.)

    Near this crossover of the two gains at f0dB, the Barkhausen criteria are almost satisfied in this example, and the

    feedback amplifier exhibits a massive peak in gain (it would be infinity if AOL

    = 1). Beyond the unity gain


    , the open-loop gain is sufficiently small thatAFB


    (examine the formula at the beginning of this

    section for the case of smallAOL


    Figure 7 shows the corresponding phase comparison: the phase of the feedback amplifier is nearly zero out to the


    where the open-loop gain has a phase of 180. In this vicinity, the phase of the feedback amplifier

    plunges abruptly downward to become almost the same as the phase of the open-loop amplifier. (Recall, AFB


    for smallAOL


    Comparing the labeled points in Figure 6 and Figure 7, it is seen that the unity gain frequencyf0dB

    and the phase-flip


    are very nearly equal in this amplifier, f180


    3.332 kHz, which means the gain margin and

    phase margin are nearly zero. The amplifier is borderline stable.
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    Bode plot 8

    Figures 8 and 9 illustrate the gain margin and phase margin for a different amount of feedback . The feedback

    factor is chosen smaller than in Figure 6 or 7, moving the condition | AOL

    | = 1 to lower frequency. In this example,

    1 / = 77 dB, and at low frequenciesAFB

    77 dB as well.

    Figure 8 shows the gain plot. From Figure 8, the intersection of 1 / and AOL

    occurs atf0dB

    = 1 kHz. Notice that the

    peak in the gainAFB


    is almost gone.[5][6]

    Figure 9 is the phase plot. Using the value of f0dB = 1 kHz found above from the magnitude plot of Figure 8, theopen-loop phase atf

    0dBis 135, which is a phase margin of 45 above 180.

    Using Figure 9, for a phase of 180 the value of f180

    = 3.332 kHz (the same result as found earlier, of course[7]


    The open-loop gain from Figure 8 atf180

    is 58 dB, and 1 / = 77 dB, so the gain margin is 19 dB.

    Stability is not the sole criterion for amplifier response, and in many applications a more stringent demand than

    stability is good step response. As a rule of thumb, good step response requires a phase margin of at least 45, and

    often a margin of over 70 is advocated, particularly where component variation due to manufacturing tolerances is

    an issue.[8]

    See also the discussion of phase margin in the step response article.


    Figure 6: Gain of feedback amplifierAFB

    in dB and corresponding

    open-loop amplifierAOL

    . Parameter 1/ = 58 dB, and at low

    frequenciesAFB 58 dB as well. The gain margin in this amplifier isnearly zero because | A

    OL| = 1 occurs at almostf =f


    Figure 7: Phase of feedback amplifier AFB

    in degrees and

    corresponding open-loop amplifier AOL

    . The phase margin in this

    amplifier is nearly zero because the phase-flip occurs at almost theunity gain frequencyf =f

    0dBwhere | A

    OL| = 1.

    Figure 8: Gain of feedback amplifierAFB in dB and correspondingopen-loop amplifierA

    OL. In this example, 1 / = 77 dB. The gain

    margin in this amplifier is 19 dB.

    Figure 9: Phase of feedback amplifierAFB in degrees andcorresponding open-loop amplifierA

    OL. The phase margin in this

    amplifier is 45.
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    Bode plot 9

    Bode plotter

    Figure 10: Amplitude diagram of a 10th order Chebyshev filter plotted using a Bode

    Plotter application. The chebyshev transfer function is defined by poles and zeros which

    are added by clicking on a graphical complex diagram.

    The Bode plotter is an electronic

    instrument resembling an oscilloscope,

    which produces a Bode diagram, or a

    graph, of a circuit's voltage gain or

    phase shift plotted against frequency in

    a feedback control system or a filter.

    An example of this is shown in Figure

    10. It is extremely useful for analyzing

    and testing filters and the stability of

    feedback control systems, through the

    measurement of corner (cutoff)

    frequencies and gain and phase


    This is identical to the functionperformed by a vector network analyzer, but the network analyzer is typically used at much higher frequencies.

    For education/research purposes, plotting Bode diagrams for given transfer functions facilitates better understanding

    and getting faster results (see external links).

    Related plots

    Two related plots that display the same data in different coordinate systems are the Nyquist plot and the Nichols plot.

    These are parametric plots, with frequency as the input and magnitude and phase of the frequency response as the

    output. The Nyquist plot displays these in polar coordinates, with magnitude mapping to radius and phase to

    argument (angle). The Nichols plot displays these in rectangular coordinates, on the log scale.

    Related Plots

    A Nyquist plot. A Nichols plot of the same response.
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    Bode plot 10


    [1] Van Valkenburg, M. E. University of Illinois at Urbana-Champaign, "In memoriam: Hendrik W. Bode (1905-1982)", IEEE Transactions on

    Automatic Control, Vol. AC-29, No 3., March 1984, pp. 193-194. Quote: "Something should be said about his name. To his colleagues at Bell

    Laboratories and the generations of engineers that have followed, the pronunciation is boh-dee. The Bode family preferred that the original

    Dutch be used as boh-dah."

    [2] Ordinarily, as frequency increases the magnitude of the gain drops and the phase becomes more negative, although these are only trends and

    may be reversed in particular frequency ranges. Unusual gain behavior can render the concepts of gain and phase margin inapplicable. Thenother methods such as the Nyquist plot have to be used to assess stability.

    [3] Thomas H. Lee (2004). The design of CMOS radio-frequency integrated circuits(http://worldcat. org/isbn/0-521-83539-9) (Second Edition

    ed.). Cambridge UK: Cambridge University Press. p. 14.6 pp. 451453. ISBN 0-521-83539-9. .

    [4] William S Levine (1996). The control handbook: the electrical engineering handbook series(http://books. google. com/


    sig=ad5DJ7EvVm6In_zhI0MlF_6vHDA) (Second Edition ed.). Boca Raton FL: CRC Press/IEEE Press. p. 10.1 p. 163. ISBN 0849385709. .

    [5] The critical amount of feedback where the peak in the gainjust disappears altogether is the maximally flat or Butterworth design.

    [6] Willy M C Sansen (2006).Analog design essentials(http://worldcat. org/isbn/0-387-25746-2). Dordrecht, The Netherlands: Springer.

    p. 0517-0527 pp. 157163. ISBN 0-387-25746-2. .

    [7] The frequency where the open-loop gain flips signf180

    does not change with a change in feedback factor; it is a property of the open-loop

    gain. The value of the gain atf180

    also does not change with a change in . Therefore, we could use the previous values from Figures 6 and 7.

    However, for clarity the procedure is described using only Figures 8 and 9.

    [8] Willy M C Sansen. 0526 p. 162( ISBN 0-387-25746-2. .


    External links

    Explanation of Bode plots with movies and examples (


    How to draw piecewise asymptotic Bode plots (

    Summarized drawing rules (

    pdf) (PDF)

    Bode plot applet ( - Accepts transfer function coefficients as

    input, and calculates magnitude and phase response

    Circuit analysis in electrochemistry (

    Tim Green: Operational amplifier stability(

    Includes some Bode plot introduction

    Gnuplot code for generating Bode plot: DIN-A4 printing template (pdf)
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    Current mirror 11

    Current mirror

    A current mirror is a circuit designed to copy a current through one active device by controlling the current in

    another active device of a circuit, keeping the output current constant regardless of loading. The current being

    'copied' can be, and sometimes is, a varying signal current. Conceptually, an ideal current mirror is simply an ideal

    inverting current amplifier that reverses the current direction as well or it is a current-controlled current source

    (CCCS). The current mirror is used to provide bias currents and active loads to circuits

    Mirror characteristics

    There are three main specifications that characterize a current mirror. The first is the transfer ratio (in the case of a

    current amplifier) or the output current magnitude (in the case of a constant current source CCS). The second is its

    AC output resistance, which determines how much the output current varies with the voltage applied to the mirror.

    The third specification is the minimum voltage drop across the output part of the mirror necessary to make it work

    properly. This minimum voltage is dictated by the need to keep the output transistor of the mirror in active mode.

    The range of voltages where the mirror works is called the compliance range and the voltage marking the boundarybetween good and bad behavior is called the compliance voltage. There are also a number of secondary performance

    issues with mirrors, for example, temperature stability.

    Practical approximations

    For small-signal analysis the current mirror can be approximated by its equivalent Norton impedance .

    In large-signal hand analysis, a current mirror is usually and simply approximated by an ideal current source.

    However, an ideal current source is unrealistic in several respects:

    it has infinite AC impedance, while a practical mirror has finite impedance

    it provides the same current regardless of voltage, that is, there are no compliance range requirements it has no frequency limitations, while a real mirror has limitations due to the parasitic capacitances of the


    the ideal source has no sensitivity to real-world effects like noise, power-supply voltage variations and component


    Circuit realizations of current mirrors

    Basic idea

    A current mirror consists of two cascaded inverse converters with mirrored transfer


    A bipolar transistor can be used as the

    simplest current-to-current converter

    but its transfer ratio would highly

    depend on temperature variations,

    tolerances, etc. To eliminate these

    undesired disturbances, a current

    mirror is composed of two cascaded

    current-to-voltage and

    voltage-to-current converters placed at

    the same conditions and having reverse characteristics. They have not to be obligatory linear; the only requirement is

    their characteristics to be mirrorlike (for example, in the BJT current mirror below, they are logarithmic and
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    Current mirror 12

    exponential). Usually, two identical converters are used but the characteristic of the first one is reversed by applying

    a negative feedback. Thus a current mirror consists of two cascaded equal converters (the first - reversed and the

    second - direct).

    Figure 1: A current mirror implemented with npn

    bipolar transistors using a resistor to set the reference

    current IREF

    ; VCC

    = supply voltage

    Basic BJT current mirror

    If a voltage is applied to the BJT base-emitter junction as an input

    quantity and the collector current is taken as an output quantity,

    the transistor will act as an exponential voltage-to-current

    converter. By applying a negative feedback (simply joining the

    base and collector) the transistor can be "reversed" and it will

    begin acting as the opposite logarithmic current-to-voltage

    converter; now it will adjust the "output" base-emitter voltage so

    as to pass the applied "input" collector current.

    The simplest bipolar current mirror (shown in Figure 1)

    implements this idea. It consists of two cascaded transistor stages

    acting accordingly as a reversed and direct voltage-to-current

    converters. Transistor Q1

    is connected to ground. Its collector-base

    voltage is zero as shown. Consequently, the voltage drop across Q1

    is VBE

    , that is, this voltage is set by the diode law and Q1

    is said to

    be diode connected. (See also Ebers-Moll model.) It is important to have Q1

    in the circuit instead of a simple diode,

    because Q1

    sets VBE

    for transistor Q2. If Q

    1and Q

    2are matched, that is, have substantially the same device

    properties, and if the mirror output voltage is chosen so the collector-base voltage of Q2

    is also zero, then the


    -value set by Q1

    results in an emitter current in the matched Q2

    that is the same as the emitter current in Q1.

    Because Q1

    and Q2

    are matched, their 0-values also agree, making the mirror output current the same as the

    collector current of Q1. The current delivered by the mirror for arbitrary collector-base reverse bias VCB of the outputtransistor is given by (see bipolar transistor):



    = reverse saturation current or scale current, VT

    = thermal voltage and VA

    = Early voltage. This current is

    related to the reference currentIREF

    when the output transistor VCB

    = 0 V by:

    as found using Kirchhoff's current law at the collector node of Q1:

    The reference current supplies the collector current to Q1

    and the base currents to both transistors when both

    transistors have zero base-collector bias, the two base currents are equal, IB1




    Parameter 0

    is the transistor -value for VCB

    = 0 V.
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    Current mirror 13

    Output resistance

    If VCB

    is greater than zero in output transistor Q2, the collector current in Q

    2will be somewhat larger than for Q


    to the Early effect. In other words, the mirror has a finite output (or Norton) resistance given by the rO

    of the output

    transistor, namely (see Early effect):


    where VA

    = Early voltage and VCB

    = collector-to-base bias.

    Compliance voltage

    To keep the output transistor active, VCB

    0 V. That means the lowest output voltage that results in correct mirror

    behavior, the compliance voltage, is VOUT

    = VCV

    = VBE

    under bias conditions with the output transistor at the output

    current levelIC

    and with VCB

    = 0 V or, inverting theI-V relation above:

    where VT

    = thermal voltage andIS

    = reverse saturation current or scale current.

    Extensions and complications

    When Q2

    has VCB

    > 0 V, the transistors no longer are matched. In particular, their -values differ due to the Early

    effect, with

    where VA

    is the Early voltage and 0

    = transistor for VCB

    = 0 V. Besides the difference due to the Early effect, the

    transistor -values will differ because the 0-values depend on current, and the two transistors now carry different

    currents (see Gummel-Poon model).

    Further, Q2

    may get substantially hotter than Q1

    due to the associated higher power dissipation. To maintain

    matching, the temperature of the transistors must be nearly the same. In integrated circuits and transistor arrayswhere both transistors are on the same die, this is easy to achieve. But if the two transistors are widely separated, the

    precision of the current mirror is compromised.

    Additional matched transistors can be connected to the same base and will supply the same collector current. In other

    words, the right half of the circuit can be duplicated several times with various resistor values replacing R2

    on each.

    Note, however, that each additional right-half transistor "steals" a bit of collector current from Q1

    due to the non-zero

    base currents of the right-half transistors. This will result in a small reduction in the programmed current.

    An example of a mirror with emitter degeneration to increase mirror resistance is found in two-port networks.

    For the simple mirror shown in the diagram, typical values of will yield a current match of 1% or better.
  • 7/23/2019 Analog Integrated Circuits


    Current mirror 14

    Figure 2: An n-channel MOSFET current mirror with a

    resistor to set the reference current IREF

    ; VDD

    is the

    supply voltage

    Basic MOSFET current mirror

    The basic current mirror can also be implemented using MOSFET

    transistors, as shown in Figure 2. Transistor M1

    is operating in the

    saturation or active mode, and so is M2. In this setup, the output

    currentIOUT is directly related toIREF, as discussed next.

    The drain current of a MOSFET ID

    is a function of both the

    gate-source voltage and the drain-to-gate voltage of the MOSFET

    given by ID

    = f (VGS

    , VDG

    ), a relationship derived from the

    functionality of the MOSFET device. In the case of transistor M1

    of the mirror, ID

    = IREF

    . Reference current IREF

    is a known

    current, and can be provided by a resistor as shown, or by a

    "threshold-referenced" or "self-biased" current source to ensure

    that it is constant, independent of voltage supply variations.[1]

    Using VDG=0 for transistor M1, the drain current in M1 is ID = f(V


    DG=0), so we find:f (V

    GS, 0) =I

    REF, implicitly determining

    the value of VGS

    . Thus IREF

    sets the value of VGS

    . The circuit in the diagram forces the same VGS

    to apply to

    transistorM2. If M

    2is also biased with zero V

    DGand provided transistors M


    2have good matching of their

    properties, such as channel length, width, threshold voltage etc., the relationshipIOUT

    =f (VGS


    =0 ) applies, thus

    setting IOUT

    = IREF

    ; that is, the output current is the same as the reference current when VDG

    =0 for the output

    transistor, and both transistors are matched.

    The drain-to-source voltage can be expressed as VDS



    . With this substitution, the Shichman-Hodges

    model provides an approximate form for functionf (VGS



    where, is a technology related constant associated with the transistor, W/L is the width to length ratio of the

    transistor, VGS

    is the gate-source voltage, Vth

    is the threshold voltage, is the channel length modulation constant,

    and VDS

    is the drain source voltage.

    Output resistance

    Because of channel-length modulation, the mirror has a finite output (or Norton) resistance given by the ro

    of the

    output transistor, namely (see channel length modulation):


    where = channel-length modulation parameter and VDS

    = drain-to-source bias.

    Compliance voltage

    To keep the output transistor resistance high, VDG

    0 V.[2]

    (see Baker).[3]

    That means the lowest output voltage that

    results in correct mirror behavior, the compliance voltage, is VOUT

    = VCV

    = VGS

    for the output transistor at the output

    current level with VDG

    = 0 V, or using the inverse of thef-function,f1



    For Shichman-Hodges model,f-1

    is approximately a square-root function.
  • 7/23/2019 Analog Integrated Circuits


    Current mirror 15

    Extensions and reservations

    A useful feature of this mirror is the linear dependence of f upon device width W, a proportionality approximately

    satisfied even for models more accurate than the Shichman-Hodges model. Thus, by adjusting the ratio of widths of

    the two transistors, multiples of the reference current can be generated.

    It must be recognized that the Shichman-Hodges model[4]

    is accurate only for rather dated technology, although it

    often is used simply for convenience even today. Any quantitative design based upon new technology uses computermodels for the devices that account for the changed current-voltage characteristics. Among the differences that must

    be accounted for in an accurate design is the failure of the square law in Vgs

    for voltage dependence and the very

    poor modeling of Vds

    drain voltage dependence provided by Vds

    . Another failure of the equations that proves very

    significant is the inaccurate dependence upon the channel lengthL. A significant source ofL-dependence stems from

    , as noted by Gray and Meyer, who also note that usually must be taken from experimental data.[1]

    Feedback assisted current mirror

    Figure 3: Gain-boosted current mirror with op amp feedback to increase output


    Figure 3 shows a mirror using negative

    feedback to increase output resistance.

    Because of the op amp, these circuits are

    sometimes called gain-boosted current

    mirrors. Because they have relatively low

    compliance voltages, they also are called

    wide-swing current mirrors. A variety of

    circuits based upon this idea are in


    particularly for MOSFET

    mirrors because MOSFETs have rather low

    intrinsic output resistance values. A

    MOSFET version of Figure 3 is shown inFigure 4 where MOSFETs M

    3and M


    operate in Ohmic mode to play the same

    role as emitter resistors RE

    in Figure 3, and

    MOSFETs M1

    and M2

    operate in active

    mode in the same roles as mirror transistors


    and Q2

    in Figure 3. An explanation

    follows of how the circuit in Figure 3 works.

    The operational amplifier is fed the

    difference in voltages V1

    - V2

    at the top of

    the two emitter-leg resistors of value RE

    . This difference is amplified by the op amp and fed to the base of output

    transistor Q2. If the collector base reverse bias on Q

    2is increased by increasing the applied voltage V

    A, the current in


    increases, increasing V2

    and decreasing the difference V1

    - V2

    entering the op amp. Consequently, the base

    voltage of Q2

    is decreased, and VBE

    of Q2

    decreases, counteracting the increase in output current.
  • 7/23/2019 Analog Integrated Circuits


    Current mirror 16

    Figure 4: MOSFET version of wide-swing current mirror; M1

    and M2

    are in active

    mode, while M3

    and M4

    are in Ohmic mode and act like resistors

    If the op amp gain Av

    is large, only a very

    small difference V1

    - V2

    is sufficient to

    generate the needed base voltage VB

    for Q2,


    Consequently, the currents in the two leg

    resistors are held nearly the same, and the

    output current of the mirror is very nearly

    the same as the collector current IC1

    in Q1,

    which in turn is set by the reference current


    where 1

    for transistor Q1

    and 2

    for Q2

    differ due to the Early effect if the reversebias across the collector-base of Q



    Figure 5: Small-signal circuit to determine output resistance of mirror; transistor Q2

    is replaced with its hybrid-pi model; a test currentIX

    at the output generates a

    voltage VX

    , and the output resistance isRout

    = VX/I


    Output resistance

    An idealized treatment of output resistance is given in the footnote.[8]

    A small-signal analysis for an op amp with

    finite gainAv

    but otherwise ideal is based upon Figure 5 (, rO

    and r

    refer to Q2). To arrive at Figure 5, notice that

    the positive input of the op amp in Figure 3 is at AC ground, so the voltage input to the op amp is simply the AC

    emitter voltage Ve

    applied to its negative input, resulting in a voltage output of AvV

    e. Using Ohm's law across the

    input resistance r

    determines the small-signal base currentIb


    Combining this result with Ohm's law forRE, Ve can be eliminated, to find:[9]
  • 7/23/2019 Analog Integrated Circuits


    Current mirror 17

    Kirchhoff's voltage law from the test sourceIX

    to the ground ofRE


    Substituting forIb

    and collecting terms the output resistanceRout

    is found to be:

    For a large gainAv>> r

    / R

    Ethe maximum output resistance obtained with this circuit is

    a substantial improvement over the basic mirror whereRout

    = rO


    The small-signal analysis of the MOSFET circuit of Figure 4 is obtained from the bipolar analysis by setting = gm


    in the formula forRout

    and then letting r

    . The result is

    This time, RE is the resistance of the source-leg MOSFETs M3, M4. Unlike Figure 3, however, as Av is increased(holdingR

    Efixed in value),R

    outcontinues to increase, and does not approach a limiting value at largeA


    Compliance voltage

    For Figure 3, a large op amp gain achieves the maximumRout

    with only a small RE

    . A low value for RE

    means V2

    also is small, allowing a low compliance voltage for this mirror, only a voltage V2

    larger than the compliance voltage

    of the simple bipolar mirror. For this reason this type of mirror also is called a wide-swing current mirror, because it

    allows the output voltage to swing low compared to other types of mirror that achieve a large Rout

    only at the

    expense of large compliance voltages.

    With the MOSFET circuit of Figure 4, like the circuit in Figure 3, the larger the op amp gain Av, the smallerR


    be made at a givenRout, and the lower the compliance voltage of the mirror.

    Other current mirrors

    There are many sophisticated current mirrors that have higher output resistances than the basic mirror (more closely

    approach an ideal mirror with current output independent of output voltage) and produce currents less sensitive to

    temperature and device parameter variations and to circuit voltage fluctuations. These multi-transistor mirror circuits

    are used both with bipolar and MOS transistors. These circuits include:

    the Widlar current source

    the Wilson current source

    the Cascoded current sources
  • 7/23/2019 Analog Integrated Circuits


    Current mirror 18


    [1] Paul R. Gray, Paul J. Hurst, Stephen H. Lewis, Robert G. Meyer (2001).Analysis and Design of Analog Integrated Circuits(Fourth Edition

    ed.). New York: Wiley. p. 308309. ISBN 0471321680.

    [2] Keeping the output resistance high means more than keeping the MOSFET in active mode, because the output resistance of real MOSFETs

    only begins to increase on entry into the active region, then rising to become close to maximum value only when VDG

    0 V.

    [3] R. Jacob Baker (2010). CMOS Circuit Design, Layout and Simulation(Third ed.). New York: Wiley-IEEE. pp. 297, 9.2.1 and Figure 20.28,

    p. 636. ISBN 978-0-470-88132-3.[4] NanoDotTek Report NDT14-08-2007, 12 August 2007 (http://www.nanodottek. com/NDT14_08_2007. pdf)

    [5] R. Jacob Baker. 20.2.4 pp. 645646. ISBN 978-0-470-88132-3.

    [6] Ivanov VI and Filanovksy IM (2004). Operational amplifier speed and accuracy improvement: analog circuit design with structural

    methodology(http://books. google. com/books?id=IuLsny9wKIIC& pg=PA110&dq=gain+boost+wide++"current+mirror"#PPA107,M1)

    (The Kluwer international series in engineering and computer science, v. 763 ed.). Boston, Mass.: Kluwer Academic. p. 6.1, p. 105108.

    ISBN 1-4020-7772-6. .

    [7] W. M. C. Sansen (2006).Analog design essentials. New York ; Berlin: Springer. p. 0310, p. 93. ISBN 0-387-25746-2.

    [8] An idealized version of the argument in the text, valid for infinite op amp gain, is as follows. If the op amp is replaced by a nullor, voltage V2

    = V1, so the currents in the leg resistors are held at the same value. That means the emitter currents of the transistors are the same. If the V



    increases, so does the output transistor because of the Early effect: = 0

    ( 1 + VCB

    / VA

    ). Consequently the base current to Q2

    given by


    =IE/ ( + 1) decreases and the output currentI


    E/ (1 + 1 / ) increases slightly because increases slightly. Doing the math,

    where the transistor output resistance is given by rO

    = ( VA

    + VCB

    ) /Iout

    . That is, the ideal mirror resistance for the

    circuit using an ideal op amp nullor isRout

    = ( + 1 ) rO

    , in agreement with the value given later in the text when the

    gain .

    [9] Notice that asAv , V

    e 0 andI




    External links Patent (1968) by RJ Widlar:Biasing scheme especially suited for integrated circuits(

    patents?hl=en&id=IhlMAAAAEBAJ&dq=+"Biasing+scheme+especially+suited+for+ integrated+circuits"&



    Current mirrors (


    gs_upl=1781l6281l0l14l14l0l3l3l0l328l2455l0.4.6. 1)

    4QD tec - Current sources and mirrors ( html) Compendium of circuits and

  • 7/23/2019 Analog Integrated Circuits


    Differential amplifier 19

    Differential amplifier

    Differential amplifier symbolThe inverting and

    non-inverting inputs are distinguished by "" and

    "+" symbols (respectively) placed in the amplifier

    triangle. Vs+ and Vs are the power supply

    voltages; they are often omitted from the diagram

    for simplicity, but of course must be present in

    the actual circuit.

    A differential amplifier is a type of electronic amplifier that amplifies

    the difference between two voltages but does not amplify the particular



    Many electronic devices use differential amplifiers internally. The

    output of an ideal differential amplifier is given by:

    Where and are the input voltages and is the differential


    In practice, however, the gain is not quite equal for the two inputs. This

    means, for instance, that if and are equal, the output will not

    be zero, as it would be in the ideal case. A more realistic expression for

    the output of a differential amplifier thus includes a second term.

    is called the common-mode gain of the amplifier.

    As differential amplifiers are often used to null out noise or bias-voltages that appear at both inputs, a low

    common-mode gain is usually desired.

    The common-mode rejection ratio (CMRR), usually defined as the ratio between differential-mode gain and

    common-mode gain, indicates the ability of the amplifier to accurately cancel voltages that are common to bothinputs. The common-mode rejection ratio is defined as:

    In a perfectly symmetrical differential amplifier, is zero and the CMRR is infinite. Note that a differential

    amplifier is a more general form of amplifier than one with a single input; by grounding one input of a differential

    amplifier, a single-ended amplifier results.

    Long-tailed pair

    Historical background

    The long-tailed pair was originally implemented using a pair of vacuum tubes. The circuit works the same way for

    all three-terminal devices with current gain. Today, its main feature is mostly vestigial, by virtue of the fact that

    long-tail resistor circuit bias points are largely determined by Ohm's Law and less so by active component


    The long-tailed pair was developed from earlier knowledge of push-pull circuit techniques and measurement


    The earliest circuit that is truly recognizable as a long-tailed pair in its conventional form is given by

    Matthews (1934)[2]

    and the same circuit form appears in a patent submitted by Alan Blumlein in 1936.[3]

    By the end

    of the 1930s the topology was well established and had been described by various authors including Offner (1937),[4]

    Schmitt (1937)[5]

    and Toennies (1938) and It was particularly used for detection and measurement of physiologicalimpulses.

  • 7/23/2019 Analog Integrated Circuits


    Differential amplifier 20

    The long-tailed pair was very successfully used in early British computing, most notably the Pilot ACE Model and


    Wilkes' EDSAC, and probably others designed by people who worked with Blumlein or his peers.

    The long-tailed pair has many attributes as a switch: largely immune to tube (transistor) variations (of great

    importance when machines contained 1,000 or more tubes), high gain, gain stability, high input impedance,

    medium/low output impedance, good clipper (with not-too-long tail), non-inverting (EDSAC contained no

    inverters!) and large output voltage swings. One disadvantage is that the output voltage swing (typically 1020 V)

    was imposed upon a high DC voltage (200 V or so), requiring care in signal coupling, usually some form of

    wide-band DC coupling. Many computers of this time tried to avoid this problem by using only AC-coupled pulse

    logic, which made them very large and overly complex (ENIAC: 18,000 tubes for a 20 digit calculator) or unreliable.

    DC-coupled circuitry became the norm after the first generation of vacuum tube computers.


    A differential (long-tailed,[8]

    emitter-coupled) pair amplifier consists of two amplifying stages with common

    (emitter, source or cathode) degeneration.

    Differential output

    Figure 2: A classic long-tailed pair

    With two inputs and two outputs, this forms a differential amplifier

    stage (Fig. 2). The two bases (or grids or gates) are inputs which are

    differentially amplified (subtracted and multiplied) by the pair; they

    can be fed with a differential (balanced) input signal, or one input

    could be grounded to form a phase splitter circuit. An amplifier with

    differential output can drive floating load or another stage with

    differential input.

    Single-ended output

    If the differential output is not desired, then only one output can be

    used (taken from just one of the collectors (or anodes or drains),

    disregarding the other output without a collector resistor; this

    configuration is referred to as single-ended output. The gain is half that

    of the stage with differential output. To avoid sacrificing gain, a

    differential to single-ended converter can be utilized. This is often

    implemented as a current mirror (Fig. 3).


    To explain the circuit operation, four particular modes are isolated below although, in practice, some of them act

    simultaneously and their effects are superimposed.


    In contrast with classic amplifying stages that are biased from the side of the base (and so they are highly

    -dependent), the differential pair is directly biased from the side of the emitters by sinking/injecting the total

    quiescent current. The series negative feedback (the emitter degeneration) makes the transistors act as voltage

    stabilizers; it forces them to adjust their VBE

    voltages (base currents) so that to pass the quiescent current through

    their collector-emitter junctions.[9]

    So, due to the negative feedback, the quiescent current depends slightly on the

    transistor's .
  • 7/23/2019 Analog Integrated Circuits


    Differential amplifier 21

    The biasing base currents needed to evoke the quiescent collector currents usually come from the ground, pass

    through the input sources and enter the bases. So, the sources have to be galvanic (DC) to ensure paths for the

    biasing currents and low resistive enough to not create significant voltage drops across them. Otherwise, additional

    DC elements should be connected between the bases and the ground (or the positive power supply).

    Common mode

    At common mode (the two input voltages change in the same directions), the two voltage (emitter) followers

    cooperate with each other working together on the common high-resistive emitter load (the "long tail"). They all

    together increase or decrease the voltage of the common emitter point (figuratively speaking, they together "pull up"

    or "loose" it so that it moves). In addition, the dynamic load "helps" them by changing its instant ohmic resistance in

    the same direction as the input voltages (it increases when the voltage increases and vice versa.) thus keeping up

    constant total resistance between the two supply rails. There is a full (100%) negative feedback; the two input base

    voltages and the emitter voltage change simultaneously while the collector currents and the total current do not

    change. As a result, the output collector voltages do not change as well.

    Differential mode

    Normal. At differential mode (the two input voltages change in opposite directions), the two voltage (emitter)

    followers oppose each other - while one of them tries to increase the voltage of the common emitter point, the other

    tries to decrease it (figuratively speaking, one of them "pulls up" the common point while the other "looses" it so that

    it stays immovable) and v.v. So, the common point does not change its voltage; it behaves like a virtual ground with

    a magnitude determined by the common-mode input voltages. The high-resistive emitter element does not play any

    role since it is shunted by the other low-resistive emitter follower. There is no negative feedback since the emitter

    voltage does not change at all when the input base voltages change. he common quiescent current vigorously steers

    between the two transistors and the output collector voltages vigorously change. The two transistors mutually ground

    their emitters; so, although they are common-collector stages, they actually act as common-emitter stages with

    maximum gain. Bias stability and independence from variations in device parameters can be improved by negativefeedback introduced via cathode/emitter resistors with relatively small resistances.

    Overdriven. If the input differential voltage changes significantly (more than about a hundred millivolts), the

    base-emitter junction of the transistor driven by the lower input voltage becomes backward biased and its collector

    voltage reaches the positive supply rail. The other transistor (driven by the higher input voltage) saturates and its

    collector voltage begins following the input one. This mode is used in differential switches and ECL gates.

    Breakdown. If the input voltage continues increasing and exceeds the base-emitter breakdown voltage, the

    base-emitter junction of the transistor driven by the lower input voltage breaks down. If the input sources are low

    resistive, an unlimited current will flow directly through the "diode bridge" between the two input sources and will

    damage them.

    At common mode, the emitter voltage follows the input voltage variations; there is a full negative feedback and the

    gain is minimum. At differential mode, the emitter voltage is fixed (equal to the instant common input voltage); there

    is no negative feedback and the gain is maximum.
  • 7/23/2019 Analog Integrated Circuits


    Differential amplifier 22

    Single-ended input

    The differential pair can be used as an amplifier with a single-ended input if one of the inputs is grounded or fixed to

    a reference voltage (usually, the other collector is used as a single-ended output) This arrangement can be thought as

    of cascaded common-collector and common-base stages or as a buffered common-base stage.[10]

    The emitter-coupled amplifier is compensated for temperature drifts, VBE

    is cancelled, Miller effect and transistor

    saturation are beated. That is why it is used to form emitter-coupled amplifiers (avoiding Miller effect), phase splittercircuits (obtaining two inverse voltages), ECL gates and switches (avoiding transistor saturation), etc.


    Emitter constant current source

    Figure 3: An improved long-tailed pair with

    current-mirror load and constant-current biasing

    The quiescent current has to be constant to ensure constant collector

    voltages at common mode. This requirement is not so important in the

    case of a differential output since the two collector voltages will vary

    simultaneously but their difference (the output voltage) will not vary;

    only the output range will decrease. But in the case of a single-ended

    output, it is extremely important to keep a constant current since the

    output collector voltage will vary. Thus the higher the resistance of the

    current source , the lower is, and the better the CMRR. The

    constant current needed can be produced by connecting an element

    (resistor) with very high resistance between the shared emitter node

    and the supply rail (negative for NPN and positive for PNP transistors)

    but this will require high supply voltage. That is why, in more

    sophisticated designs, an element with high differential (dynamic)

    resistance approximating a constant current source/sink is substituted for the long tail (Fig. 3). It is usually

    implemented by a current mirror because of its high compliance voltage (small voltage drop across the output


    The same arrangement is widely used in cascode circuits as well. It can be generalized by an equivalent circuit

    consisting of a constant current source loaded by two connected in parallel voltage sources with equal voltages. The

    current source determines the common current flowing through the voltage sources while the voltage sources fix the

    voltage across the current source. The emitter current source is usually implemented as a common-emitter transistor

    stage with constant base voltage driving with current the two common-base transistor stages. So, this arrangement

    can be considered as a cascode consisting of cascaded common-emitter and common-base stages.

    Collector current mirrorThe collector resistors can be replaced by a current mirror, whose output part acts as an active load (Fig. 3). Thus the

    differential collector current signal is converted to a single ended voltage signal without the intrinsic 50% losses and

    the gain is extremely increased. This is achieved by copying the input collector current from the right to the left side

    where the magnitudes of the two input signals add. For this purpose, the input of the current mirror is connected to

    the right output and the output of the current mirror is connected to the left output of the differential amplifier.

    The current mirror inverts the right collector current and tries to pass it through the left transistor that produces the

    left collector current. In the middle point between the two left transistors, the two signal currents (current changes)

    are subtracted. In this case (differential input signal), they are equal and opposite. Thus, the difference is twice the

    individual signal currents (I - (-I) = 2I) and the differential to single ended conversion is completed without gain

  • 7/23/2019 Analog Integrated Circuits


    Differential amplifier 23

    Interfacing considerations

    Floating input source

    It is possible to connect a floating source between the two bases, but it is necessary to ensure paths for the biasing

    base currents. In the case of galvanic source, only one resistor has to be connected between one of the bases and the

    ground. The biasing current will enter directly this base and indirectly (through the input source) the other one. If the

    source is capacitive, two resistors have to be connected between the two bases and the ground to ensure different

    paths for the base currents.

    Input/output impedance

    The input impedance of the differential pair highly depends on the input mode. At common mode, the two parts

    behave as common-collector stages with high emitter loads; so, the input impedances are extremely high. At

    differential mode, they behave as common-emitter stages with grounded emitters; so, the input impedances are low.

    The output impedance of the differential pair is high (especially for the improved differential pair from Fig. 3).

    Input/output range

    The common-mode input voltage can vary between the two supply rails but cannot closely reach them since some

    voltage drops (minimum 1 volt) have to remain across the output transistors of the two current mirrors.

    Other differential amplifiers

    Figure 4: Op-amp differential amplifier

    An operational amplifier, or op-amp, is a

    differential amplifier with very high

    differential-mode gain, very high input

    impedances, and a low output impedance. By

    applying negative feedback an op-ampdifferential amplifier (Fig. 4) with predictable

    and stable gain can be built.[11]

    Some kinds of

    differential amplifier usually include several

    simpler differential amplifiers. For example, an

    instrumentation amplifier, a fully differential

    amplifier, an instrument amplifier, or an

    isolation amplifier are often built from several



    Differential amplifiers are found in many circuits that utilise series negative feedback (op-amp follower,

    non-inverting amplifier, etc.), where one input is used for the input signal, the other for the feedback signal (usually

    implemented by operational amplifiers). For comparison, the old-fashioned inverting single-ended op-amps from the

    early 40's could realize only parallel negative feedback by connecting additional resistor networks (an op-amp

    inverting amplifier is the most popular example). A common application is for the control of motors or servos, as

    well as for signal amplification applications. In discrete electronics, a common arrangement for implementing a

    differential amplifier is the long-tailed pair, which is also usually found as the differential element in most op-amp

    integrated circuits. A long-tailed pair can be used as an analog multiplier with the differential voltage as one input

    and the biasing current as another.
  • 7/23/2019 Analog Integrated Circuits


    Differential amplifier 24

    A differential amplifier is used as the input stage emitter coupled logic gates and as switch. When used as a switch,

    the "left" base/grid is used as signal input and the "right" base/grid is grounded; output is taken from the right

    collector/plate. When the input is zero or negative, the output is close to zero (but can be not saturated); when the

    input is positive, the output is most-positive, dynamic operation being the same as the amplifier use described above.

    Footnotes[1] Eglin, J. M.A Direct Current Amplifier for Measuring Small Currents. Journal of the Optical Society of America and Review of Scientific

    Instruments, May 1929, p.393-402.

    [2] Matthews, B. H. C.A Special Purpose Amplifier. The Journal of Physiology, March 17, 1934, 81, 28P-29P.

    [3][3] Blumlein, A. D. UK Patent No: 482740; July 4, 1936. US Patent 2185367; filed June 24, 1937

    [4] Offner, F.Push-Pull Resistance Coupled Amplifiers. Review of Scientific Instruments, January 1937, p.20-21.

    [5] Schmitt, O. H. Cathode Phase Inversion. Review of Scientific Instruments, 1937, p.100-101.

    [6] Geddes, L. A. Who Invented the Differential Amplifier?. IEEE Engineering in Medicine and Biology, May/June 1996, p.116-117.

    [7][7] Details of the long-tailed pair circuitry used in early computing can be found in "Alan Turing's Automatic Computing Engine" (Oxford

    University Press, 2005, ISBN 0-19-856593-3) in Part IV, 'ELECTRONICS'

    [8] Long-tail is a figurative name of high resistance that represents the high emitter resistance at common mode with a common long tail with a

    proportional length (at differential mode this tail shortens up to zero). If additional emitter resistors with small resistances are included

    between the emitters and the common node (to introduce a small negative feedback at differential mode), they can be figuratively representedby short tails.

    [9][9] It is interesting fact that the negative feedback as though has reversed the transistor behavior - the collector current has become an input

    quantity while the base current serves as an output one.

    [10][10] More generally, this arrangement can be considered as two interacting voltage followers with negative feedback: the output part of the

    differential pair acts as a voltage follower with constant input voltage (a voltage stabilizer) producing constant output voltage; the input part

    acts as a voltage follower with varying input voltage trying to change the steady output voltage of the stabilizer. The stabilizer reacts to this

    intervention by changing its output quantity (current, respectively voltage) that serves as a circuit output.

    [11] It seems strange that, in this arrangement, a high-gain differential amplifier (op-amp) is used as a component to build a low-gain differential

    amplifier like a high-gain inverting amplifier (op-amp) serves as a component in a low-gain inverting amplifier; but just this paradox of

    negative feedback amplifiers has impeded Harold Black to obtain his patent.


    External links

    BJT Differential Amplifier (

    Circuit and explanation

    A testbench for differential circuits (

    A discrete OpAmp with complimentary differential amplifier (

    KompDiffAmp.html) (in German)

    Application Note: Terminating a Differential Amplifier in Single-Ended Input Applications (http://www.analog.





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    Operational amplifier 25

    Operational amplifier

    A Signetics a741 operational amplifier, one of the most successful op-amps.

    An operational amplifier ("op-amp") is a

    DC-coupled high-gain electronic voltage

    amplifier with a differential input and,

    usually, a single-ended output.[1] An op-amp

    produces an output voltage that is typically

    hundreds of thousands times larger than the

    voltage difference between its input


    Operational amplifiers are important

    building blocks for a wide range of

    electronic circuits. They had their origins in

    analog computers where they were used in

    many linear, non-linear andfrequency-dependent circuits. Their

    popularity in circuit design largely stems from the fact that characteristics of the final op-amp circuits with negative

    feedback (such as their gain) are set by external components with little dependence on temperature changes and

    manufacturing variations in the op-amp itself.

    Op-amps are among the most widely used electronic devices today, being used in a vast array of consumer,

    industrial, and scientific devices. Many standard IC op-amps cost only a few cents in moderate production volume;

    however some integrated or hybrid operational amplifiers with special performance specifications may cost over

    $100 US in small quantities. Op-amps may be packaged as components, or used as elements of more complex

    integrated circuits.

    The op-amp is one type of differential amplifier. Other types of differential amplifier include the fully differential

    amplifier (similar to the op-amp, but with two outputs), the instrumentation amplifier (usually built from three

    op-amps), the isolation amplifier (similar to the instrumentation amplifier, but with tolerance to common-mode

    voltages that would destroy an ordinary op-amp), and negative feedback amplifier (usually built from one or more

    op-amps and a resistive feedback network).

    Circuit notation

    Circuit diagram symbol for an op-amp

    The circuit symbol for an op-amp is shown to the right, where:

    V+: non-inverting input

    V: inverting input


    : output


    : positive power supply


    : negative power supply

    The power supply pins (VS+

    and VS

    ) can be labeled in different ways

    (See IC power supply pins). Despite different labeling, the function

    remains the same to provide additional power for amplification of

    the signal. Often these pins are left out of the diagram for clarity, and

    the power configuration is described or assumed from the circuit.
  • 7/23/2019 Analog Integrated Circuits


    Operational amplifier 26


    An op-amp without negative feedback (a


    The amplifier's differential inputs consist of a V+

    input and a V


    and ideally the op-amp amplifies only the difference in voltage

    between the two, which is called the differential input voltage. The

    output voltage of the op-amp is given by the equation,

    where V+

    is the voltage at the non-inverting terminal, V

    is the voltage

    at the inverting terminal andAOL

    is the open-loop gain of the amplifier

    (the term "open-loop" refers to the absence of a feedback loop from the

    output to the input).

    The magnitude of AOL

    is typically very large10,000 or more for

    integrated circuit op-ampsand therefore even a quite small difference

    between V+

    and V

    drives the amplifier output nearly to the supply

    voltage. This is called saturation of the amplifier. The magnitude of

    AOL is not well controlled by the manufacturing process, and so it is impractical to use an operational amplifier as a

    stand-alone differential amplifier. Without negative feedback, and perhaps with positive feedback for regeneration,

    an op-amp acts as a comparator. If the inverting input is held at ground (0 V) directly or by a re