AM Stereo Tuner

AM Stereo Tuner/FM Stereo IF Receiver Written by BuSan    Wednesday, 08 August 2007 This application combines a Sanyo LA1832M with the Motorola MC13028A AM Stereo decoder IC. The LA1832 provides an FM IF, FM multiplex detection, AM tuning, and the AM IF functions. The MC13028A provides the AM Stereo detection as well as Left and Right audio outputs. An MC145151 synthesizer provides the frequency control of the local oscillator contained within the LA1832. Frequency selection is by means of a switch array attached to the synthesizer.

The MC13028A is designed as a low voltage, low cost decoder for the C–QUAM AM Stereo technology and is completely compatible with existing monaural AM transmissions. The IC requires relatively few, inexpensive external parts to produce a full featured C–QUAM AM Stereo implementation. The layout is straightforward and should produce excellent stereo performance. This device performs the function of IF amplification, AGC, modulation detection, pilot tone detection, signal quality inspection, and left and right audio output matrix operation. The IC is targeted for use in portable and home AM Stereo radio applications.

The amplified C–QUAM IF signal is fed simultaneously to the envelope detector circuit, and to a C–QUAM converter circuit. The envelope detector provides the L+R (mono) signal output which is fed to the stereo matrix. In the converter circuit, the C–QUAM signal is restored to a Quam signal. This is accomplished by dividing the C–QUAM IF signal by the demodulated cos Φ term. The cosΦ term is derived from the phase modulated IF signal in an active feedback loop. Cosine Φ is detected by comparing the envelope detector and the in–phase detector outputs in the high speed comparator/feedback loop. Cosine Φ is extracted from the I detector output and is actively transferred through feedback to the output of the comparator. The output of the comparator is in turn fed to the control input of the divider, thus closing the feedback loop of the converter circuit. In this process, the cos Φ term is removed from the divider IF output, thus allowing direct detection of the L–R by the quadrature detector. The audio outputs from both the envelope and the L–R detectors are first filtered to minimize the second harmonic of the IF signal. Then they are fed into a matrix circuit where the Left channel and the Right channel outputs can be extracted at Pins 15 and 16. (The outputs from the I and Q detectors are also filtered similarly.) At this time, a stereo indicator driver circuit, which can sink up to 10 mA, is also enabled. The stereo output will occur if the input IF signal is: larger than the stereo threshold level, not too noisy, and if a proper pilot tone is present. If these three conditions are not met, the blend circuit will begin to force monaural operation at that time.

A blend circuit is included in this design because conditions occur during field use that can cause input signal strength fluctuation, strong unwanted co–channel or power line interference, and/or multi–path or re–radiation. When these aberrant conditions occur, rapid switching between stereo and mono might occur, or the stereo quality might be degraded enough to sound displeasing. Since these conditions could be annoying to the normal listener, the stereo information is blended towards a monaural output. This circuit

action creates a condition for listening where these aberrant effects are better tolerated by the consumer.

Intentional mono operation is a feature sometimes required in receiver designs. There are several ways in which to accomplish this feat. First, a resistor from Pin 10 to ground can be switched into the circuit. A value of 1.0 k is adequate as is shown in the schematic in Figure 18. A second method to force the decoder into mono is simply to shunt Pin 10 to ground through an NPN transistor (collector to Pin 10, emitter to ground), where the base lead is held electrically “high” to initiate the action.

A third method to force a mono condition upon the decoder is to shunt Pin 8 of the decoder to ground through an NPN transistor as described above. Effectively, this operation discharges the blend capacitor (10 μF), and the blend function takes over internally forcing the decoder into mono. This third method does not necessarily require extra specific parts for the forced mono function as the first two examples do. The    reason for this is that most electronically tuned receiver designs require an audio muting function during turn on/turn off, tuning/scanning, or band switching (FM to AM). When the muting function is designed into an AM Stereo receiver, it also should include a blend capacitor reset (discharge) function which is accomplished in this case by the use of an NPN transistor shunting Pin 8 to ground, (thus making the addition of a forced mono function almost “free”). The purpose of the blend reset during muting is to re–initialize the decoder back into the “fast lock” mode from which stereo operation can be attained much quicker after any of the interruptive activities mentioned earlier, (i.e. turn on, tuning, etc.).

The VCO in this IC is a phase shift oscillator type design that operates with a ceramic resonator at eight times the IF frequency, or 3.60 MHz. With IF input levels below the stereo threshold level, the oscillator is not operational. This feature helps to eliminate audio tweets under low level, noisy input conditions.

The following information provides circuit function, part number, and the manufacturer’s name for special parts identified by their schematic symbol. Where the part is not limited to a single source, a description sufficient to select a part is given.U1     IC – AM Stereo Decoder         MC13028AD by MotorolaU2     IC – AM/FM IF and Multiplex Tuner         LA1832M by SanyoU3     IC – Frequency Synthesizer         MC145151DW2 by MotorolaT1     AM IF Coil         A7NRES–11148N by TOKOF1     AM IF Ceramic Filter         SFG450F by MurataF2     FM IF Detector Resonator         CDA10.7MG46A by Murata

F3     FM Multiplex Decoder Resonator         CSB456F15 by MurataF4     AM Tuner Block         BL–70 by Korin GikenX1     10.24 MHz Crystal, Fundamental Mode,         AT Cut, 18 pF Load Cap, 35 Ω maximum series R.         HC–18/U HolderX2     3.6 MHz AM Stereo Decoder Resonator         CSA3.60MGF108 by MurataS5     8 SPST DIP Switch

Figure 1. AM STEREO TUNER/FM STEREO IF Circuit with MC13028, MC145151 and LA1832

Figure 2. AM STEREO TUNER/FM STEREO IF PCB with MC13028, MC145151 and LA1832

The LA1832 tuner IC (U2) is set for AM operation by switch S2 connecting Pin 12 to ground. An AM Stereo signal source is applied to Pin 2 of the RF coil contained within the BL–70 tuning block. That coil applies the signal to Pin 21 of U2. The L.O. coil is connected from Pin 23 to VCC. The secondary is tuned by a varactor which is controlled by a dc voltage output from the synthesizer circuit. The reactance of this oscillator tank is coupled back to Pin 23. It is through this reactance that the frequency of the L.O. is determined. A buffered output from the L.O. emerges at Pin 24. This signal is routed to Pin 1 of the synthesizer (U3), thus completing the frequency control loop.

The mixer output at Pin 2 is applied to the IF coil T1. Coil T1 provides the correct impedance to drive the ceramic bandpass filter F1. The IF signal returns to U2 through Pin 4, and also to the input, Pin 4 of the AM Stereo decoder (U1). The ceramic filter F1 is designed to operate into a load resistance of 2.0 kΩ. This load is provided at Pin 4 of U2.

The stereo outputs exit from Pins 15 and 16 of U1. The design amplitudes of the audio outputs will vary according to the values used for the resistors to ground at Pins 15 and 16 of the decoder, While the values chosen for RO are left to the discretion of the designer, the numbers chosen in this data sheet are reflective of those required to set the general industry standard levels of audio outputs in receiver designs.

Pins 15 and 16 are also good locations for the insertion of simple RC filters that are used to comply with the United States NRSC requirement for the shape of the overall receiver audio response. There are many design factors that affect the shape of the receiver response, and they must all be considered when trying to approximate the NRSC de–emphasis response. The mixer output transformer (IF coil, T1), and ceramic filter probably have the greatest contribution to the frequency response. The ceramic filter can be tailored from its rated response by the choice of transformer impedance and bandwidth. When designing an overall audio response shape, the response of the speakers or earphones should also be considered.

The frequency to which the test circuit will tune is set by the eight binary switches contained in the S5 assembly, numbered from 1 to 8. Number 1 connects to Pin 11 of U3 and number 8 connects to Pin 18. The other switches connect to the pins in between and in order. Each individual switch is a SPST type.

To tune to a specific RF frequency, a computation must be made in order to ascertain the divide ratio to input to the synthesizer via the switch array. The divide ratio is simply the eight digit binary equivalent number for the local oscillator frequency divided by 10 kHz. The local oscillator frequency is the desired RF frequency plus 450 kHz, the IF frequency. Any local oscillator value within the AM band can be representedby a binary number. Each binary bit represents a switch setting where a “1” is an open switch and a “0” is a closed switch. The most significant bit represents switch 8 which is connected to Pin 18.

To illustrate, consider the setting for an input frequency of 1070 kHz. (This frequency was used to test the circuit board as described further on.) The local oscillator frequency is 1070 kHz plus 450 kHz which equals 1520 kHz. Dividing by 10 kHz yields the number 152. The binary number for 152 is 10011000. Thus the switches are set to:

Switch Configuration

The MC13025 is the complementary ETR® Electronically Tuned Radio front–end for the second generation MC13022 C–QUAM® AM stereo IF and decoder. The MC13025 provides a high dynamic range mixer, voltage controlled oscillator, and first IF that with the MC13022 and synthesizer form a complete digitally controlled AM stereo tuner system. This system in turn may drive a dual channel audio processor and high power amplifiers for car radio or home stereo applications. Other applications include portable

radio “boom boxes”, table radios and component stereo systems. Functionally, the MC13022A and MC13022 are very similar. The MC13022A has 10 dB more audio output and a CMOS compatible logic level output (Pin 15) for stop sense. The stop sense/AGC function has been internally connected to the output notch filter control.

Figure 1. MC13025 and MC13022 Cascode RF ETR AM Stereo Receiver Application Circuit

Blend and Noise Reduction

Although AM stereo does not have the extreme difference in S/N between mono and stereo that FM does (typically less than 3.0 dB versus greater than 20 dB for FM), sudden switching between mono and stereo is quite apparent. Some forms of interference such as co–channel have a large L–R component that makes them more annoying than would ordinarily be expected for the measured level. The MC13022A measures the interference level and reduces L–R as interference increases, blending smoothly to monaural. The pilot indicator remains on as long as a pilot signal is detected, even when interference is severe, to minimize annoying pilot light flickering.

Signal Strength

A dc voltage proportional to the log of signal strength is provided at Pin 6 MC13022A. This can be used for signal strength indication, and it directly controls the post detection filter. Normal operation is above 2.2 V as shown is Figure NO TAG.

Stop Sense

The signal strength information is multiplexed with the stop sense signal. The stop sense is activated when scanning by externally pulling the blend, Pin 23 MC13022A, below 0.3 V. This would typically be done from the mute line in a frequency synthesizer.

If at any time Pin 23 is low and there is either no signal in the IF or a noisy signal of a predetermined interference level, Pins 6 and 15 will go low. This low can be used to tell the frequency synthesizer to immediately scan to the next channel. The interference detection prevents stopping on many unlistenable stations, a feature particularly useful at night when many frequencies may have strong signals from multiple co–channel stations. Pin 6 drives a comparator which has a 1.7 V reference. Therefore the comparator output, Pin 15, is low if Pin 6 is <1.7 V and high if Pin 6 is >1.7 V.

IF Bandwidth Control

IF AGC attenuates the signal by shunting the signal at the IF input. This widens the IF bandwidth by decreasing the loaded Q of the input coupling coil as signal strength increases.

Post Detection Filtering

With weak, noisy signals, high frequency rolloff greatly improves the sound. Conventional tone controls do not attenuate the highs sufficiently to control noise without also significantly affecting the mid–range. Also, notch filters are necessary with any wide–band AM radio to eliminate the 10 kHz whistle from adjacent stations.

By using a twin–T filter with variable feedback to the normally grounded center leg, a variable Q notch filter is formed that provides both the 10 kHz notch and variable high frequency rolloff functions. Typical range of response is shown in Figure NO TAG. Response is controlled by Pin 6 MC13022A for automatic audio bandwidth control as a function of signal strength.

Supply Voltage

Ranger= +10V.....+18VSupply Current whit LED= 30 mA

Stereo Separation= 45 dBT.H.D at 0,5Vrms-1KHZ = 0,4%

Input Resistance= 45 KOutput Resistance= 1,3K

SCA Rejection= 70 dB  LM1800N: See Database

SpecificationPower supply =8V-14V

Current with out LED =13 mAInput signal = 1V RMS.Output signal= 485 mV

Distortion = 0,3%Channel Separation= 40dBMC1310P: See Database

FM Receiver Written by BuSan    Friday, 17 November 2006 An FM waveform carries its information in the form of frequency, so the amplitude is constant. Thus the information is held in the zero crossings. The FM waveform can be clipped at a low level without the loss of information. Additive noise has less of an effect on zero crossings than the amplitude. Receivers therefore often clip, or limit the amplitude of the received waveform prior to frequency detection. This produces a constant waveform as an input to the discriminator. This clipping has the effect of introducing higher harmonic terms which are rejected by a pos-detection low-pass filter. A simplified FM receiver is shown in figure 1a, a more sophisticated system is shown in figure 1b.

DEMODULATORS

FM demodulators are frequency-dependant circuits that produce an output voltage that is directly proportional to the instantaneous frequency at its input. The signal received is lfm(t) and is known to the receiver in the form,

Several circuits are used for demodulating FM signals slope detector, Foster-Seeley disciminator, ratio detector, PLL demodulator and quadrature detector. The first three are tuned circuit frequency discriminators, they work by converting the FM signal to AM then demodulate using conventional peak detectors.

Descriminators

A block diagram of a descriminator is shown in figure 2.

The differentiator effectively converts the FM signal into an AM signal. The differentiated FM signal is,

The envelope detector removes the sine term, this is possible because the slight changes in frequency are not detected by the envelope detector. The envelope is given by

from which the signal s(t) can be found.

When a differentiator is used like this it is called a slope detector or discriminator. A requirement for a descriminator is that the transfer function be linear throughout the range of frequencies of the FM wave. This is the simplest type of decriminator. Two descriminators can be used by subtracting the characteristic of one from a shifted version of itself, see figure 3.

This method is called a balanced slope detector. It has several disadvantages like poor linearity and difficulty in tuning. Another way is to approximate the derivative by using the difference between two adjacent sample values of the waveform, see figure 4, the Foster-Seeley discriminator or also known as a phase shift demodulator.

The Foster-Seeley circuit is easier to tune but must be preceeded by a separate limiter circuit to clip the amplitude before demodulating. The ratio detector has the property that

it is immune to amplitude variations in its input signal, so a preceeding limiter is not required. Last Updated ( Friday, 08 December 2006 ) FM Stereo Written by BuSan    Friday, 17 November 2006 Until 1961, all commercial FM broadcast-band transmissions were monophonic. The FCC

authorized stereophonic transmission for the commercial FM broadcast band. With stereophonic transmission, the information signal is spatially divided into two 50-Hz to 15-kHz audio channels (a left and a right). Music that originated on the left side is reproduced only on the left speaker, and music that originated on the right side is reproduced only on the right speaker. The main problem with introducing stereophonic transmission is the compatibility with monophonic receivers.

The spectrum shown in figure 1 is the standard spectrum used today. The spectrum compromises the 50 Hz to 15 kHzstereo channel plus an additional stereo channel frequency division multiplexed into a composite baseband signal with a 19 kHz pilot. The three channels are (1) the left (L) plus the right (R) audio channels, (2) the left plus the inverted right audio channels, and (3) the SCA subcarrier and its associated sidebands.

The L + R stereo channel occupies the 0 - 15 kHz passband. The L - R audio channel amplitude modulates a 38 kHz subcarrier and produces the L - R stereo channel, which is a double-sideband suppressed carrier that occupies the 23 - 53 kHz passband, used only for FM stereo transmission. SCA transmission occupy the 60 - 74 kHz frequency spectrum. Mono receivers can demodulate the total baseband spectrum but only the 50 - 15 kHz L + R channel is amplified and fed to all speakers. Stereophonic receivers must provide additional demodulation of the 23 - 53 kHz L - R stereo channel, separate the left and right audio channels and then feed them to their respective speakers.

The process of multiplexing two audio signals is shown in figure 2.

Stereo Transmission

Figure 3 shows a simplified block diagram for a stereo FM transmitter. The L and R audio channels are combined in a matrix network to produce the L + R and the L - R audio channels. The L - R audio channel modulates a 38 kHz subcarrier and produces a 23 to 53 kHz L - R stereo channel. Because there is a time delay introduced in the L - R signal path as it propagates through the balanced modulator, the L + R stereo channel must be artificially delayed somewhat to maintain phase integrity with the L - R stereo channel for demodulation purposes. Also for demodulation purposes, a 19 kHz pilot is transmitted rather than the 38 kHz subcarrier because it is considerably more difficult to recover the 38 kHz subcarrier in the receiver. The composite baseband signal is fed to the FM transmitter, where it modulates the main carrier.

Stereo Reception

FM stereo receivers are identical to standard FM receivers up to the output of the audio

detector stage. The output of the discriminator is the total baseband spectrum that was shown in figure 1. Figure 4 shows a simplified block diagram for an FM receiver that has both mono and stereo audio outputs. In the mono section of the signal processor, the L + R stereo channel, which contains all of the original information from both the L and R audio channels, is simply filtered, amplified and the fed to both the L and R speakers. In the stereo section of the signal processor, the baseband signal is fed into the stereo demodulator where the L and R audio channels are separated and then fed to their respective speakers. The L + R and L - R stereo channels and the 19 kHz pilot are separated from the composite baseband signal with filters. The 19 kHz pilot is filtered with a high-Q bandpass filter, multiplied by 2, amplified and the fed to the L - R demodulator. The L + R stereo channel is filtered off by a low-pass filter with an upper cutoff frequency of 15 kHz. The L - R double-sideband signal is separated with a broadly tuned bandpass filter and then mixed with the recovered 38 kHz carrier in a balanced modulator to produce the L - R audio information. The matrix network combines the L + R and L - R signals in such a way as to separate the L and R audio information signals, which are fed to their respective deemphasis networks and speakers.

The block diagram for the stereo matrix decoder if shown in figure 5. The L - R audio channel is added directly to the L + R audio channel. The ouput from the adder is

The L - R audio channel is inverted and then added to the L + R audio channel. The output from the adder is

Frequency Modulation Principles Written by BuSan    Friday, 17 November 2006 Frequency modulation of a carrier signal (with frequency fC) involves changing the frequency of this carrier based on some input signal s(t). The modulated carrier then contains the information in s(t). This process is useful in radio communications because it is not practical to directly transmit audio frequencies through the air.

The modulated carrier is of the form Acos [ (t)] , where (t) is the varying phase angle of the signal. We define the instantaneous frequency of the signal to be

Note that the power of the FM signal is a constant (with value A2).

Figures 1 & 2 show some examples of frequency modulated signals and the corresponding inputs.

To examine the FM signal's spectrum, we will look at the various harmonics comprising the input signal. If the input signal is s(t) = X cos 2 fmt, then the phase angle is

and the FM signal is

We define

as the modulation index of the FM signal. This gives us a measure of how much the frequency of the FM signal changes, ie this is a measure of the amount of modulation.

Since the FM signal is periodic (for a sine wave input), we can perform a Fourier series expansion of the FM signal to find its spectrum. This analysis yields a Bessel function of the first kind. This tells us that the magnitude of the harmonics is eventually decreasing (ie the function does not monotonicly decrease, but it eventually gets smaller). In fact, the magnitude of the ( +1)th harmonic is negligible, so we assume that there are in fact only b harmonics.

Having done this analysis (skipping the messy mathematical details), we can say that the bandwidth of the FM signal is approximately 2( fm + fm). This equation is known as Carson's Rule and can be written as BW 2(kX + fm). We can thus see that the bandwidth of the FM signal depends on the size of the frequency deviations from the carrier frequency. Also, increasing the frequency of the input signal requires more bandwidth for transmission, which is also intuitive.

For general signals with maximum frequency deviation f (ie f = max{kX}), we can say that BW 2( f + fm). Note that fm is the maximum frequency of the input signal (ie the input bandwidth is 2fm), but this is not necessarily the maximum frequency deviation, since this depends of the magnitude of the Fourier transform.

Phase Modulation involves changing the phase of the carrier signal based on the input g(t). In this case, we use (t) = 2 fCt + kg(t). This is the same as frequency modulation, but with

Thus, we can see that phase modulation and frequency modulation are very similar, so the analysis presented applies to both. In fact, collectively the are described as Angle Modulation . The concept of phase modulation will be useful in later sections .

Demokratisasi Pemancar Radio

Written by BuSan    Friday, 17 November 2006 Media adalah kunci strategis proses demokratisasi. Media memungkinkan seseorang / sekelompok orang untuk mengekspresikan pendapatnya kepada banyak kalangan. Media juga memudahkan interaksi dalam sebuah komunitas, oleh komunitas, untuk komunitas itu sendiri. Proses ini mungkin mudah di implementasikan di media cetak, yang praktis

tidak menggunakan sumber daya alam yang terbatas. Pada media radio, cerita menjadi lain, karena kita menggunakan sumber daya alam yang terbatas, yaitu frekuensi radio.

Pada hari ini, FM dengan alokasi frekuensi 88-108MHz dengan spasi 350KHz antar pemancar merupakan band favorit diantara seluruh band radio siaran yang ada. Kapling band FM di Jakarta bahkan naga-naganya sudah penuh, sehingga ?tidak mungkin? menerima sebuah radio siaran baru kecuali dilakukan realokasi frekuensi.

 Band radio siaran sebetulnya masih banyak, AM atau MW pada frekuensi 535-1705KHz biasanya kosong karena tidak ada yang berminat memancar pada band ini. AM cukup baik untuk penggunaan radio siaran terutama di tepi laut, yang akan menempuh jarak sangat jauh. Pada gelombang pendek (SW), juga terdapat alokasi band radio siaran pada frekuensi 2.300-2.495MHz, 3.2-3.4MHz, 3.9-4.0MHz, 4.75-5.06MHz, 5.95-6.20MHz, 7.1-7.3MHz, 9.5-9.9MHz, 11.65-12.05MHz, 13.6-13.8MHz, 15.1-15.6MHz, 17.55-17.9MHz, 21.45-21.85MHz, dan tertinggi pada 25.6-26.1MHz. Tentunya masih di tambah lagi siaran radio melalui satelit maupun Internet.

 FM praktis penuh di beberapa kota besar, tapi banyak kosong di kota-kota kecil maupun di kota kabupaten / kecamatan. Band AM & SW praktis tidak digunakan, sialnya, peralatan pemancar AM & SW sebetulnya jauh lebih sederhana dan jauh lebih murah daripada pemancar FM. Tapi memang tidak sebaik FM untuk menyiarkan lagu-lagu & entertainment.

 Konsep radio siaran komunitas, atau radio rakyat ini, sebetulnya sangat sederhana. Pada hari ini radio siaran swasta niaga lebih banyak di fokuskan untuk usaha, bisnis & mencari fulus yang halal. Konsekuensinya, kebutuhan komunitas yang terbatas seperti komunitas sekolah, warga masyarakat, komunitas nelayan, komunitas petani, komunitas pengrajin yang lingkupnya terbatas dan terkadang tidak terakomodasi oleh radio siaran swasta niaga. Berdasarkan kebutuhan tersebut, muncul kebutuhan untuk membangun radio siaran komunitas, tentunya dengan kebutuhan yang ada jarak pancarnya maupun pembebanan frekuensi dapat menjadi sangat terbatasi.

 Sebetulnya pada hari ini teman-teman radio komunitas sudah mulai memancarkan diri di band FM dengan pemancar buatan sendirinya di Wanayasa, di Jogya, di banyak kota-kota kecil di Indonesia. Mereka umumnya tergabung pada Jaringan Radio Komunitas. Band FM menjadi favorit karena kualitas pancarannya yang baik. Rangkaiannya tersedia dengan mudah di Internet, salah satunya adalah buatan Andik yang dapat dilihat rangkaian & disain papan rangkaiannya di RBS Skematik .

 Bagi pembaca yang penasaran dengan teknik pemancar FM, ada baiknya mencari di search engine Internet, seperti www.google.com. Gunakan keyword "pirate FM", anda akan menemukan 36,000 situs yang berisi berbagai rangkaian & informasi untuk membuat pemancar FM sendiri. Yang perlu di garis bawahi, pemancar-pemancar FM buatan sendiri ini dapat dibuat dengan mudah & murah, tidak lebih dari satu (1) juta rupiah setiap pemancarnya, bahkan versi yang berdaya rendah dapat dibuat beberapa ratus ribu rupiah, sangat terjangkau untuk sebuah sekolah atau kelompok warga untuk mulai

memancar.

 Pertanyaan mendasar yang perlu dipikirkan bersama, benarkah keberadaan pemancar komunitas ini merupakan pemborosan frekuensi? Bagaimana disain alokasi frekuensi, penataan frekuensi agar kebutuhan komunitas ini menjadi terpenuhi?

Jawaban sederhananya sebetulnya tidak sukar, jika sebuah frekuensi di duduki oleh sebuah transmitter yang sangat kuat (misalnya 10-30 KiloWatt) dengan jangkauan puluhan mungkin ratusan km, maka kepentingan banyak orang menjadi terkangkangi oleh satu pemancar tersebut. Tapi bayangkan jika atas kesepakatan bersama, kekuatan maksimum pemancar yang digunakan di batasi misalnya 10 Watt. Dengan perhitungan propagasi adanya redaman di udara di 89MHz, dengan ketinggian antenna sekitar 37 meter, akan di peroleh jarak pancar sekitar 5-10 km untuk dapat diterima di radio biasa yang tidak terlalu baik.

Artinya apa? Dalam setiap jarak 10-20 km, frekuensi tersebut dapat dipakai ulang oleh pemancar yang berlainan. Tanpa saling mengganggu antar pemancar walaupun ke duanya memancar pada saat yang bersamaan, pada frekuensi yang bersamaan. Konsep ini di sebut frekuensi re-use. Konsep frekuensi re-use ini merupakan jawaban yang telak mementahkan argumentasi bahwa terjadi pemborosan frekuensi oleh radio pemancar komunitas. Tentunya masing-masing pemancar harus mentaati batasan daya yang di sepakati, dan daya tersebut harus rendah supaya frekuensi re-use dapat dilakukan sebanyak mungkin. Tidak heran, radio komunitas di amerika serikat sana di kenal sebagai pemancar FM berdaya rendah (low power).

 Masalah lain yang perlu diperhitungkan juga adalah lebar band setiap pemancar. Pada saat ini band FM menggunakan standar spasi 350KHz antar pemancarnya untuk mencapai kualitas suara yang sangat baik, konsekuensinya alokasi frekuensi untuk pemancar di FM 88-108MHz menjadi sekitar 56-57 pemancar saja.

 Dengan kualitas suara yang baik (lebar band 350KHz setiap pemancar), jika kita mengorbankan 88-90MHz untuk pemancar radio komunitas, maka kita akan memperoleh alokasi untuk 5 pemancar sekaligus yang dapat bekerja bersamaan dalam sebuah kecamatan atau kelurahan dalam radius pancaran 10-20 km. Jika di Indonesia ada sekitar 4000-an kecamatan (400-an kabupaten), dapat dengan mudah kita membangun 20.000 pemancar radio komunitas seluruh Indonesia.

 Jika kita sepakat untuk menurunkan sedikit kualitas suara, dengan menurunkan index modulasi sehingga sebuah pemancar hanya menggunakan lebar band 200KHz, maka 88-89MHz dapat dengan mudah di penuhi oleh 10 pemancar sekaligus. Konsekuensinya kita akan melihat 40.000 pemancar radio komunitas, sebuah kekuatan warga yang sangat besar yang tidak dapat di kesampingkan begitu saja oleh siapapun.

Dengan semakin banyaknya radio pemancar komunitas, akan lebih banyak lagi bangsa Indonesia terpenuhi hak asasinya untuk dapat berkomunikasi dan memperoleh informasi

untuk mengembangkan pribadi dan lingkungan sosialnya, seperti tertuang pada Bab VI Hak Atas Kebebaskan Informasi, Pasal 20 dari Ketetapan MPR XVII/MPR/1998. 

Semoga para eksekutif, aparat & wakil rakyat Indonesia masih ingat kesepakatan yang tertuang di atas.Last Updated ( Tuesday, 24 July 2007 ) Basics of Stereo Multiplexing Written by BuSan    Friday, 17 November 2006 One of the promises of FM in the early days was the possibility of transmission in Stereo. This stemmed directly from some of FM's properties :

Better rejection than AM of spurious signals from your girlfriend's Hair Dryer Electrical sparks, Lightning etc are normally AM sources, rejected by FM receivers

More distance on the dial between FM stations FM Stations are 200 KHz away from each other. FM Bandwidth = 2 ( deviation + input signal frequency). As FM is designed to handle Audio from 20Hz to 15 KHz, this gives lots of room to put in the Audio spectrum twice over + even more information in one channel without infringing on the other channel. AM stations on the other hand are spaced 9 or 10 KHz away from each other.

So the FM Gurus set out to decide on standards on how to use the available bandwidth to broadcast both the Left and the Right Channel of a Stereo Broadcast on the same FM Channel. The process of combining multiple signals onto one composite signal in such a way that the the original signals can be reconstituted by the receiver is called Multiplexing.

One of the design goals was that the Stereo Multiplexed signal ( MPX) was to be compatible with the earlier Mono FM Radios. As older FM Radios only looked at the 0-15KHz part of the incoming FM signal, it was a straightforward decision that the lower part of the Stereo MPX should contain the sum of the Left and Right channels i.e. L+R. In this way, the Mono FM Receiver would not notice any change, it would just play back the L+R signal. One would obviously not get a Stereo sound stage but on the other hand, one would not loose any information from the L and the R channels.

If one is already transmitting a L+R channel and one finally needs the L and the R channels separately, an option would be to transmit the L signal signal hidden somewhere higher than 0-15 KHz as part of the Stereo MPX signal.

The receiver could then direct the L Channel from the MPX signal directly to the Left Amplifier. To regenerate the Right Channel, the Receiver would convert L into -L by an Inverting Amp, Add L+R to -L and extract a R signal i.e. the original Right channel.

However, FM scientists found a more elegant approach that above. They decided to transmit an L-R signal as part of the MPX signal. If the receiver adds the L+R signal to

the L-R signal, it gets 2L i.e. the Left signal amplified two times. If the receiver subtracts L+R and the L-R signal, it gets 2R, i.e. the Right signal amplified two times. Summing and subtracting signals is easy with just simple Summing and Difference Amps.

As the 0-15 KHz part of the MPX signal has already been occupied by the L+R signal, one now has to think up a way of accommodating the L-R signal somewhere higher up in the bandwidth allocated for the MPX signal. One way to shift the L-R signal is to actually use it to amplitude modulate a higher frequency signal. The higher frequency signal used in this way is then called a Subcarrier.

Here we need to know two rules of Electronics :

A) If you Amplitude Modulate a carrier of frequency fc Hz with Audio that has a bandwidth of 0-fmaxa Hz, you will get side bands extending from fc-fmaxa right up to fc+fmaxa. All the Audio information we had from 0 to fmaxa is now stored in the side bands from fc-fmaxa to fc+fmaxa.

B) The carrier frequency has to be at least 2 times the highest modulation frequency otherwise one does not have enough "samples" to reproduce the original signal.

If we intend to transmit Audio upto 15 KHz, the Subcarrier frequency for the L-R channel has to be higher than 30 KHz otherwise, from (A) above, the side bands of the L-R channel will overlap the L+R signal. Also from (B) above, if the Subcarrier frequency is below 30KHz, you cannot reproduce signals of 15KHz.

We actually do not need the AM Subcarrier at fc to be transmitted at all. It contains no information. It actually has a lot of energy in it and we would just be wasting our FM transmitter's power by sending out an internal signal that contains no information. For an AM transmission, we would need an actual carrier to carry the signal across the airwaves, but in our application, we need the AM Subcarrier only to modulate it and generate the side bands storing L-R information. The actual carrier is the FM frequency we are transmitting on. ( Such an AM signal is called a Double Side Band Suppressed Carrier DSBSC Signal )

However, the receiver needs to regenerate the AM Subcarrier at the exact frequency and phase as the original so that it can extract the side band information. We cannot just use a local oscillator at the agreed upon AM Subcarrier frequency in the Receivers, since it would be too hard for all FM Receivers to exactly generate the same frequency and phase.

There is also the issue of somehow adding a tiny signal in the MPX signal that informs the Receiver that the incoming signal is in Stereo, so it should decode information higher than 15 KHz as that's not just Audio, it is the L-R signal hidden somehow. One simple way is to also transmit a sine wave as a pilot tone at a fixed frequency. If the receiver receives this tone, it can switch in its Stereo decoding circuitry, else it just treats the

incoming signal as Mono.

The Pilot tone has to be higher than 15 KHz as that part is used up by the L+R signal. If we choose a Pilot Tone somewhere in the 18-22 KHz range, the Mono Receiver will not decode it. Now comes a stroke of genius : Suppose we choose a Pilot Tone of 19 KHz, it can act as the signal informing the Receiver that the signal is Stereo. More-over, the Receiver can use this 19KHz signal to generate a 38KHz signal using a simple frequency doubler. As 38KHz is more than 2x 15KHz, it is good enough to act as our AM Subcarrier. The 38 KHz signal generated by the Receiver will be in phase with the 19 KHz pilot tone, so in case the Transmitter uses a 38 KHz Subcarrier in phase with the Pilot Tone, the Receiver would be able to exactly regenerate the missing AM SubCarrier.

The Side Bands would be from 38-15 KHz to 38+15 KHz i.e. 23 KHz to 53 KHz, so they would not overlap with the Pilot Tone or the L+R information.

By now we have finalized out how to generate the Stereo Multiplex signal. The MPX Coder / Transmitter will :

A) Add Left and Right signals to get a L+R signal.

B) Generate a Pilot Tone of 19 KHz.

C.1) Use an AM Subcarrier of 38 KHz in phase with the 19 KHz Pilot.

C.2) Amplitude Modulate the 38 KHz AM Subcarrier with a L-R signal

C.3) Cancel out the 38 KHz Subcarrier

D) Add up A, B and C above to get the complete MPX Signal. Use the above MPX signal to Frequency Modulate a carrier in the 88-108 MHz band.

A Mono receiver will :

A) React only to signals below 15 KHz so it would decode L+R

A Stereo Receiver will :

A) Sense the 19 KHz pilot signal so it will switch in the Stereo Decoder Circuitry

B.1) Decoder will extract the L+R signal from the MPX signal

B.2) Decoder will generate a 38 KHz signal by frequency doubling the received 19 KHz Pilot. It will ensure that the 38 KHz signal is in phase with the 19 KHz signal.

B.3) Decoder will demodulate the signals from 23KHz to 53 KHz to get back a L-R

channel

B.4) Decoder will use the L+R and the L-R signals to regenerate the original L and R signals

OK, the above scheme looks workable. Now comes the interesting part, How to actually generate the MPX signal at the transmitting end? The hard way is to actually follow faithfully the above instructions in building up the MPX signal. Have a Summing Amp to add the L + R. Have a Difference Amp create the L-R signal. Build a Double Balanced Modulator and feed it the L-R signal. Make a 38 KHz oscillator to feed the Double Balanced Modulator. Digitally divide the 38 KHz signal to get a 19 KHz pilot tone. Build a final summing amp that will add L+R, the 19 KHz pilot and the modulated L-R to get the final MPX signal.

It can however be shown ( not by me though ;) that rapidly switching between L and R channels at 38 KHz does most of the hard work and is a near equivalent to the hard way. Switching channels as above makes a L+R signal and a DSBSC L-R channel centered around 38 KHz. It also generates a lot of harmonics but they can be filtered away. All that is left is to add the Pilot Tone. Maybe the FM MPX standards were actually decided upon the other way around, the ease of the switching method lead to it being accepted as the standard ? Rapidly sampling L and R signals alternatively is called Time Division Multiplexing (TDM).

Winter 1933 : Edwin H. Armstrong demonstrates frequency modulation to executives and engineers of RCA. July 18, 1939 : First day of regular programming for W2XMN Alpine (Armstrong), 42.8 MHz, 35,000 watts. Only 25 FM receivers in the world at that time !!!Dec. 8, 1940 : The first advertising contract for FM broadcasts was signed by the Longines Watch Company Jan. 15, 1945 : FCC announces allocations proposals, moving FM to 84-108 MHz, with 84-88 MHz reserved for noncommercial FM broadcasting. Apr. 24, 1961 : Broadcasting reports the FCC approves stereo multiplex standardsJune 1, 1961 : FM stereo broadcasting is authorized to begin; on this date the FCC received its first notifications of such regular operation, from WEFM Chicago and WGFM Schenectady. Above information from FM Broadcasting Chronology, an interesting page with many more facts.

Now we really need some graphics to illustrate the various waveforms involved. I have just begun playing around with Micro-Cap SPICE so at first I thought I would simulate a complete Stereo Multiplexor and show the various waveforms generated. However I soon realized that SPICE can show virtual waveforms based on expressions. In case I need to show a L+R signal, I do not really need to simulate a mixer. I can just ask SPICE to scope out the expression "V(Left)+V(Right)". Similarly, to show the chopped up MPX signal, all I need to do is ask for a plot of "IF(V(Chopper), V(Left),V(Right))"

In the first graph above, We have a Left Audio signal of a single Sine wave tone at 2 KHz , with 2 Vpp. The Right channel is a 8 KHz signal at 1.4 Vpp. We have a digital chopper in Blue running at 38 KHz. When the chopper is high, we follow the Left channel and when it is low, we follow the Right channel. The chopped up basic MPX signal is shown in Black.What we wanted is not really a chopped signal as above. We wanted a smooth sine wave kind of thing whose amplitude varies between the instantaneous amplitude of the Left and the Right channels. The chopped up waveform as above has sharp edges so it will have a lot of odd harmonics. Let's run a harmonic analysis on the chopped MPX signal.....

Here we show on the top, the original chopped MPX signal. At the bottom is the harmonic analysis of the same signal. We can see the 2 KHz and 8 KHz signals in the L+R part of the MPX. Further on we see, centered around 38 KHz, the side bands of the L-R signal. The original 8 KHz signal transforms to 38-8 = 30 KHz and 38+8 = 46 KHz signals. Similarly the 2 KHz signal transforms to signals of 36 and 40 KHz. The 38 KHz carrier signal is most obvious in its absence.

As predicted, we also see strong odd harmonics of the DSBSC signal, centered around 114, 190, 266 KHz and so on. These extraneous harmonics dirty up the signal. Extra energy in the wrong place means that we waste transmitter power and we also limit the maximum modulation deviation for the signals we really want to transmit. They also use up space that could have been used for other signals as part of the MPX bundle. Besides, I am not sure how this makes the final Audio sound. Clearly, we need a Low Pass Filter around 57 KHz to remove the harmonics in the MPX signal.

( Incidentally , the BA1404 chip works by using exactly the same chopping method described above. BA1404 Kits could have used an external LPF on the MPX signal for a much better performance. Unfortunately most kit manufacturers decided to not take the design trouble, they lowered costs by eliminating the vital MPX LPF)

The smoother red line in the top chart is the MPX signal after it has gone through a very simple multistage RC Low Pass Filter. At the bottom, we see the harmonic content of the MPX signal after the LPF. We can see much lower harmonics around 114 KHz while all other higher harmonics have been almost totally suppressed. However the LPF has introduced enormous phase shifts, more about which follows later. A better LC LPF could have been used instead for better results.

One way of looking at the MPX generation method is that it is simply sampling the L and the R channels at 38 KHz. A standard method of reducing extraneously generated harmonic content in sampled Analog signals is to increase the number of samples. The first thought that comes to mind is to increase the sampling frequency. While this idea works on many other ocassions, we cannot use it for FM MPX because shifting the 38 KHz carrier upwards would use up more bandwidth and we would have a larger wasted gap between the L+R and the L-R part of the signal. So what we need to do is intrapolate between the 38 KHz samples.

The examples above show that we use the Left signal in case the Chopper signal is high and Right signal in case the Chopper signal is low. In each period, we read the L signal once and the R signal once i.e. there are two states. We get an odd looking wave that

appears as if it is made up of square wave segments. As expected, it is high in 3rd and 5th order harmonics.

If we look at the red curve showing the same chopped MPX signal passed through a LPF, we see that what we really wanted was a smoother sine-like curve with amplitude oscillating between the instantaneous L and R signals. If we took weighted samples, let's say 8 times per period, we would get a "Digital Sine" wave. This would be much lower in harmonics than a square wave because it looks much closer to the signal that we really want.

If we break one period into 8 time intervals, each bit will be 45 degrees. SIN(45) is 0.707, so we multiply the sample taken at 45 degrees by 0.707.

So instead of old algorithm of just switching between full L and full R,

IF(V(Chopper), V(Left),V(Right))

we now use following snippet from SPICE :

.define WTD_MPX IF((tm2>=0 AND tm2<45), V(Left)*.707+ V(Right)*.293,+ (IF((tm2>=45 AND tm2<135), V(Left),+ (IF((tm2>=135 AND tm2<180),V(Left)*.707+ V(Right)*.293,+ (IF((tm2>=180 AND tm2<225),V(Right)*.707+ V(Left)*.293,+ (IF((tm2>=225 AND tm2<315),V(Right),+ (IF((tm2>=315 AND tm2<=360),V(Right)*.707+ V(Left)*.293,+ 0)))))))))))

We need only 4 different states to generate the new waveform.

I was amazed at the reduction of Harmonic content caused by this simple weighted oversampling technique !!

The top part shows the Left Audio, the Right Audio, Basic chopped MPX and oversampled weighted MPX signal all on top of each other. The Bottom part shows the harmonic analysis of the MPX and the weighted MPX signals. The results of oversampling are dramatic indeed, with the raw oversampled signal looking better than the basic MPX through a LPF. We find that the first serious harmonic in the weighted MPX signal is around 266 KHz ie the 7th harmonic. As that is so far away from the 0-53 KHz signals we want, a gentler MPX filter with lower phase shifts in the 0-53KHz can now be designed, leading to better channel separation.

Even better results would have been possible if we oversampled at a higher rate i.e. used 8 states and weight sampled at 608 KHz instead of the 4 states weight sampled at 304 KHz as shown. Professional Stereo Coders do use 8 State sampling.

Exactly the same weighted sampling technique can be used to generate the 19 KHz Pilot. If we use a digitally generated Square wave like in the BA1404 and many kits, it contains a lot of odd harmonics. If we tried to create a "Digital Sine Wave" by making weighted samples at 45 degree intervals in each period, we would get a final Pilot tone with drastically reduced harmonics, much closer to the sine wave we really wanted, but with greater stability ( Locked to a Crystal digitally) than an analog generated Sine wave.

So far, we have used only Sine Wave inputs to study the MPX waveform. Let's try and simulate a bit of music going through a basic chopped MPX with a low pass filter.....

The Black line in the top half is the basic chopped MPX signal passed through a LPF, but no Pilot Tone has been added. In the Harmonic Analysis graph, we can easily make out the L+R region between 0 and 15 KHz, followed by the L-R region between 23 to 53 KHz.

We can see that the area between 15 KHz and 23 KHz is not blank as we would have imagined. All musical instruments create harmonics. Even a Bass drum (which we would assume contains only low frequencies) is actually rich in higher harmonics at the moment when the drum is struck. If you play a CD with 20 KHz information, it will ruin the MPX signal due to overlapping the 19 KHz Pilot. Obviously one needs pretty sharp 15 KHz Low Pass Filters on both Audio channels before they are fed into an MPX generator. Such sharp filters are called brickwall filters, maybe because they suddenly and sharply stop frequencies past their cutoff frequency analogous to one driving a car into a brickwall.

One of the interesting effects of the MPX process is non-harmonic distortion in case of

clipping. Lets analyze how this comes about. Suppose we have a 10 KHz signal that is clipped. It will create harmonic distortion at 2x, 3x etc. The third harmonic will fall at 30 KHz. But that's right in the territory of the L-R signal !! As a 30 KHz signal is 8 KHz away from the 38 KHz AM carrier, a stereo receiver would be within its rights to reproduce a 8 KHz tone during decoding process. We have mysteriously created a 8 KHz signal from a clipped 10 KHz signal !! This is called "Aliasing".

It is also interesting to study the effects of phase shifts within the Audio Chain. Suppose a filter in the Audio Chain shifts the phase around 30 KHz, it will shift part of the L-R signal. Let's call the phase shifted signal Lp-Rp. When we add L+R and Lp-Rp to try and get the 2L signal, we see that R and Rp will not cancel each other exactly even if they are of the same Amplitude. A small component of the Right channel will remain in the Left Channel. Similarly a small bit of the Left channel will remain in the right channel. So a phase shift in the Audio MPX Chain causes loss of Channel Separation !!

This is a serious issue : If you dont use oversampling techniques, the nearest unwanted harmonic in the signal is the 3rd harmonic around 114 KHz. To try and filter that by about 20 dB will need a filter that would probably cause a phase shift around 50-70 degrees around 45 KHz upwards. This will make you loose about 6-10 dB of separation. So if the rest of the electronics gave you 50 dB of separation, the MPX filter will reduce that to 40-44 dB.

As the phase shift could also have happened in the other stages following the Stereo Multiplexer (Transmitter's PLL section or even at the Receiver), one would tend to unnecessarily place the whole blame on the Multiplexer.

Similarly, a change in gain over the 0-53 KHz region will also negatively affect the Channel Separation. Let's analyse how that comes about.

Suppose we have a Multiplexing stage that has 0 dB gain from 0 to 19KHz and then -0.1 dB from there to 53 KHz.

So the L-R part of the MPX signal will be 0.1 dB down. This means that the signal is less by a factor of 0.988 times

Suppose we inject enough Audio to get 1L + 1R in the L+R part. The Decoder will add the L+R and the L-R signals to get a 2L

But we will get (1 L + 1 R) + (0.988 L - 0.988 R)

= 1.988 L + 0.0114 R

Hey, we wanted a theoretical signal of 2L over here. But we see that loss of gain in the 19-53 KHz part of the signal --

A) Reduces our L ( Receiver needs more Audio Gain. Loss of S/N)

B) Puts in a bit of R into the L channel. (Loss of Channel Separation)

So to compensate for the less L, suppose we turn up the volume by 2/1.9885531 times.

We get 2L + 0.0115 R instead of 2L

We see that the MPX process has added 0.0115/2 of crosstalk ie -44.79 dB

Here's a small table re gain change vs. Channel Separation, calculated the same way as above.

Gain Change dB Channel Separation dB

-0.01 -64.79

-0.05 -50.81

-0.1 -44.79

-0.2 -38.77

-0.5 -30.81

-1 -24.80

-2 -18.81

-3 -15.34

-4 -12.91

So looks like we need a gain change below 0.05 db across 0 - 53 KHz for separation better than 50 dB. Phew !!!!

As changes in Gain and Phase shift are frequency dependant, the Channel Separation of a Stereo Decoder is frequency dependent. Normally Channel Separation is best in the 200-8000 Hz range, degrading sharply at both the low end and the high end.

A Stereo transmitter has to transmit various parts of the MPX signal and has to be set such that it does not over-modulate on the sum of all the MPX signal. A Mono Transmitter would be set up to allow maximum allowed deviation on the peaks of the L+R signal it is transmitting. The Stereo Transmitter would have to be turned down a bit as it also needs to consider the deviation caused by the L-R signals. Hence a Transmitter signal will travel less distance in Stereo than when it is transmitting in Mono. Put another way, if you need to start transmitting in Stereo, you need a more powerful transmitter.

A Stereo transmission tends to sound noisier that a Mono signal. This is mainly caused by the noise in the L-R channel. As the noise in the 23 KHz to 53 KHz segment is also brought down to the audible 0-15 KHz region by the decoding process, we now have

more noise than receiving the same signal in Mono. The decoder circuits in the receiver also contribute extra noise. On top of that, as MPX signals have more bandwidth than a Mono signal, the station has to use less modulation with MPX to remain in the deviation limits. All above tend to increase the noise. (Visit USENET message link below for more info)

The last step is to add in about 10% of the 19 KHz Pilot Tone. The MPX signals would look quite the same as the ones shown above; the Harmonic analyses would show a blip at the 19 KHz point. As an MPX signal is not really a simple Audio signal, you cannot Compress/Limit/Equalise/Pre-emphasise an MPX signal. All the audio processing should have been done before the MPX process. All you can do now is feed the MPX signal to the transmitter instead of the Mono Audio signal it had before,

and Voila, You are transmitting in glorious STerEO !!!FM Transmitters Written by BuSan    Friday, 17 November 2006

MODULATORS

There are two types of FM modulators - direct and indirect. Direct FM involves varying the frequency of the carrier directly by the modulating input. Indirect FM involves directly altering the phase of the carrier based on the input (this is actually a form of direct phase modulation.

Direct modulation is usually accomplished by varying a capacitance in an LC oscillator or by changing the charging current applied to a capacitor.

The first method can be accomplished by the use of a reverse biased diode, since the capacitance of such a diode varies with applied voltage. A varactor diode is specifically designed for this purpose. Figure 1 shows a direct frequency modulator which uses a varactor diode.

This circuit deviates the frequency of the crystal oscillator using the diode. R1 and R2 develop a DC voltage across the diode which reverse biases it. The voltage across the diode determines the frequency of the oscillations. Positive inputs increase the reverse bias, decrease the diode capacitance and thus increase the oscillation frequency. Similarly, negative inputs decrease the oscillation frequency.

The use of a crystal oscillator means that the output waveform is very stable, but this is only the case if the frequency deviations are kept very small. Thus, the varactor diode modulator can only be used in limited applications.

The second method of direct FM involves the use of a voltage controlled oscillator, which

is depicted in figure 2.

The capacitor repeatedly charges and discharges under the control of the current source/sink. The amount of current supplied by this module is determined by vIN and by the resistor R. Since the amount of current determines the rate of capacitor charging, the resistor effectively controls the period of the output. The capacitance C also controls the rate of charging. The capacitor voltage is the input to the Schmitt trigger which changes the mode of the current source/sink when a certain threshold is reached. The capacitor voltage then heads in the opposite direction, generating a triangular wave. The output of the Schmitt trigger provides the square wave output. These signals can then be low-pass filtered to provide a sinusoidal FM signal.

The major limitation of the voltage controlled oscillator is that it can only work for a small range of frequencies. For instance, the 566 IC VCO only works a frequencies up to 1MHz.

A varactor diode circuit for indirect FM is shown in figure 3.

The modulating signal varies the capacitance of the diode, which then changes the phase shift incurred by the carrier input and thus changes the phase of the output signal. Because the phase of the carrier is shifted, the resulting signal has a frequency which is more stable than in the direct FM case.

TRANSMITTERS

As previously stated, if a crystal oscillator is used to provide the carrier signal, the frequency cannot be varied too much (this is a characteristic of crystal oscillators). Thus, crystal oscillators cannot be used in broadcast FM, but other oscillators can suffer from frequency drift. An automatic frequency control (AFC) circuit is used in conjunction with a non-crystal oscillator to ensure that the frequency drift is minimal.

Figure 4 shows a Crosby direct FM transmitter which contains an AFC loop. The frequency modulator shown can be a VCO since the oscillator frequency as much lower than the actual transmission frequency. In this example, the oscillator centre frequency is 5.1MHz which is multiplied by 18 before transmission to give ft = 91.8MHz.

When the frequency is multiplied, so are the frequency and phase deviations. However, the modulating input frequency is obviously unchanged, so the modulation index is multiplied by 18. The maximum frequency deviation at the output is 75kHz, so the maximum allowed deviation at the modulator output is

Since the maximum input frequency is fm = 15kHz for broadcast FM, the modulation index must be

The modulation index at the antenna then is = 0.2778 x 18 = 5.

The AFC loop aims to increase the stability of the output without using a crystal oscillator in the modulator.

The modulated carrier signal is mixed with a crystal reference signal in a non-linear device. The band-pass filter provides the difference in frequency between the master oscillator and the crystal oscillator and this signal is fed into the frequency discriminator. The frequency discriminator produces a voltage proportional to the difference between the input frequency and its resonant frequency. Its resonant frequency is 2MHz, which will allow it to detect low frequency variations in the carrier.

The output voltage of the frequency discriminator is added to the modulating input to correct for frequency deviations at the output. The low-pass filter ensures that the frequency discriminator does not correspond to the frequency deviation in the FM signal (thereby preventing the modulating input from being completely cancelled).

Indirect transmitters have no need for an AFC circuit because the frequency of the crystal is not directly varied. This means that indirect transmitters provide a very stable output, since the crystal frequency does not vary with operating conditions.

Figure 5 shows the block diagram for an Armstrong indirect FM transmitter. This works by using a suppressed carrier amplitude modulator and adding a phase shifted carrier to this signal. The effect of this is shown in figure 6, where the pink signal is the output and the blue signal the AM input. The output experiences both phase and amplitude modulation. The amplitude modulation can be reduced by using a carrier much larger than the peak signal amplitude, as shown in figure 7. However, this reduces the amount of phase variation.

The disadvantage of this method is the limited phase shift it can provide. The rest of figure 5 shows the frequency shifting to the FM broadcast band by means of frequency multiplication (by a factor of 72), frequency shifting and frequency multiplication again. This also multiplies the amount of phase shift at the antenna, allowing the required phase shift to be produced by a small phase variation at the modulator output.

Note that a phase modulator can be used as a frequency modulator if the input signal is integrated with respect to time (see FM principles section).